






t 



SEP 2 3)96tf 

f d "“ e, “”»2*w ae 

Lm RARY~ojr~ 

















































































































































































































































































































SUMMARY TECHNICAL REPORT 
OF THE 

NATIONAL DEFENSE RESEARCH COMMITTEE 



* * ec reta» 

aty Of 

$EP 2 ? 


^ess 


This^cfflWI^^^ontains inform^®# affecting the national defense of 
the United Sto^l^Wain the me^Kng of the Espionage Act, 50 U. S. C., 
31 and 32, as amenaSfc|ts taMPnission or the revelation of its con- 
tents in any manner to ai^l^Kthorized person is prohibited by law. 

This volume is classifiedaccordance with security 
regulations of the War a^^Wavy Depar^^^s because certain chap¬ 
ters contain material wjnH was at the date of 

printing. Other chapterj|Kiy have had a lower clas^||ation or none. 
The reader is advised t||wisult the War and Navy ageiW^tlisted on 
the reverse of this pagjJJr the current classification of anj^^terial. 


Manuscript and illustrations for this volume were pre¬ 
pared for publication by the Summary Reports Group of the 
Columbia University Division of War Research under con¬ 
tract OEMsr-1131 with the Office of Scientific Research and 
Development. This volume was printed and bound by the 
Columbia University Press. 

Distribution of the Summary Technical Report of NDRC 
has been made by the War and Navy Departments. In¬ 
quiries concerning the availability and distribution of the 
Summary Technical Report volumes and microfilmed and 
other reference material should be addressed to the War 
Department Library, Room 1A-522, The Pentagon, Wash¬ 
ington 25, D. C., or to the Office of Naval Research, Navy 
Department, Attention: Reports and Documents Section, 
Washington 25, D. C. 


Copy No. 

239 


This volume, like the seventy others of the Summary Tech¬ 
nical Report of NDRC, has been written, edited, and printed 
under great pressure. Inevitably there are errors which 
have slipped past Division readers and proofreaders. There 
may be errors of fact not known at time of printing. The 
author has not been able to follow through his writing to 
the final page proof. 

Please report errors to: 

JOINT RESEARCH AND DEVELOPMENT BOARD 
PROGRAMS DIVISION (STR ERRATA) 

WASHINGTON 25, D. C. 

A master errata sheet will be compiled from these reports 
and sent to recipients of the volume. Your help will make 
this book more useful to other readers and will be of great 
value in preparing any revisions. 


SUMMARY TECHNICAL REPORT OF DIVISION 13, NDRC 


VOLUME 1 


DIRECTION FINDER 
AND ANTENNA RESEARCH 


OFFICE OF SCIENTIFIC 



VANNEYAR BUSH, DIRECTOR 


NATIONAL DEFENSE RESEARCH COMMITTEE 
JAMES B. CONANT, CHAIRMAN 


DIVISION 13 
HARADEN PRATT, CHIEF 



WASHINGTON, D. C., 1946 







NATIONAL DEFENSE RESEARCH COMMITTEE 

James B. Conant, Chairman 
Richard C. Tolman, Vice Chairman 
Roger Adams Army Representative 1 

Frank B. Jewett Navy Representative 2 

Karl T. Compton Commissioner of Patents 3 

Irvin Stewart, Executive Secretary 


1 Army representatives in order of service: 


Maj. Gen. G. V. Strong 
Maj. Gen. R. C. Moore 
Maj. Gen. C. C. Williams 
Brig. Gen. W. A. Wood, Jr. 

Col. E. A. 


Col. L. A. Denson 
Col. P. R. Faymonville 
Brig. Gen. E. A. Regnier 
Col. M. M. Irvine 
Routheau 


2 Navy representatives in order of service: 

Rear Adm. H. G. Bowen Rear Adm. J. A. Furer 
Capt. Lybrand P. Smith Rear Adm. A. H. Van Keuren 
Commodore H. A. Schade 
3 Commissioners of Patents in order of service: 
Conway P. Coe Casper W. Ooms 


NOTES ON THE ORGANIZATION OF NDRC 


The duties of the National Defense Research Committee 
were (1) to recommend to the Director of OSRD suitable 
projects and research programs on the instrumentalities 
of warfare, together with contract facilities for carrying 
out these projects and programs, and (2) to administer 
the technical and scientific work of the contracts. More 
specifically, NDRC functioned by initiating research 
projects on requests from the Army or the Navy, or on 
requests from an allied government transmitted through 
the Liaison Office of OSRD, or on its own considered 
initiative as a result of the experience of its members. 
Proposals prepared by the Division, Panel, or Committee 
for research contracts for performance of the work in¬ 
volved in such projects were first reviewed by NDRC, 
and if approved, recommended to the Director of OSRD. 
Upon approval of a proposal by the Director, a contract 
permitting maximum flexibility of scientific effort was 
arranged. The business aspects of the contract, including 
such matters as materials, clearances, vouchers, patents, 
priorities, legal matters, and administration of patent 
matters were handled by the Executive Secretary of 
OSRD. 

Originally NDRC administered its work through five 
divisions, each headed by one of the NDRC members. 
These were: 

Division A—Armor and Ordnance 
Division B—Bombs, Fuels, Gases, & Chemical Problems 
Division C—Communication and Transportation 
Division D—Detection, Controls, and Instruments 
Division E—Patents and Inventions 


In a reorganization in the fall of 1942, twenty-three 
administrative divisions, panels, or committees were 
created, each with a chief selected on the basis of his 
outstanding work in the particular field. The NDRC 
members then became a reviewing and advisory group 
to the Director of OSRD. The final organization was as 
follows: 

Division 1—Ballistic Research 

Division 2—Effects of Impact and Explosion 

Division 3—Rocket Ordnance 

Division 4—Ordnance Accessories 

Division 5—New Missiles 

Division 6—Sub-Surface Warfare 

Division 7—Fire Control 

Division 8—Explosives 

Division 9—Chemistry 

Division 10—Absorbents and Aerosols 

Division 11—Chemical Engineering 

Division 12—Transportation 

Division 13—Electrical Communication 

Division 14—Radar 

Division 15—Radio Coordination 

Division 16—Optics and Camouflage 

Division 17—Physics 

Division 18—War Metallurgy 

Division 19—Miscellaneous 

Applied Mathematics Panel 

Applied Psychology Panel 

Committee on Propagation 

Tropical Deterioration Administrative Committee 


iv 


Library of Congress 



2015 


490940 














NDRC FOREWO 


LC R E GUL ATION: BEFORE SERVICING 
XJR REPRODUCING ANY PART OF THIS 
TJUCUMENT, ALL CLASSIFICATION 
MARKINGS MUST RE CANCELLED. 


A S events of the years preceding 1940 re- 
L vealed more and more clearly the serious¬ 
ness of the world situation, many scientists in 
this country came to realize the need of or¬ 
ganizing scientific research for service in a 
national emergency. Recommendations which 
they made to the White House were given care¬ 
ful and sympathetic attention, and as a result 
the National Defense Research Committee 
[NDRC] was formed by Executive Order of 
the President in the summer of 1940. The mem¬ 
bers of NDRC, appointed by the President, 
were instructed to supplement the work of the 
Army and the Navy in the development of the 
instrumentalities of war. A year later, upon 
the establishment of the Office of Scientific Re¬ 
search and Development [OSRD], NDRC be¬ 
came one of its units. 

The Summary Technical Report of NDRC is 
a conscientious effort on the part of NDRC to 
summarize and evaluate its work and to pre¬ 
sent it in a useful and permanent form. It 
comprises some seventy volumes broken into 
groups corresponding to the NDRC Divisions, 
Panels, and Committees. 

The Summary Technical Report of each Di¬ 
vision, Panel, or Committee is an integral sur¬ 
vey of the work of that group. The first volume 
of each group’s report contains a summary of 
the report, stating the problems presented and 
the philosophy of attacking them, and sum¬ 
marizing the results of the research, develop¬ 
ment, and training activities undertaken. Some 
volumes may be “state of the art” treatises 
covering subjects to which various research 
groups have contributed information. Others 
may contain descriptions of devices developed 
in the laboratories. A master index of all these 
divisional, panel, and committee reports which 
together constitute the Summary Technical Re¬ 
port of NDRC is contained in a separate vol¬ 
ume, which also includes the index of a micro¬ 
film record of pertinent technical laboratory 
reports and reference material. 

Some of the NDRC-sponsored researches 
which had been declassified by the end of 1945 
were of sufficient popular interest that it was 
found desirable to report them in the form of 
monographs, such as the series on radar by 
Division 14 and the monograph on sampling 
inspection by the Applied Mathematics Panel. 
Since the material treated in them is not dupli¬ 
cated in the Summary Technical Report of 
NDRC, the monographs are an important part 
of the story of these aspects of NDRC research. 


In contrast to the information on radar, 
which is of widespread interest and much of 
which is released to the public, the research on 
subsurface warfare is largely classified and is 
of general interest to a more restricted group. 
As a consequence, the report of Division 6 is 
found almost entirely in its Summary Tech¬ 
nical Report, which runs to over twenty vol¬ 
umes. The extent of the work of a Division can¬ 
not therefore be judged solely by the number 
of volumes devoted to it in the Summary Tech¬ 
nical Report of NDRC; account must be taken 
of the monographs and available reports 
published elsewhere. 

Of all the NDRC Divisions, few were larger 
or charged with more diverse responsibilities 
than Division 13. Under the urgent pressure 
of wartime requirements, the staff of the Di¬ 
vision developed navigation and communica¬ 
tions devices and systems which not only con¬ 
tributed to the successful Allied war effort, but 
will continue to be of value in time of peace in 
the fields of transportation and communica¬ 
tions. The work of the Division, under the di¬ 
rection first of C. B. Jolliffe and later of Hara- 
den Pratt, furnishes a foundation for what 
promises to be even more radical developments 
than those of the war—for one example, direc¬ 
tion finders which will operate at all elevations 
and azimuth angles, in other words, hemispher- 
ically. „ . . 

The Summary Technical Report of Division 
13 was prepared under the direction of the 
Division Chief and authorized by him for publi¬ 
cation. The report presents the methods and 
results of the widely varied research and de¬ 
velopment program, and, in the case of work 
with speech scrambling and decoding, it pre¬ 
sents for the first time a comprehensive review 
of the state of the art. The report is also a 
notable record of the skill and integrity of the 
scientists and engineers, who, with the coopera¬ 
tion of the Army and Navy and Division con¬ 
tractors, contributed brilliantly to the defense 
of the nation. To all of these we express our 
sincere appreciation. 


Vannevar Bush, Director 
Office of Scientific Research and Development 


SEP 2 3 Wee 
Defense memo 2 AugUi 
tJBRARY ~OF~c ONg 

























































FOREWORD 


N early sixty years ago Heinrich Hertz 
experimentally produced electromagnetic 
waves, determined the direction of the waves, 
and wrote, “Thus we now have a means of dis¬ 
cerning the direction of the electric force at 
every point.” The waves were not detected out¬ 
side of his lecture room, and it is unlikely that 
he foresaw the application of direction finding 
to navigation. Later, as the direction finder 
art advanced, many types of directional anten¬ 
nas were devised, including loops, crossed 
loops, spaced loops, Adcocks, and arrays. Some 
who contributed most effectively were Adcock, 
Ballantine, Barfield, Bellini, Busignies, Dellin¬ 
ger, Dieckmann, Eckersley, Heil, Roister, Mar¬ 
coni, Mesny, Pickard, Smith-Rose, and Tossi. 

During the fifteen years prior to World War 
II, the art advanced relatively slowly. Most 
progress was made in England. Equipment per¬ 
formance was reasonably satisfactory. Ground 
installations of direction finders were used to 
inform ships at sea of their positions. A similar 
use of ground direction finders was made by 
Pan American Airways and by various Euro¬ 
pean air lines. Direction finders on ships at sea 
were almost universally used as a navigational 
aid, and most commercial airliners employed 
automatic direction finders for navigation. 
Thus, by the advent of World War II, direction 
finding was established as an important means 
of navigation. 

Early in World War II, the Communications 
Division (Division 13) of the National Defense 
Research Committee [NDRC] formed a Direc¬ 
tion Finder Committee under the Chairman¬ 
ship of Loren F. Jones of which the members 
were H. Busignies, J. H. Dellinger, D. G. C. 
Luck, and R. K. Potter. Later, as consultants 
or technical aides, the Committee was greatly 
assisted by J. Allison, E. D. Blodget, and W. 
C. Lent. This Committee was active until Sep¬ 
tember 21, 1945, with a number of Army, 
Navy, and British liaison representatives at¬ 
tending each meeting. During this period, the 
Committee issued contracts for work at Stan¬ 
ford University, California Institute of Tech¬ 
nology, Harvard University, University of New 
Mexico, Federal Telephone and Radio Labora¬ 
tories, Radio Corporation of America, Wilmotte 
Laboratories, J. A. Maurer, Inc., and Bell Tele¬ 
phone Laboratories. In addition, the Committee 
served as a coordinating agent and a clearing 
house for direction finder developments every¬ 
where. The art advanced rapidly. Such diverse 
subjects as polarization errors, ionospheric 
effects, site errors, navigational applications, 
evaluation of fixes, and location of electric 
storms were studied. 

Despite its long history, direction finding has 
been the subject of remarkably few texts. For 


years, the standard text in English was Wire¬ 
less Direction Finding by R. Keen, published 
in England in 1922 and now undergoing its 
fourth revision. Radio Direction Finders by 
D. S. Bond was published in 1944. 

The present publication, for which Keith 
Henney has acted as general editor and has 
devoted much time to coordinating the materi¬ 
al, is Volume 1 of four books covering the war¬ 
time work of the Communications Division of 
NDRC. In this volume, there are accounts of 
developments sponsored by the Direction 
Finder Committee and of the results obtained. 
This book is not intended for the layman, and 
will be of only moderate assistance to equip¬ 
ment operators. It is intended for scientists, 
engineers, military personnel, students, and 
others who are interested in radio direction 
finding. 

Radar, which combines direction finding and 
ranging, is already extensively used for naviga¬ 
tion. To some extent, it will replace direction 
finding. However, direction finding will remain 
as one of the primary navigational methods 
and will be used for new functions such as 
locating electric storms. As the art advances, 
developments will facilitate direction finding 
at higher frequencies, will minimize errors, 
and will simplify equipment. Recent progress 
made in these directions by NDRC is outlined 
in the following pages. 

The future holds promise, of more radical 
developments, such as direction finders which 
will operate at all elevation and azimuth angles, 
in other words, hemispherically, with an ac¬ 
curacy adequate for fire control purposes. Pos¬ 
sibly all directions and frequencies will be 
under continuous uninterrupted observation 
with some kind of panoramic presentation. 
Possibly there will be a need for direction 
finders with automatic tracking wherein the 
equipment will lock on and automatically fol¬ 
low a moving source of emission. No doubt 
there will be still other developments not now 
envisioned. 

All radio communication, of course, involves 
the proper design and use of many components, 
among them antennas. Direction finding, radar, 
altimeters, and countermeasures for jamming 
enemy radio communication require means for 
imparting to and receiving from space the re¬ 
quired radio energy. For this reason it was 
natural that certain research on the design, 
measurement, and application of antennas 
should fall to Division 13 to sponsor. Following 
the material on direction finding will be found 
summaries of the several antenna projects 
supervised by the Division. 

Haraden Pratt 

Chief, Division 13 

vii 



































































































































































PREFACE 


I N SUMMARIZING the several hundred reports 
of contractors on the hundred-odd research 
projects sponsored by Division 13 of the Na¬ 
tional Defense Research Committee, [NDRC], 
the editor has had to settle in his own mind 
how much or how little of each project report 
should be included; in other words, how far the 
boiling-down process should go. 

The editor has an abhorrence for seeing good 
scientific or technical material go unpublished. 
Only by publication can the facts or methods 
developed by a few researchers become available 
for all researchers. On this basis, substantially 
all Division 13’s program should be included in 
the volumes, of which this is one, summarizing 
the work of the Division. On the other hand, 
time moves forward inexorably so that it is 
quite likely that, by the day of publication, much 
of the data would already be out of date. Fur¬ 
thermore, time and human energy are always 
scarce. On these bases, all that might be re¬ 
quired would be a paragraph or two summariz¬ 
ing the aims of the project and its accomplish¬ 
ments. 

A middle course was steered, a course be¬ 
tween the easiest solution of publishing prac¬ 
tically all of each report and the more difficult 
job of really digesting the project purpose and 
results. The editor, however, deliberately chose 
to publish too much rather than too little. In 
most cases it will be unnecessary for the reader 
to search out the original source material unless 
he wishes to dig deep into the subject. In those 
cases where fundamental information was as¬ 
sembled and printed in the project report, that 
is, information on which future research might 
be based, the summaries have been permitted 
to take as much space as required. 

This volume covers two aspects of Division 
13’s work—that dealing with research and de¬ 
velopment in direction finding, and that on an¬ 
tennas. The work on direction finders has been 
divided broadly into two aspects, that describ¬ 
ing physical equipment, and that covering fun¬ 
damental research leading to better knowledge 
of the manner in which ground constants, mul¬ 
tiple rays, polarization by the ionosphere, and 
other factors affect the accuracy with which 


bearings can be measured. All this was necessi¬ 
tated by the fact that direction finding had gone 
into a sort of intellectual slump by the begin¬ 
ning of World War II. Antennas were generally 
of the loop or the Adcock type. Errors in bear¬ 
ings were deplored but accepted. Need had not 
risen for direction finding on the higher fre¬ 
quencies which came into such wide use during 
World War II. Above all, new ideas, new and 
basic analytical research were needed. 

Throughout all the fundamental work on di¬ 
rection finding, the subject of errors was most 
important, simply because direction finders of 
various types do not give consistent nor accu¬ 
rate bearings in spite of the fact they can be 
erected with great care and constructed of pre¬ 
cision apparatus. In fact, exploration of the 
vagaries of direction finding occupied a great 
deal of the attention of the Division and its 
research men and engineers. Finally, through 
the means of a new instrument, the polariscope, 
it was proved that many d-f troubles are due, 
not to the apparatus itself, nor to the ground on 
which it is located, nor to the operation of the 
equipment, nor to the fact that the ionosphere 
polarized radio waves heterogenously. Many of 
the errors which would remain, even if all the 
other sources of difficulty were removed, come 
from the fact that radiation from a transmitter 
arrives at a receiving point over multiple paths, 
and it is the many possible interrelations be¬ 
tween these multiple rays that produce direc¬ 
tion-finding aberrations. Thus it appears that 
there is a point beyond which much greater ac¬ 
curacy in bearing determination cannot be ob¬ 
tained by refining the apparatus. 

Fundamental studies, analytical in nature, 
are reported rather fully in this report. Part I, 
dealing with basic studies in direction finding, 
includes means of measuring ground constants, 
and of rating d-f systems in terms of wanted-to- 
unwanted pickups; the effects of connecting 
cables with Adcock systems; a new means of 
controlling the amplification of a d-f receiver by 
means of a local transmitter; and the virtues of 
direction finding on pulse transmissions. 

Part II deals with physical equipment and 
systems developed under the aegis of Division 


IX 


X 


PREFACE 


13. Here will be found the work which led to 
the SCR-291, a single-band d-f system widely 
used by the Air Transport Service, a workable 
Radio-Sonde, a d-f system for the region of 140 
to 600 me, portable beacons which would lead 
a foot soldier to his objective on the field of 
battle regardless of weather or time of day, 
and means for locating tanks by radio. Finally, 
one of the last and most elegant accomplish¬ 
ments of the Division was an electrical and elec¬ 
tronic instrument for evaluating the responses 
obtained from multiple d-f receivers so that the 
origin of signals could be more closely pinned 
down to a circle of small radius. 

Part III records early work of sferics, the use 
of radio direction finding for locating storms. 


The portion of the Division’s activities deal¬ 
ing with antenna research is found in Part IV. 
Here is described the early work on determina¬ 
tion of the characteristics of antennas for air¬ 
craft and tanks by means of scaled-down 
models; the work on faired-in antennas; a com¬ 
plete survey of airborne antennas as of early 
1945, including what was then known about 
wide-band antennas. Work on disguised anten¬ 
nas, on improvised d-f antennas for use in the 
field, and on antennas for use in the region of 
150 to 600 me are also recorded here. 


Keith Henney 
Editor 




CONTENTS 


CHAPTER PART I PAGE 

STUDIES OF 

HIGH-FREQUENCY DIRECTION FINDING 

1 BTL High-Frequency Direction-Finder Research. 3 

2 NBS High-Frequency Direction-Finder Research.22 

3 Study of Radio Pulse Propagation.55 

4 Ultra-High-Frequency Direction-Finding Study.59 

5 Errors in Direction Finders.100 

6 Correlation of D-F Errors with Ionosphere Measurements . . 119 

7 Miscellaneous Direction-Finder Research.122 

PART II 

APPARATUS DESIGN 

8 U-H-F Radio-Sonde Direction Finder.127 

9 Demountable Short-Wave Direction Finder.136 

10 Direction Finding by Improvised Means.148 

11 Portable Radio Assault Beacon.100 

12 U-H-F Direction-Finding Antenna Study.172 

13 Locating Tanks by Radio.183 

14 U-H-F Friendly Aircraft Locator.185 

15 Electrical Direction-Finder Evaluator.190 

PART III 

RADIO AND WEATHER 

16 A Study of Sferics.197 

PART IV 

ANTENNA RESEARCH 

17 Antenna Patterns for Aircraft.203 

18 Airborne Antenna Design at U-H-F and V-H-F.219 

19 Development of Faired-in Antennas ..267 

20 Miscellaneous Antenna Research.274 

Bibliography.278 

OSRD Appointees.283 

Contract Numbers.284 

Service Project Numbers.286 

Index.287 




























































































































































































PART I 


STUDIES OF HIGH-FREQUENCY DIRECTION FINDING 


D uring the years immediately preceding 
World War II only a limited amount of 
basic research was devoted to the development 
of direction-finding [d-f] techniques and equip¬ 
ment. As in most other branches of scientific 
and engineering endeavor, the advent of the 
war accelerated such research first by making 
only too evident the need for it as related to 
ordinary peacetime applications, and second by 


bringing into important focus new uses for d-f 
equipment. Much information was required on 
the use of d-f technique for use on high radio 
frequencies, on the causes of and solutions for 
certain vagaries in high-frequency direction 
finding, on the correlations between d-f mea¬ 
surements and the state of the ionosphere which 
reflects back to earth radio frequencies most 
likely to be used during the war. 





Chapter 1 


BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


Research leading to the general design requirements 
for circular-array direction-finder systems and Adcock 
antennas, including a determination of antenna spacing 
to minimize interaction effects, requirements for buried- 
conductor arrangements in both systems, design of an¬ 
tenna elements and coupling units to extend the fre¬ 
quency range of operation, specifications for receiver 
for a crossed buried U antenna direction-finder system, 
setting up of a complete direction-finder system using 
commercial receivers, and development of a unique 
method of automatic gain control. The text herein is 
condensed from the contractor’s final report. 1 


1.1 STATE OF THE ART 

A t the time of this project/ 1 the most prom- 
L ising high-frequency direction finders from 
the standpoint of simplicity and ease of opera¬ 
tion were based upon the Adcock principle, but 
all such systems were subject at times to serious 
errors, the main cause being unwanted hori¬ 
zontal pickup in the antenna system. The most 
accurate high-frequency direction finders were 
the large fixed installations, but they were usu¬ 
ally complicated, clumsy, and slow in operation. 

The object of this project was to make a brief 
survey of the various types of high-frequency 
direction finders, to pick the most promising, to 
study the causes of the errors, and to determine, 
whenever possible, methods for reducing these 
errors. 

The conclusion was reached that, for fixed 
installations and where speed of operation was 
important, Adcock systems were most promis¬ 
ing. Accordingly, a crossed Adcock antenna 
system was designed and built. The errors in 
its receiving characteristics were studied, and 
methods were derived for their reduction. The 
final result was an Adcock antenna system with 
greatly reduced polarization errors. A receiving 
system was designed for operation with the 
antenna. 

a Project C-16, Contract No. NDCrc-155, Western 
Electric Company. 


12 INTRODUCTION 

Fundamentally, operation of all radio direc¬ 
tion finders depends upon the fact that the rela¬ 
tive phases of the currents induced by a radio 
wave in two or more fixed, spaced wires vary 
as the direction of arrival of the wave varies. 
In some systems this phase difference is mea¬ 
sured directly and the direction of arrival de¬ 
termined by a comparison of the measured 
phase difference with a previous calibration of 
phase difference versus direction. For conve¬ 
nience we will call this the phase-comparison 
method. The Navy’s CXK direction finder, for 
example, works by this method. In other sys¬ 
tems, instead of actually measuring the relative 
phases of the currents in the various antennas, 
the latter are connected together in such a way 
that, as a result of phase interference, different 
outputs are obtained from the antenna system 
for different directions of arrival. This will be 
called the amplitude-comparison method. Direc¬ 
tion finders which use the loop antenna, the 
Adcock or any of its variations, or a sharp di¬ 
rectional array are all examples of systems of 
this type. 

For either case the determination of the di¬ 
rection of arrival from the amplitude or phase 
difference is straightforward when the signal 
arrives over a single path. Long-distance short¬ 
wave radio transmission, however, usually takes 
place by several paths of continuously varying 
lengths. Due to the interference among the 
waves arriving over these various paths, in gen¬ 
eral the field strengths will not be identical at 
two or more spaced antennas and the phase dif¬ 
ferences will not be the same as for any of the 
component waves. However, if the directions 
of arrival are very nearly the same, the field 
strengths at the various antennas will also be 
very nearly the same and the phase differences 
very nearly what would have been obtained for 
a single wave arriving in the mean direction, 


3 





4 


BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


except for those periods when the relative phases 
of the component waves are such as to produce 
a weak signal at any place within the area oc¬ 
cupied by the antennas. At such times the rela¬ 
tive field strengths at the various antennas may 
differ greatly and the phase differences may dif¬ 
fer by as much as ±180° from the correct value. 
For these reasons accurate bearings cannot be 
obtained during the minima of a fading signal. 

The closer the antenna spacing the shorter 
will be the period when the relative field 
strengths will differ appreciably and when the 
phase differences will be incorrect. When the 
directions of arrival of the various waves are 
radically different, the field strength and phase 
difference will vary rapidly and considerably, so 
that, in general, direction finding with simple 
antennas consisting of only a few elements be¬ 
comes impossible.' 3 However, if the directions 
of arrival are confined to a single vertical plane, 
the azimuth of the direction of arrival may still 
be measured with certain types of direction 
finders such as the crossed Adcock, described 
later, although the periods of weak fields when 
correct bearings cannot be obtained will occur 
very frequently. 

13 PHASE-COMPARISON METHOD 

A simple form of direction finder using the 
phase-comparison method would be one consist¬ 
ing of two fixed vertical antennas connected, 
through appropriate receiving and amplifying 
equipment, to a phase-measuring device. One 
way of accomplishing this phase measurement 
is to introduce a phase shifter, either in the 
radio- or lower-frequency branches of one of 
the receivers. This phase shifter is then varied 
until the outputs of the two receivers cancel 
(differ by 180°). The phase difference between 
the currents in the antennas is then found by 
subtracting 180° from the phase-shifter reading. 

In Figure 1 let A and B represent two such 
antennas and let d be the distance between 
them. If a radio wave arrives at an angle f3 
with respect to the line AB, then the phase dif¬ 
ference cf> between the currents induced in the 


b Highly directive steerable antenna systems such as 
the Musa type are required for this type of transmission. 


two antennas will be given by the equation 

4> (degrees) —— ^ • cos ft (1) 

A 

where 

cos (3 — cos a cos 8 (2) 

where a and 8 are the horizontal and vertical 
angles of arrival respectively and A is the wave¬ 
length. 



Figure 1 . Diagram of simple phase-comparison 
direction finder. 


Since /3 is measured from the array axis, 
equation (1) represents a cone whose axis is 
the line joining the two antennas and whose 
generator is at an angle ft with respect to this 
axis. Thus all wave directions which lie in this 
cone will produce currents of the same phase 
difference in the two antennas, and, in general, 
additional information is needed to obtain the 
azimuth of the direction of arrival or the ap¬ 
parent bearing of the station. 

This information can be obtained from an¬ 
other pair of antennas having a different orien¬ 
tation. The measurements obtained with this 
second pair of antennas will determine another 
cone with a different axis than the first. The 
line of intersection of these two cones will coin¬ 
cide with the actual direction of arrival of the 
wave and will, accordingly, determine not only 
the azimuth of the direction of arrival, but the 
vertical angle of arrival as well. 

Disregarding, for the moment, the difficulties 
associated with the taking of two sets of data 


% 


ONFJDENTI4J 


I 












PHASE-COMPARISON METHOD 


5 


simultaneously, a satisfactory direction finder 
might be made using two pairs of antennas 
arranged in two lines mutually perpendicular. 
In fact three antennas arranged at three corners 
of a square would answer the purpose. 

On the other hand, if two vertical antennas 
are mounted on a structure which can be ro¬ 
tated about a vertical axis until the phase dif¬ 
ference between the currents in the two an¬ 
tennas is zero, then, if d is shorter than A the 
apparent bearing of the station will be perpen¬ 
dicular to the line joining the antennas. Such 
a system cannot distinguish between signals 
having bearings 180° apart. To remove this 
180° ambiguity requires the addition of a third 
antenna and greatly complicates the receiving 
equipment. This is the principle of the Navy s 
three-antenna CXK direction finder. One of the 
objections to this direction finder is the size of 
the rotating structure and the resulting time 
consumed in taking a bearing. 


Circular Array 

A variation of the foregoing scheme which 
overcomes the disadvantage just mentioned, at 
the expense of a slight decrease in accuracy, 
would make use of several fixed antennas spaced 
on the perimeter of a semicircle. These anten¬ 
nas would be used in pairs, any two adjacent 
antennas constituting such a pair. For making 
a measurement, that pair would be selected 
which was most nearly perpendicular to the 
direction of arrival and which, therefore, would 
give the smallest phase difference. 

Figure 2 shows such an arrangement con¬ 
sisting of 19 antennas, one pair for every 10°. 
A wave is shown arriving at a bearing of 57° 
for which pair FG would be used to obtain the 
bearing. 

The information obtained from a single pair 
of antennas is not, in general, sufficient to deter¬ 
mine the apparent bearing of a station. How¬ 
ever, if each pair is used to take bearings over 
only a small angular range approximately per¬ 
pendicular to the line joining the antennas, 
then the phase difference of the currents in the 
two antennas can be used to determine the 


apparent bearing with a reasonable degree of 
accuracy for all but very high vertical angles 
of arrival, except for an approximate 180° 
ambiguity which could be removed only by the 
use of additional equipment. For a system such 
as is shown in Figure 2, where each pair is 
used over a range of only 5° on each side of 
the perpendicular, the maximum error for dif¬ 
ferent vertical angles of arrival is given by 
curve A of Figure 3. Curve B gives the errors 


WAVE DIRECTION X* ANTENNAS 



Figure 2. Diagram of antennas arranged in semi¬ 
circle for direction finding. 


for a system consisting of 10 antennas, one 
pair for every 20°. It will be observed that for 
the occasional signal suspected of having a 
high vertical angle of arrival the error may be 
eliminated by taking an additional measure¬ 
ment with another pair of antennas, preferably 
that pair the axis of which is perpendicular to 
that of the first pair. If the phase-measuring 
device used for making this measurement is 
capable of operating over the full 360° range 
this measurement would also give the sense of 
the signal. In Figure 2, pair OP would be used 
to obtain this additional information. 

As the separation between the antennas is in¬ 
creased, any given value of <f> will correspond to 
smaller and smaller values of p [equation (1)]. 
Thus, for any given uncertainty in the value 
of <f> f the greatest accuracy in bearing deter¬ 
mination will be obtained with the largest pos¬ 
sible value of d. However, for systems using 
only two sets of antennas with axes mutually 
perpendicular, the separation must be kept to 























6 


BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


less than half the shortest wavelength upon 
which observations are to be taken. Otherwise, 
for some values of <j>, there will be more than 
one possible value of /3. For a system of several 
antennas located on a semicircle the separation 
may be increased somewhat as long as it is kept 
below a/2 sin 0, where 6 is the maximum angle 
on each side of the perpendicular at which 



O 10 20 30 40 50 60 70 80 90 

VERTICAL ANGLE OF ARRIVAL IN DEGREES 

Figure 3. Errors in semicircle system. A shows 
errors when each antenna pair is used over range 
of only 5° on each side of the perpendicular; 

B shows similar errors for ten-antenna system, one 
pair for every 20°. 

measurements will be made (5° for Figure 2). 
However, if this is done it will no longer be 
possible to make check measurements on the 
vertical angle of arrival with the in-line pair. 
For this reason it is recommended that the 
spacing always be kept below A/2. Then, if 
greater accuracy is desired, after the approxi¬ 
mate bearing is obtained a more widely spaced 
pair can be used to get a more accurate meas¬ 
urement. Thus in Figure 2, pair NQ could be 
used to get a more accurate measurement after 
a preliminary measurement is made with pair 
OP. 

Interaction Effects 

The accuracy of all the above systems de¬ 
pends on the accuracy with which the phase 
difference between the currents in the separate 


antennas is given by equation (1). Among 
other things this phase difference is affected 
by the interaction among the various antennas 
that make up the system. 

One of the objects of this project was the 
determination of the extent of this interaction 
and of the amount of error it would introduce 
in direction finders working by the phase-com¬ 
parison method. In this study, use was made 
of some of the antennas of the broadside cage 
Musa system at the Holmdel, N. J., laboratories 
of the Bell System. These were vertical cage 
antennas 2i/ 2 feet in diameter and 23i/ 2 feet 
high. They had a half-wave resonant impedance 
of about 300 ohms at 18.15 meters and a 
quarter-wave impedance of about 36 ohms at 
36.3 meters. The broad-band characteristic of 
this type of antenna makes it desirable for di¬ 
rection-finding systems which are to be used 
over a relatively wide frequency range. The 
low impedance makes it a simple matter to 
connect them to the receivers by means of low 
impedance, concentric transmission lines. The 
interaction will be a function of the dimensions 
of the antennas, but measurements were made 
with antennas of only one size since, in general, 
antennas used to cover the frequency range 
from 5 to 18 me would be of approximately 
the same dimensions. 

Figure 4 shows a ground plan of the antenna 
arrangement used for making these measure¬ 
ments. The antenna at point X was used as a 
reference antenna and all phases were meas¬ 
ured with reference to the phase of the current 
in that antenna. Antennas 4, 5, and 6 in Figure 
4 were antennas 4, 5, and 6 of the broadside 
Musa. They were fixed in location but could be 
easily lowered to the ground when not in use. 
These antennas as well as antenna X were all 
connected to buried coaxial transmission lines 
which terminated on the antenna termination 
panel in the Musa building and could, there¬ 
fore, be connected to the Musa phase measur¬ 
ing equipment. Another exactly similar cage 
antenna was carried on a trolley suspended be¬ 
tween the supporting poles of antennas 5 and 
6. It was always connected through an 80-ohm 
terminating resistance to one of several ground 
rods which had been driven into the ground at 
approximately 6-foot intervals between anten- 


























PHASE-COMPARISON METHOD 


7 


nas 5 and 6. Antennas 4, 5, and 6 were spaced 
49 feet (15 meters) apart. 

By comparing the output of antenna 5 with 
that of antenna X while moving the traveling 
antenna between 5 and 6, with 4 and 6 lowered 
to the ground, the effect of an interacting an¬ 
tenna at distances of 6 feet to 49 feet was 
determined. By comparing the output of an¬ 
tenna 4 with that of antenna X while moving 
the traveling antenna between 5 and 6, with 



100 FT 

^OSCILLATOR 

LOCATION A 



GROUND RODS FOR 

MOVABLE ANTENNA 
\ 

X 

• 

r— OSCILLATOR 
\ LOCATION C 

o 

5 500 FT TO ( 



^Vocation b * 


Figure 4. Arrangement of antennas for interac¬ 
tion tests. 


antennas 5 and 6 lowered to the ground, the 
effect of an interacting antenna at distances 
of 49 feet to 98 feet was determined. Measure¬ 
ments were made on five different wavelengths 
and with three different oscillator positions; 
one broadside to the antennas, at location A in 
Figure 4; one end-on, in the direction of the 
interacting antenna, at B\ and one end-on in 
the opposite direction, at C. 

In Figures 5, 6, and 7, the curves marked 
with open circles give the effect of the inter¬ 
action on the amplitude of the current in the 
fixed antenna, those marked with crosses give 
the effect on the phase of the current, and 
those marked with solid circles give the cor¬ 
responding error which the change in phase 
would introduce in the value obtained for /3 . 
For wavelengths between 16 and 64 meters a 
spacing of 49 feet does not introduce any sig¬ 
nificant error. This would be a perfectly satis¬ 
factory spacing for wavelengths greater than 


30 meters but for shorter wavelengths this 
spacing becomes greater than A/2 and would 
introduce an uncertainty in the bearings. For 
these shorter wavelengths a closer spacing is 
required, a spacing of 26 feet (8 meters) being 
satisfactory for wavelengths as short as 16 
meters. The interaction for antennas at this 
spacing would introduce only a small error for 




\ 








— 

468C 

1 KC 
































O 10 20 30 4 0 5 0 60 70 80 90 100 

DISTANCE BETWEEN ANTENNAS IN FEET 

Figure 5. Interaction effects between antennas; 
oscillator at A in Figure 4. 




























































































8 


BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 









T i 

18,750 KC 











y 

& 


\ 







/ 

y 










UJ — 

o i- 

§5 


;t 


• 20 

10 

0 

-10 

-20 

-30 

20 

10 

0 

-10 

-20 

-30 









i 

12,500 KC 











x v 


—X— 


-- 
















cr 


















c 

i i 

)375 KC 





"m —■ 






S 










/ 










C 












0 10 20 30 40 50 60 70 80 90 100 

DISTANCE BETWEEN ANTENNAS IN FEET 


Figure 6. Interaction effects between antennas; 
oscillator at B in Figure 4. 


wavelengths up to 24 meters but would be en¬ 
tirely unsatisfactory for wavelengths greater 
than 30 meters. Thus two complete antenna 


systems would be needed to cover the range 
from 16 to 64 meters (18.75 to 4.68 me). It 
might be possible to balance out these inter¬ 
action effects by using only the middle pair of 




< 




Figure 7. Interaction effects between antennas; 
oscillator at C in Figure 4. 


a line of 4 or 6 equal-spaced antennas with the 
unused antennas terminated in a dummy load 
of the same impedance as the load impedance 
of the used antennas. However no tests have 
been made of such a system. 







































































































































































AMPLITUDE-COMPARISON METHOD 


9 


Phase-Comparison Method Possibilities 

No attempt will be made to go into the de¬ 
tails of the equipment that would be needed for 
a direction finder operating by the phase-com¬ 
parison method. If only moderate accuracy is 
demanded a system could be built using several 
antennas arranged in a semicircle, each pair 
to be used for observing over only a limited 
range of azimuth. Once the correct antenna 
pair had been selected the taking of a bearing 
could be made practically instantaneous, but it 
might be necessary to try several different 
pairs before the correct one was selected and 
in that time the signal might be lost. It is con¬ 
ceivable that an instantaneous, direct-reading 
direction finder based on the phase-comparison 
principle could be devised, but the equipment 
would necessarily have to be very complicated 
and would require considerable time to develop. 
For these reasons attention was turned to sys¬ 
tems working by the amplitude-comparison 
method. 

14 AMPLITUDE-COMPARISON METHOD 

Direction finders which use the balanced loop 
for a collector system are perhaps the most 
commonly known and simplest form of direc¬ 
tion finder based upon the amplitude-compari¬ 
son method. When properly constructed, they 
work very well for waves which are entirely 
vertically polarized; however, if there is any 
horizontally polarized component to the wave, 
currents will be induced in the loop which will 
mask the normal ‘‘figure eight” directional 
characteristic and will prevent the taking of 
accurate hearings. Since all radio waves which 
have suffered reflection from the ionosphere are 
more or less randomly polarized, this suscepti¬ 
bility to horizontally polarized waves makes the 
loop antenna practically useless for long-range 
direction finding on the short wavelengths. 

Adcock Antenna 

The Adcock antenna was designed to over¬ 
come the effect of horizontally polarized waves. 
In its simplest form an Adcock direction finder 
consists of two spaced vertical doublets con¬ 
nected by a balanced transmission line with a 


receiver connected across the transmission line 
at the midpoint. The conductors of the line to 
one of the doublets are reversed with respect 
to those to the other doublet. Connected in this 
way an Adcock antenna is, in reality, a two- 
element vertical array with the outputs in 
phase opposition. When the spacing between 
the doublets is small with respect to the wave¬ 
length the simple Adcock antenna has the same 
“figure eight” directional characteristic for 
vertically polarized waves as the loop. Figure 
8A shows a schematic diagram of a simple 
Adcock antenna with its associated receiver 
and Figure 8B gives the horizontal directional 
characteristic. 



A B 

Figure 8. Diagram of simple Adcock direction 
finder. 

In free space and with perfectly balanced 
transmission lines such a system would be un¬ 
affected by horizontally polarized waves, but, 
actually, the lower halves of the doublets are 
always nearer the earth than the upper halves 
so that a perfectly balanced system can not be 
obtained except through the use of compensat¬ 
ing networks which require critical adjustment 
and must be retuned every time the receiver 
is tuned to a new wavelength. However, with¬ 
out these compensating networks, the un¬ 
balance is not serious if the antenna is elevated 
to a reasonable height above ground. Fairly 
accurate bearings can be obtained (1) if the 
rest of the equipment is kept small and simple, 
(2) if care is taken to keep the horizontal 

















10 


BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


members well balanced, and (3) if all the 
vertical elements other than the doublets are 
kept symmetrical. 

Bearings are taken by rotating the antenna 
about a vertical axis until one of the nulls of 
the directional characteristic is pointed towards 
the direction of arrival of the signal, at which 
time the output of the receiver will at at a 
minimum. The direction of arrival or apparent 
bearing of the station will then be perpendicu¬ 
lar to the line joining the antennas. The taking 
of bearings in this manner consumes an ap¬ 
preciable time, especially if the signal is weak 
or fading, when it is necessary to move the null 
of the directional characteristic back and forth 
across the signal direction several times before 
the bearing is certain. This time required for 
taking a bearing might mean that the signal 
would be lost before a bearing could be ob¬ 
tained. For this reason a quick-acting, direct- 
reading system would be desirable. 


Direct-Reading System with Crossed 
Adcock Antennas 


Figure 9A shows the schematic diagram of a 
direct-reading system using two mutually per¬ 
pendicular Adcock antennas. In Figure 8B the 
equation for the output of a single Adcock was 
given as 


where 


I = k sin 



sin 


“) 


cos 8 COS (ot 


(3) 


I is the output current; 

& is a proportionality factor; 

a is the angle between the azimuth of 
the direction of arrival and the per¬ 
pendicular to the line joining the two 
antennas; 

d is the spacing of the antennas; 

A is the wavelength of the incoming 
signal; 

w is 2?r times the frequency of the in¬ 
coming signal; 

8 is the vertical angle of arrival. 


If now we have an antenna system consist¬ 
ing of two crossed Adcock antennas, and if we 
let the axis of one run north and south and 
that of the other run east and west, then the 
equations for the current output of the two 
antennas are 


Ins — 

k sin j 

( 2nd \ 

—j— cos a 1 

| COS 8 COS (ot 

(4) 

Jew = 

k sin | 

( 2nd . \ 

—— sin a 1 

| cos 8 COS wt 

(5) 


where a is now measured clockwise from the 
north-south line to the azimuth of the direction 




oa=i NS = k SIN cos<x)cos<S COS CJ 

Ob =l EW , k SIN (— IA SIN a ) cos ,5 cos (J 


CATHODE RAY 
OSCILLOSCOPE 


A B 

Figure 9. Diagram of direct-reading crossed 
Adcock direction finder. 


of arrival and is, therefore, the apparent bear¬ 
ing of the station. If d is small with respect to 
A the equations become 0 



cos a cos 8 cos 


(4') 


2 ? T d 

I ew = k —— sin a cos 8 cos U. (5') 


If these two antenna outputs are fed through 
appropriate equal-gain high-frequency ampli¬ 
fiers with equal phase shifts to the vertical and 
horizontal deflecting plates respectively of a 


c When the spacing is not small with respect to a 
wavelength, i.e., when sin (Z'jrd/X) cos a is not approxi¬ 
mately equal to (2'7rd/A) cos a, an error is introduced 
which is zero for those bearings which are multiples of 
45° and which reaches a maximum at the eight inter¬ 
mediate directions. Accordingly it is called the “octan- 
tal” error. 





























AMPLITUDE-COMPARISON METHOD 


11 


cathode-ray oscilloscope, the spot on the oscil¬ 
loscope will trace a line which will make an 
angle cc with respect to the vertical and will 
therefore, except for the 180° uncertainty, give 


an antenna system is small the interaction be¬ 
comes large, but for the crossed Adcock an¬ 
tennas these effects are balanced out, providing 
the axes of the two pairs are exactly perpen- 



Figure 10. Block diagram of crossed buried U direction finder with injection signal. 


the bearing directly. If the phase shifts through 
the amplifiers are not equal, the spot on the 
oscilloscope will, in general, trace an ellipse 
instead of a line, the major axis of which will 
not give an accurate bearing. 

When the spacing between the elements of 


dicular and the antenna are all equispaced 
from the center. 

It is to be noted that the achievement of this 
instantaneous, direct reading feature has re¬ 
quired the complication of both the antenna 
system and of the receiving equipment, making 

















































































12 


BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


the problem of keeping the system balanced 
and symmetrical much more difficult. Some ex¬ 
perimenters have attempted to overcome these 
difficulties by housing the receiving and indi¬ 
cating equipment in a shielded coop located at 


not perfectly conducting, and if the horizontal 
members were buried to such a depth that they 
were unaffected by the incoming waves. The 
paragraphs below contain a description of an 
antenna system of this type which was built at 



Figure 11. Details of construction of Adcock antenna system with square diagonals of 15 ft arranged north- 
south and east-west. Each antenna is 1V^ ft in cross section, 28 ft high and is covered with copper-coated paper. 


the exact center of the antenna system and 
then elevating the whole structure above the 
ground on poles. Even for such a system, care 
must be taken to keep the vertical portions of 
the power leads symmetrical and the horizontal 
portions well buried in the ground. 

15 CROSSED BURIED U ANTENNA SYSTEM 

If we had perfectly conducting ground an¬ 
other way of overcoming these difficulties 
would be to use only the upper halves of the 
doublets, bringing them down to the ground 
level and shielding the horizontal leads by 
burying them in the ground. It would be ex¬ 
pected that such a system would work satis¬ 
factorily if the ground were uniform even if 


Holmdel and gives the results of various tests 
performed upon it. 

Antenna Transformers and Buried 
Conductors 

A schematic diagram of the antenna system 
and of the equipment used in testing it are 
shown in Figure 10. The four vertical antennas 
were located at the corners of a square with a 
diagonal spacing of 15 feet. The square was 
laid out so that one diagonal extended north 
and south and the other east and west. These 
four antennas consisted of box-like structures 
IV 2 ^et square in cross section, extending 28 
feet above the ground and covered with copper- 
coated paper. At 1 foot above the ground level 

















CROSSED BURIED U ANTENNA SYSTEM 


13 


the four sides of the antenna were brought to¬ 
gether at a point in the middle by an inverted 
pyramid of galvanized sheet iron while the four 
wooden corner posts were extended down 4 feet 
below the surface of the ground where they 
were bolted to a wooden framework. 

Details of construction of these antennas are 
shown in Figure 11. Great care was taken 
when they were erected to keep them located 



Figure 12. Concentric lines from antennas con¬ 
nected to broad-band transformer located in 
shielded box at center of antenna system. Housing 
is removed. 

exactly as planned. After completion, check 
measurements showed a maximum error in di¬ 
rection of only 27 minutes and in spacing of 
less than 14 inch. The half-wave resonant im¬ 
pedance of a single unit (i.e., between the an¬ 
tenna and ground) was about 250 ohms and 
occurred at a wavelength of about 22.2 meters. 
The quarter-wave impedance was about 36 
ohms and occurred at about 44.4 meters. Over 
the frequency range of 5 to 15 me the output 
of a single pair for an end-on signal was about 
9 db below that for a horizontal half-wave 
doublet at a height of 60 feet. 

The inner conductor of a 72-ohm concentric 
transmission line was connected directly to the 


apex of each inverted pyramid, all four lines 
being of equal length. These transmission lines 
ran straight down to 4% feet below the surface 
of the ground and then horizontally to the cen¬ 
ter of the system where they were brought 
back up to the ground level. Here they were 
connected to the primary, or balanced, side of 
a balanced-to-unbalanced broad-band trans¬ 
former, the lines from each diagonal pair being 
connected to the same transformer. These 
transformers were housed in the small shielded 
box at the center of the antenna system which 
is visible in Figure 11. Figure 12 is a view of 
the interior with the cover removed. 

Two more concentric lines, one from the 
secondary of each transformer, ran back down 
to 4i/2 feet below the ground, then horizontally 
for about 100 feet. Here they commenced a 
gradual rise to a depth of about 1 foot at which 
depth they remained until they reached the 
apparatus building at a distance of about 700 
feet from the antennas where they were con¬ 
nected to the inputs of two receivers. 


-I12 k 



Figure 13. Diagram showing one pair of crossed 
buried U antennas. 


Figure 13 shows a diagonal cross section of 
the antenna system and illustrates the disposi¬ 
tion of the transmission lines. They were 
buried in this manner to provide sufficient 
shielding to eliminate all pickup from the hori¬ 
zontally polarized waves. 








































14 


BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


Figure 14 shows the details of construction 
of the broad-band balanced-to-unbalanced 
transformers, and Figure 15 the frequency 
characteristics. Details of the measuring tech¬ 
nique will be found in the project final report. 1 
A balance of only 20 db would give an error 
in bearing of 5.7°. For a balance of 30 db the 
error is 1.8° and for 40 db the error is 0.6°. 



UNBALANCED WINDING 
TWO WINDINGS EACH 6 TURNS 
NO. 36 DSC WOUND AS SHOWN 
BELOW 



BALANCED WINDING 
ONE CENTER TAPPED WINDING 10 
TURNS NO. 32 DSC WOUND AS 
SHOWN BELOW 




- C, = 140 /U/Uf 
C 2 = 51.7 >UyU f 

Figure 14. Construction details of balanced-to- 
unbalanced transformer. 


Receiving Arrangements— 
Injection-Signal System 

Figure 10 is a block diagram of the d-f sys¬ 
tem. In operation, the incoming signal beats 
with the signal from the injection oscillator to 
give a beat note somewhere between 100 and 
2,000 cycles. The injection-oscillator input level 
to the two receivers is equal at all times while 
the incoming signal input level varies in ac¬ 
cordance with the directional pattern of the 
crossed Adcock antenna system. The receiving 
equipment measures the incoming signal direc¬ 
tion by comparing the audio-frequency levels 
from the two receivers. 


The audio output of the receiver connected 
to the north-south antenna pair is connected to 
the vertical deflecting plates of a cathode-ray 
oscilloscope and the output of the receiver con¬ 
nected to the east-west antenna pair is con¬ 
nected to the horizontal deflecting plates. Now 
it will be seen that if the overall gains of these 
two receivers are equal, the ratio of the audio 
outputs of the two receivers will be equal to the 
ratio of the carrier output of the two antenna 
pairs, which in turn is a function of the signal 
direction as shown above. When these voltages 





4 5 7 10 15 20 30 

FREQUENCY IN MEGACYCLES/SECOND 


Figure 15. Characteristics of broad-band trans¬ 
former. 


are applied to the oscilloscope a line is formed 
which gives the apparent bearing of the sta¬ 
tion directly. 

Gain Control by Injection Signal. The gains 
of the two receivers are kept approximately 
equal in the following manner. An injection 
signal is radiated from a fifth antenna located 



































































































TESTS ON COMPLETE SYSTEM 


15 


at a distance of 500 feet from the crossed an¬ 
tenna system and on a line which bisects one 
of the 90° angles formed by them. This signal, 
which is adjusted to differ in frequency by a 
few hundred cycles from the carrier frequency 
of the signal whose direction is being meas¬ 
ured, is used to control the gains of the two 
receivers. Since the outputs of both antenna 
pairs are equal for the injection signal the 
gains of the two receivers will be made equal 
providing the injection signal is much stronger 
than the carrier and providing the gain con¬ 
trols track. A ratio of injection signal to car¬ 
rier signal of 26 db should be enough to insure 
that the carrier will have no effect on the gain 
controls and is enough to insure that the audio 
output produced in the linear low-frequency 
detectors by the beating of the carrier with 
the injection signal is directly proportional to 
the carrier output of the antenna system. Ac¬ 
cordingly, the injection signal was kept about 
26 db above the carrier level. The gain controls 
of the two receivers used for making prelimi¬ 
nary measurements did not track within the 
required limits so that it was necessary to cali¬ 
brate the system frequently. This was accom¬ 
plished by modulating the injection oscillator 
with an audio frequency and adjusting the 
gains of the two receivers until the line on the 
cathode-ray oscilloscope was in the 45° position 
which is the condition for equal gains. This 
fault was eliminated in the final system de¬ 
scribed below. 

Besides furnishing a nonfading equal-ampli¬ 
tude signal for controlling the gain of the re¬ 
ceivers, the use of the injection oscillator 
greatly reduces the phase-shift requirements. 
Instead of equal phase shifts from antenna to 
oscilloscope, all that is required of the receivers 
when the injection oscillator is used, is that 
the phase shift from the antenna to the input 
of the audio amplifier varies in the same man¬ 
ner for the two receivers over the band be¬ 
tween the carrier frequency and the injection 
oscillator frequency. The two audio amplifiers 
must have the same phase shift for the audio 
frequency used, but this requirement is not 
difficult to meet. 


TESTS ON COMPLETE SYSTEM 

First tests on the antenna system showed 
very shallow nulls and large errors in direc¬ 
tion. For details of these measurements see the 
final report. 1 

When the possible causes for these errors 
were considered suspicion was first cast upon 
the vertical leads running to nearby antennas. 
Although the nearest of these antennas was 
over 200 feet distant, their removal and re¬ 
moval of their vertical leads made differences 
as great as 11 db in the depth and 3° in the 
direction of the nulls. Vertical wires at dis¬ 
tances of 500 feet and over had no significant 
effect in the frequency range studied. 

A set of measurements was made with all 
possible reradiating objects within a radius of 
500 feet removed. Null depths of only 36 db 
and directional errors of 12° were still being 
obtained. 

Effect of Ground 

The ground at Holmdel is ordinary farm 
land consisting of a layer of top soil 1 foot 
thick overlying several feet of sandy clay. 2 The 
particular site chosen for the antenna system 
was as fiat as reasonably could be expected of 
most antenna locations and, as far as could be 
discovered by visual examination, there was no 
reason for suspecting any troublesome varia¬ 
tions in the ground constants. However, sev¬ 
eral different ground mats were tried. The first 
consisted merely of two 100-foot galvanized 
iron wires, one stretched under each pair of 
antennas and grounded at each end to 5-foot 
ground rods and at the center to the outer 
grounded conductor of the coaxial transmission 
lines. This ground system made no appreciable 
effect on either the depth or direction of the 
nulls. 

Next a ground mat consisting of two 50-foot 
strips of 2-inch mesh wire netting 6 feet wide 
was laid in the form of a cross under the an¬ 
tennas. The maximum error in direction was 
reduced from 12° to 3.6°, although the mini¬ 
mum null depth was not changed much. Further 
improvement was desired, so a ground mat con¬ 
sisting of 8 radial strips of wire netting 6 feet 
wide and 150 feet long grounded at 50-foot 
intervals was tried. The results were not ap- 





16 


BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


preciably different so still another mat was 
tried. This final one was a circular mat 100 feet 
in diameter grounded at 6-foot intervals around 
the circumference and at the center. 

Except for the 18-mc measurements, the 
maximum null depth was over 30 db and the 
maximum error in direction was less than 2°. 
Since the antennas and transformers were not 
designed to work at frequencies higher than 15 
me it was felt that further improvement of the 
ground mat was not necessary. 

With the 100-foot diameter ground mat in 
place, test bearings were taken on the field 
oscillator placed at 16 equally spaced points on 
a circle of 300 feet radius. When the errors 
were corrected for octantal error and for those 
due to the cathode-ray tube, over the frequency 
range 5 to 15 me no error greater than 2° was 
observed, while at 18 me the maximum error 
was 3°. 

Tests on Horizontally Polarized Waves 

To test the response of the system to hori¬ 
zontally polarized waves, a 53-foot tower was 
erected at a distance of 200 feet in the direc¬ 
tion of the east null. The field oscillator was 
placed on top of the tower and the change in 
null depth noted when the antenna rods were 
turned from the vertical position to an angle 
of 45°. Measurements were taken on 5, 7.5, 10, 
15, and 18 me. No significant change in null 
depth was detected indicating that the amount 
of horizontal pickup was too small to affect 
the operation of the system under normal op¬ 
erating conditions. 

Finally, bearing measurements were made 
on fixed transmitting stations ranging in fre¬ 
quency from 5 to 18.4 me and in distance from 
about 30 miles to over 5,000 miles. Each bear¬ 
ing was checked by comparing it with that 
obtained on the same station with the crossed 
vertical Musa. A total of 107 bearings were 
taken. For 25 of these the vertical Musa gave 
either no bearing indication whatsoever or 
bearings differing significantly from the true 
bearing, indicating that either the station was 
inside the skip zone or that the transmission 
was unsatisfactory for direction-finder opera¬ 
tion. Of the 82 remaining bearings, 29 differed 
from the true bearing by 0.5° or less, 24 dif¬ 


fered by 0.6° to 1.0°, 12 by 1.1° to 1.5°, and 9 
by 1.6° to 2.0° or a total of 74 of the 82 bearings 
were in error by 2° or less. For the remaining 
8 stations the largest error was 6°. Carefully 
made repeat measurements on these 8 stations 
gave no error greater than 3°. 

17 CONCLUSIONS 

The errors in such a phase-comparison sys¬ 
tem, employing several vertical antennas ar¬ 
ranged in a semicircle, are small unless the 
vertical angle of arrival is unusually high. Be¬ 
cause of interaction effects, two complete sets 
of antennas are needed for the range from 5 
to 18 me if maxmium accuracy is required. 
With a system using three antennas, prefer¬ 
ably arranged at three corners of a square, two 
simultaneous phase measurements are required 
to obtain a bearing. Interaction among the an¬ 
tennas influences the accuracy, although the 
interaction effects might be balanced out by the 
use of several dummy antennas. 

Difficulties involved in making an instantan¬ 
eous, direct-reading, p/mse-comparison system 
led to the development of an amplitude- com¬ 
parison system using a crossed, buried U an¬ 
tenna with a separate injection oscillator and 
antenna and with a cathode-ray indicating de¬ 
vice. 

Variations in the ground not detected by the 
eye cause severe distortion of the directional 
pattern of the antenna. Several ground-mat 
systems were investigated. Conservative speci¬ 
fications indicate that a mat not less than 150 
feet in diameter made of 1-inch mesh wire net¬ 
ting or its equivalent would be required. Where 
the ground has a uniformly high conductivity 
such as would be found in a salt marsh, the 
mat could be smaller. Even in a location having 
good ground conductivity, a good ground mat 
seems desirable; For details of a system suit¬ 
able for a salt-marsh location see the final 
report. 1 

18 RECEIVER SPECIFICATIONS 

The general receiver characteristics desired 
are (1) a frequency range of 4.5 to 30 me, (2) 
an input impedance of 72 ohms, and (3) an 
image rejection ratio of better than 50 db at 



COMPLETE D-F SYSTEM 


17 


20 me. An i-f band flat over a ±2-kc range 
and down 45 db ±10 kc would be satisfactory. 
To operate ordinary commercial oscilloscopes 
an a-f output of 2.5 volts across 100,000 ohms 
at 5 per cent modulation is required. 

The lowest signal level that can be received 
is determined by the equivalent input noise 
(output noise divided by the receiver gain) 
which should not be more than 5 db above the 
theoretical thermal noise or 160 db below 1 
watt for a 4,000-cycle band width. Assuming a 
minimum signal level 20 db above the noise 
gives a minimum signal input level of 140 db 
below 1 watt. 

The ratio of the a-f output levels must be the 
same as the ratio of the incoming signal levels 
delivered by the two antenna pairs. Therefore, 
if linear low-frequency detectors are used the 
level from the injection oscillator must be at 
least 20 db above the incoming signal level. 
The effect of the incoming signal on the auto¬ 
matic-gain-control circuits may require a 30-db 
difference in level. If we assume a 26-db differ¬ 
ence in level, the modulation impressed on the 
injection oscillator signal by the incoming sig¬ 
nal will never exceed 5 per cent. If incoming 
signal levels of 140 to 80 db below 1 watt are to 
be accommodated the injection oscillator level 
must be between 114 and 54 db below 1 watt 
and the receivers must be capable of handling 
these levels. The input level range may be in¬ 
creased to 80 db by inserting 20-db attenuation 
in the antenna lines to take care of exception¬ 
ally strong signals. 

For successful operation, the gains of the 
two receivers must be kept equal at all times. 
This feature practically demands the use of 
independent and extremely “stiff” automatic 
gain controls whose action is completely con¬ 
trolled by the injection oscillator. These stiff 
gain controls may take the form of separate 
i-f gain-control amplifiers and strongly biased 
rectifiers. 

For a bearing accuracy of %°, the gains of 
the two receivers must not differ by more than 
0.1 db, and this must hold over the entire range 
of input levels. To eliminate any audio fre¬ 
quency gain controls so that the amplitude of 
the oscilloscope trace may be used as an indi¬ 
cation of the percentage modulation (i.e., ratio 
of incoming signal to injection oscillator level) 


the automatic gain controls should keep the 
output constant to within 2 db for a 60 db 
change in input level. 

Three other factors which will affect the 
bearing accuracy are (1) nonlinearity of the 
a-f circuits, (2) dissimilarities in the r-f, i-f, 
and a-f bands, and (3) crosstalk between the 
two receivers. To maintain an accuracy of Vs° 
the a-f circuits should not depart from linearity 
by more than 0.1 db as the amplitude is varied, 
and the gains should be alike to within 0.1 db as 
the frequency difference between the injection 
oscillator and the signal is varied over a ± 200- 
cycle to 2-kc range. The crosstalk from one of 
the receivers, fed with a 5 per cent modulated 
signal, into the other with an equal unmodu¬ 
lated signal, should be down 50 db. 

Factors which will distort the oscilloscope 
trace are 60-cycle hum, harmonic distortion, 
and phase shift. The hum-and-harmonic dis¬ 
tortion should be down 35 db in the output in 
order that the oscilloscope trace will not be 
widened by more than 2 per cent of its length. 
The phase difference between the two a-f out¬ 
puts should not exceed 1 per cent. 

If necessary, high-pass filters may be used 
in the a-f circuits to reduce the 60-cycle hum, 
in which case the a-f range might be 200 to 
2,000 cycles. Filters cutting off above 2,000 
cycles might be used to reduce noise. Any such 
filters must, of course, meet the phase and 
amplitude-distortion requirements within the 
pass band. 

The receivers, preferably, should have a fre¬ 
quency-calibrated dial and a tuning indicator 
to insure that the operating requirements will 
be met. Separate beating oscillators may be 
used, provided the leakage from one beating 
oscillator into the other receiver does not result 
in a-f components stronger than 50 db below 
the output of the desired frequency. 

1-9 COMPLETE D-F SYSTEM 

No commercial receivers were on the market 
which exactly met the specifications listed 
above, and previous experience with commer¬ 
cial short-wave receivers showed that their 
modification to meet these specifications would 
not be easy. Since, however, the construction 
of two entirely new receivers would have taken 






18 


BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


a relatively long time, it was decided to use, 
after modification, two short-wave measuring 
sets belonging to the Bell Telephone Labora¬ 
tories. 

Receivers Used and Changes Required 

A block diagram of one of the two receivers 
before alteration is shown in Figure 16. They 
are triple-detection receivers with a first inter¬ 
mediate frequency of 3 me and a second inter¬ 
mediate frequency of 100 kc. There are no r-f 


were capable of supplying an audio output well 
over 2.5 volts across 100,000 ohms, but the 
overall gain of the receivers was not sufficient 
to supply this output for the weakest usable 
signal which is determined by the minimum 
noise level of the receiver. By adding another 
stage to the 100-kc amplifier the overall gain 
was increased to the point where the required 
audio output was obtained for the weakest 
usable signal. 

The equivalent input noise was about 5 db 
above the theoretical thermal noise level. 



Figure 16. Block diagram of triple-detection measuring set. 


amplifiers or preselector circuits other than the 
antenna-tuning and coupling circuits. The re¬ 
ceivers cover a frequency range from 4.5 me to 
well over 30 me. The image rejection ratio is 
about 40 db which is high enough for demon¬ 
stration purposes. 

The overall pass band of the receivers was 
flat for approximately 5 kc on each side of the 
carrier, which was entirely too wide for d-f 
use. By adjusting the second i-f filters the band 
was reduced so that it was flat for only 1 kc on 
each side of the carrier and was down 45 db 
or more 10 kc on each side of the carrier. It 
should be noted here that the response curves 
of the two receivers must be exactly similar 
over the operating range. This range is deter¬ 
mined by the difference between the signal fre¬ 
quency and the injection-oscillator frequency. 
If the band width is small and the response 
curves sharply peaked it becomes extremely 
difficult to make and keep them identical. 

The linear rectifiers used as final detectors 


It was necessary to install separate, biased, 
gain-control rectifiers to obtain the desired 
automatic-gain-control stiffness. The result was 
an output variation of only 1.6 db for a 60-db 
change in input level. The gain controls of the 
two receivers tracked to better than 0.1 db over 
the range of inputs from 65 db above to 25 db 
above 10" 12 watt. At lower input levels the 
noise prevents precise measurement of the 
tracking, but no appreciable difference was dis¬ 
cernible. 

The hum level of the receivers was fairly 
high but was reduced to a satisfactory value by 
minor changes in the power supply leads. The 
nonlinearity in the audio circuits, the har¬ 
monic distortion, and the difference in phase 
shift were all less than the required minimum. 

After the above changes were made the two 
receivers wore connected together and to the 
antenna system making the complete d-f sys¬ 
tem shown in the block diagram in Figure 17. 
The first beating oscillator from one of the re- 


























COMPLETE D-F SYSTEM 


19 


ceivers was used as the common beating oscil¬ 
lator for both receivers. The first beating os¬ 
cillator for the other receiver was removed and 
mounted on a separate panel with a broad-band 
amplifier and used as the injection oscillator. 
The injection-signal level was controlled by 


cycle tone and adjusting the relative audio 
gains of the two receivers until the oscilloscope 
trace made a line at 45°. The 60-cycle modula¬ 
tion was produced by putting a small 60-cycle 
voltage on the grid of the amplifier tube in 
series with the grid bias. 


FROM 
ANTENNA 
PAIR "A* 


R-F 


1 ST 


3 MC 

CIRCUITS 


DET 


FILTER 



2 ND 
DET 


3.1 MC 
CRYSTAL 
OSC 


PAD 


3.1 MC 
CRYSTAL 
OSC 



~~r~ 



R-F 


1 ST 


3 MC 

CIRCUITS 


DET 


FILTER 


2 ND 
DET 


FROM 
ANTENNA 
PAIR "B" 


IIO V 
60~ 



IOO KC 
FILTER 


100 KC 
AMP 


7 


TO INJECTION 
ANTENNA 


MOD 8 
H-F AMP 

T 

MGC 


INJECTION 

OSC 


Figure 17. Block diagram of complete d-f system using triple-detection measuring set. 


varying the grid bias of the amplifier tube. By 
making the pass band of the amplifier very 
broad the necessity of providing tuning for the 
amplifier circuits and of meeting the resulting 
tracking requirements was eliminated. 

Calibration of the system was accomplished 
by modulating the injection signal with a 60- 


Test measurements made on the complete 
setup disclosed a small amount of 3-mc cross¬ 
talk from one receiver to the other through the 
common beating oscillator lead and a small 
amount of high-frequency crosstalk direct from 
the injection oscillator to the input circuits of 
the receivers. The 3-mc crosstalk was reduced 





























































































































BTL HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


20 


below the required minimum by inserting small 
pads in the leads to each receiver and increas¬ 
ing the beating oscillator power by a corre¬ 
sponding amount. The high-frequency cross¬ 
talk was reduced by very carefully shielding 
the injection oscillator. 

The experience gained in working with these 
receivers showed that the greatest difficulties 
in building receivers for this type of d-f system 
are likely to be in making the two pass bands 
identical over the operating range, in keeping 
the crosstalk through the common beating oscil¬ 
lator lead at a low value, and in keeping the 
leakage from the injection oscillator direct to 
the receiver below the required minimum. By 
taking normal precautions all requirements 
were met, demonstrating that the system is 
entirely practical. 


110 TECHNICAL AID FOR THE 
ARMY SIGNAL CORPS 

After a thorough testing, the two measuring 
sets and the necessary cathode-ray equipment 
were set up in a small building a short distance 
from the crossed buried U antenna system and 
connected to it by coaxial transmission lines 
making a complete d-f system. Engineers from 
the Signal Corps Laboratories then operated 
and studied this equipment to familiarize them¬ 
selves with the principles involved. 

Receivers Used and Changes Required 

After operating this equipment for a while 
and after considering the possible sources of 
supply and the urgency of the need the Signal 
Corps engineers decided to attempt to rebuild 
two National N. C.-100 receivers for their first 
system. At first it was hoped that this rebuild¬ 
ing would involve merely the addition of a 
separate automatic-gain-control amplifier and 
rectifier and of a lead between the two beating 
oscillators to keep them in synchronism. How¬ 
ever, before the equipment was finally made to 
function satisfactorily, it was found that 
rather extensive changes had to be made. In 
the final arrangement three receivers were 
used, one for each of the receiving channels 


and the third to supply the beating oscillator 
and injection signals. 

Test Results. The final tests on these re¬ 
ceivers showed that they functioned very satis¬ 
factorily except for two features. The image- 
rejection ratio was better than 40 db for 
frequencies between 5 and 11 me, but above 11 
me the rejection ratio dropped very fast until 
at 14 me it was only 28 db and even less at 15 
me. The equivalent input noise of these re¬ 
ceivers was considerably higher than the speci¬ 
fied minimum, especially on the higher fre¬ 
quencies, making it difficult to obtain accurate 
bearings on weak stations. It is suspected that 
the difficulty lies in the comparatively low gain 
of the r-f amplifier and in the low Q of the h-f 
coils. 

It is not believed that low image-rejection 
ratio at the high frequencies is so serious as 
to rule out the use of such receivers, but, since 
it is the weak signals that are the important 
ones, the low signal-to-noise ratio is very seri¬ 
ous. 

111 EXTENDING THE RANGE TO 30 MC 

It was felt that, as a result of the experience 
gained in building and testing the crossed 
buried U antenna system for the 5- to 15-mc 
range, enough was known about the character¬ 
istics of such an antenna system to predict the 
performance of a smaller model with sufficient 
accuracy to make unnecessary the building of 
an experimental model. The antenna system 
for the 15- to 30-mc range would be merely a 
half-size scale model of the present system. 
Thus, the ground mat should be 75 feet in di¬ 
ameter, the antenna spacing should be 7% feet, 
and the size of each individual vertical should 
be 9 inches by 9 inches by 14 feet. 

The Transformers 

The design for the broad-band balanced-to- 
unbalanced antenna-coupling transformers fol¬ 
lowed very closely that of the ones for the 
lower frequency range. For details of construc¬ 
tion of the transformers and loss, balance, 
and impedance characteristics see the final re¬ 
port. 1 




POLARIZATION ERRORS-SYSTEM MEASUREMENTS 


21 


112 POLARIZATION ERRORS— 

SYSTEM MEASUREMENTS 

D. G. C. Luck and Kenneth A. Norton of the 
RCA Manufacturing Company brought their 
balloon equipment to Holmdel and made meas¬ 


urements of the response of the crossed buried 
U-antenna system to vertically and horizon¬ 
tally polarized waves for various vertical angles 
of arrival. The standard wave error 3 was found 
to be 8.5° at 5.1 me, 6.25° at 9.23 me, 2.2° at 
13 me, and 1.8° at 17.3 me. 




Chapter 2 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


Theoretical and experimental investigation of direc¬ 
tion-finder characteristics, particularly polarization er¬ 
rors; development of a figure of merit for direction¬ 
finder comparison; examination of typical direction¬ 
finder systems as an application of the methods for 
measurement and analysis developed; origin of a new 
method for measuring ground constants. The major 
portion of the theoretical analysis developed in this 
project 21 is included in this summary report, the chief 
abridgement from the contractor’s final report 1 being 
in the work conducted on direction-finder systems of 
the time 1941-1942. 

INTRODUCTION 

he objectives in setting up this project 
were: 

1. Study of errors due to polarization, col¬ 
lector spacing, and diversity factor, and meth¬ 
ods to minimize these. 

2. Study of errors of site and personnel. 

3. Examination of improved models from 
any source. 

4. Basic research on one or more improved 
types as appears desirable. 

5. Measurement technique for the study of 
d-f errors. 

After some preliminary study it was found 
that polarization and site errors constituted the 
largest errors in existing direction finders. The 
program therefore was chiefly devoted to a 
study of those errors over a frequency range 
of 2 to 80 me. 

22 ANALYSIS 

For the study of polarization errors a method 
was developed having advantages over previ¬ 
ously used methods and applicable to many d-f 
antenna systems. In this method a figure of 
merit designated as the “pickup ratio” was 
introduced. The pickup ratio is the ratio of the 
pickup factor, h, of the d-f antenna system for 
desired radiation field components to its pickup 


a Project C-18: The work covered in this report was 
performed by the National Bureau of Standards under 
a contract terminating June 30, 1942. 


factor, k, for undesired field components. A 
knowledge of the pickup ratio together with the 
directional pattern of response of the d-f sys¬ 
tem makes possible the determination of the 
polarization errors for downcoming sky waves. 
Since it is possible to measure the pickup ratio 
for a wave at horizontal incidence, all measure¬ 
ments may be made near the ground. This is a 
principal advantage of the method; other ad¬ 
vantages are that the method yields the maxi¬ 
mum polarization error, and a figure of merit 
for polarization error which is independent of 
the ground constants and of the height of the 
direction finder above the ground. 

After developing the technique of determin¬ 
ing polarization errors through measurements 
of pickup ratios, measurements were carried 
out on several direction finders of various 
types. Reports issued during the project are 
listed in the Bibliography. 

Polarization errors were investigated com¬ 
prehensively, both theoretically and experi¬ 
mentally. The polarization of the field at a d-f 
site for downcoming ionospheric waves was 
determined theoretically. The d-f directional 
pattern was then calculated in such a field and 
equations obtained for the observed bearing. 
The difference between the observed bearing 
given by the actual directional pattern and the 
true bearing obtained from the ideal direc¬ 
tional pattern gave the polarization errors. 
Equations were derived for the polarization 
errors in this way for the several basic direc¬ 
tion finders and were used to determine, by 
means of experimental measurements of the 
constants h and k, the polarization errors of 
these direction finders. 

2 21 Summary of Theoretical Aspects 

Study of the state of polarization of down¬ 
coming ionospheric waves showed that these 
waves were elliptically polarized, having elec¬ 
tric components E p and E n polarized parallel 
and perpendicular respectively to the plane of 



22 



ANALYSIS 


23 


incidence. These components are present in¬ 
dependently of the state of polarization of the 
wave incident on the ionosphere and therefore 
of the polarization of the transmitting antenna. 
The wave incident on the ionosphere is split into 
ordinary and extraordinary waves which on 
returning to the earth combine vectorially to 
give the total downcoming wave. Equations for 
the fading of these ordinary and extraordinary 
wave components were derived and expressions 
found for the variations in the state of polari¬ 
zation of the downcoming wave. In general, the 
state of polarization varies in a random way 
so that the average of a series of swinging 
bearings will usually give a bearing close to 
the true bearing, provided the swinging is 
caused by polarization error. 

The total field at the direction finder for 
downcoming waves was next calculated by tak¬ 
ing the vector sum of the direct and ground- 
reflected waves. It was shown that the ground 
reflection acts to suppress E n at points near the 
ground, the suppression increasing as the index 
of refraction of the ground increases. This 
showed that direction finders designed to re¬ 
spond to E p and to suppress response to E n 
should be placed as near the ground as possi¬ 
ble. On the other hand, it is shown that a direc¬ 
tion finder designed to respond to E n and to 
suppress response to E p should be located at a 
height above ground equal to A/4. 

The response of an arbitrary direction finder 
in the field of a downcoming wave was next 
calculated in terms of the known directional 
patterns of the antenna elements of the direc¬ 
tion finders. In these expressions unknown pro¬ 
portionality constants, h and k, occur. These 
constants, which correspond to output voltages 
produced by the desired and undesired field 
components respectively, were called pickup 
factors, and the ratio of h to k, the pickup 
ratio. It was shown that, by placing the direc¬ 
tion finder in plane-wave fields of special struc¬ 
ture, all terms in the expressions for the output 
voltages became zero except one. A measure¬ 
ment of the output voltage and the field inten¬ 
sity for this case provided a means for deter¬ 
mining the pickup factor. The pickup factor 
for each field component desired could be meas¬ 
ured in this way by using special fields at hori¬ 
zontal incidence. After determining the pickup 


factors experimentally, the d-f response in the 
field of any downcoming wave could be calcu¬ 
lated and therefore also the polarization errors, 
since this calculation gave the actual azimuthal 
directional pattern. The departure of this di¬ 
rectional pattern from the ideal- desired pattern 
gave the bearing error. 

It was shown that the polarization errors 
were dependent on the ratio of desired to un¬ 
desired responses and therefore on the pickup 
ratio h/k. The pickup ratio was therefore pro¬ 
posed as a figure of merit for measuring polari¬ 
zation errors. This figure of merit is indepen¬ 
dent of the ground constants of the d-f site and 
of the height of the direction finder above the 
ground. It can be used to determine the results 
of development work designed to reduce the 
polarization errors of a given direction finder 
while the complete curve of polarization error 
versus angle of elevation of the incident wave, 
as determined by measurements of h/k, can be 
used to compare the accuracy of different types 
of direction finders. 

In applying the pickup ratio method in prac¬ 
tice, the required special test fields must be 
generated by means of local transmitters. Such 
transmitters generate waves which only ap¬ 
proximately simulate the plane waves assumed 
in the theory used to calculate the polarization 
errors. Accordingly, theoretical and experi¬ 
mental studies were made of the techniques 
required for a proper measurement of h and k 
where a local transmitter is used. It was shown 
that a horizontal loop antenna should be used 
with the local transmitter when generating a 
horizontally polarized test field to avoid an 
error called radiator parallax. Similarly, when 
testing a spaced, vertical, coaxial loop-antenna 
direction finder, special procedures were nec¬ 
essary to avoid an error called collector paral¬ 
lax. Equations were also derived to show the 
proper procedure required when measuring 
electric field components by means of a field 
intensity meter using a loop antenna. 

Using the procedures outlined above, meas¬ 
urements of h and k were made and the polari¬ 
zation errors computed for the direction finders 
under consideration. 

A study of d-f sites was made in which it 
was shown that direction finders designed to 
respond to E n should have smaller site errors 



24 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


caused by reradiation than those designed to 
respond to E p . Equations were derived for the 
field intensity at any given depth below the 
ground for incident downcoming waves and an 
approximate table prepared showing the rec¬ 
ommended depth to which cables and lines 
should be buried in order to avoid reradiation 
difficulties. A new method was evolved for 
rapidly measuring the ground constants of a 
proposed site, at various points of the site, both 
to determine its electrical homogeneity and to 
make sure that its conductivity and dielectric 
constant would be high enough for the best 
results with the direction finder to be installed. 
This method should be useful in selecting the 
best site when a choice is possible. 

2 ' 2,2 Historical Development 

The single loop-antenna direction finder has 
large bearing errors when used on downcoming 
waves from the ionosphere. Since long-distance 
communication makes use of propagation via 
the ionosphere, the single loop-antenna direc¬ 
tion finder could not be successfully used for 
short-wave direction finding in the band from 
2 to 30 me. In 1919, Adcock introduced the 
spaced-antenna direction finder which at¬ 
tempted to reduce pickup in the horizontal 
members of the antenna structure and thus the 
polarization errors. To measure the success of 
such attempts, R. H. Barfield 2 in 1935 intro¬ 
duced a figure of merit for polarization error 
which he called the ‘‘standard wave error.” 
This was defined as the bearing error of the 
direction finder for an incident wave having 
an angle of elevation of 45° and components 
of electric field intensity parallel and perpen¬ 
dicular to the plane of incidence equal to each 
other and of such phase difference as to make 
the error a maximum. The standard wave error 
was commonly measured by using a local trans¬ 
mitter elevated at 45° to lay down a field 
simulating the standard wave. A dipole trans¬ 
mitting antenna oriented at 45° generated 
equal and cophased parallel and perpendicular 
wave components in this method. The cophased 
wave components of the above method did not 
result in the measurement of the maximum 
polarization error as required in the definition 
of standard wave error. 3 Furthermore, the 


difficulties introduced in the use of elevated 
transmitters led to lack of precision in meas¬ 
urement. A method was needed which would be 
easier to use in practice; for example, one 
which required ground measurements only. 
Such a method was developed using a local 
transmitter near the ground to generate a wave 
field of measured intensity at nearly horizontal 
incidence. The response of the direction finder 
was measured in this field for vertical and for 
horizontal polarization. The response of the 
antenna system to sky waves was then deter¬ 
mined from a calculation of the vertical and 
horizontal field components of such sky waves, 
and this in turn gave the polarization errors, 
including the standard wave error. This method 
yielded the maximum polarization error, while 
the ratio of the responses or pickup factors, 
called the pickup ratio, yielded *a fundamental 
constant of the antenna system which was in¬ 
dependent of the ground constants and from 
which the response in any assumed sky-wave 
field could be calculated. About the same time 
as the development of this method, the Radio 
Corporation of America [RCA] 4 modified the 
Barfield method by using an elevated trans¬ 
mitter emitting waves polarized, first, in the 
plane of incidence and, second, perpendicular 
to the plane of incidence. The response of the 
direction finder to these waves was measured, 
but a measurement of field intensity was not 
required to determine the polarization error. 
This method usually could be made to give the 
same data as the National Bureau of Standards 
[NBS] method and vice versa, each method 
having particular advantages. 

22,3 Nature of Polarization Errors 

In short-wave direction finding, bearings are 
taken on sky waves coming down from the 
ionosphere. In general these waves are ellip- 
tically polarized, having components polarized 
both parallel and perpendicular to the plane of 
incidence. However, most direction finders are 
designed to measure the bearing by utilizing 
the directional pattern of the antenna-system 
response to only one of the components. If both 
components are present and if the antenna 
system has different azimuthal responses to the 
several field components, the directional pat- 




ANALYSIS 


25 


tern will be modified so as to give incorrect 
bearings or even to prevent the taking of bear¬ 
ings altogether. These bearing errors will de¬ 
pend on the relative amount and phase of de¬ 
sired to undesired field intensity which in turn 
will depend on the state of polarization of the 
incident wave. Such bearing errors are called 
polarization errors and are one of the largest 
sources of inaccuracy in present-day direction 
finders. 

Polarization of Downcoming 
Ionospheric Radio Waves 

As a background for the study of polariza¬ 
tion errors and the techniques used to measure 
such errors, it will be desirable to consider the 
nature of the downcoming sky waves, the in¬ 
fluence of the ground reflection on the fields 
set up by these waves at the direction finder, 
and finally the problems involved in attempting 
to simulate such sky waves by the use of local 
transmitters. A more complete account than 
can be given here is available in a report 5 by 
K. A. Norton. 

The following account is intended to serve 
only as a brief review of the way in which the 
problem is set up. 

The presence of the magnetic field of the 
earth causes a wave incident on the ionosphere 
to split into ordinary and extraordinary waves, 
each of which thereafter travels independently 
in the ionosphere and is reflected at different 
heights. To calculate the intensities of the com¬ 
ponents parallel and perpendicular to the plane 
of incidence in a downcoming wave, it is nec¬ 
essary to calculate the intensities of these com¬ 
ponents for the ordinary and extraordinary 
waves and then to take the vector sum of these 
two waves to give the total field components. 

The following symbols will be used together 
with Heaviside-Lorentz units; bold face sym¬ 
bols are vectors. A dot over a symbol denotes 
time differentiation. 

e = charge on an electron. 
m — mass of an electron. 

N = electron density. 

H° = earth’s magnetic field. 
v = mean frequency of collisions between 
free electrons and neutral air mole¬ 
cules. 


c = velocity of light. 

/ = frequency of radio wave. 

co — 2,7rf. 

E = electric field intensity of the radio 
wave. 

H = magnetic field intensity of the radio 
wave. 

J = total current (displacement plus con¬ 
vection current). 

V = velocity of electron motion in the 
ionosphere. 

i, j, k = right-handed set of mutually perpen¬ 
dicular unit vectors. 

The problem of propagation of radio waves 
through the ionosphere is of course solved by 
looking for the appropriate solution of Max¬ 
well’s equations. 

V X E =-- H (1) 

c 

V X H = — J (2) 

c 

In the case of the ionosphere, the free electrons 
present give rise to a convection current NeY 
so that the total current J is given by 

J = E + NeV. (3) 

Furthermore the motion of the electrons must 
satisfy the force equation 

eE - mV - vmW + —V X H° = 0. (4) 

c 

Here the term (e/c) V X H° is the force due to 
the magnetic field of the earth. This term 
causes the paths of the electrons which are 
oscillating under the influence of the electric 
field of the radio wave to be bent (if moving 
with uniform speed they would be bent into 
circular helices) and thus causes the wave to 
be split into ordinary and extraordinary com¬ 
ponents. The term vmV is a dissipative force 
produced by collisions with the neutral air 
molecules. This term gives rise to the absorp¬ 
tion of the wave. 

Since we are looking for a plane-wave solu¬ 
tion, we substitute into equations (1) to (4) 
the plane-wave function, exp [£(&/*, n . r— <ot]. 
Here /a denotes the complex index of refrac¬ 
tion, n denotes a unit vector normal to the wave 
front, and r is a vector denoting the position of 
the field point from a fixed origin. The field 
equations can be satisfied providing the pa¬ 
rameters fulfill certain relations which are 



26 


N BS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


found by substituting the plane-wave function 
into the equations. Thus n comes out to be 
double-valued showing the splitting into ordi¬ 
nary and extraordinary wave components. For 
each of these values of the index of refraction, 
the field equations can be solved for the electric 
and magnetic intensities. When such a solution 
is carried out for the case of plane waves in¬ 
cident on the ionosphere from a transmitter on 
the ground, it is found that the ordinary and 
extraordinary waves are both elliptically polar¬ 
ized, the state of polarization being indepen¬ 
dent of the polarization of the incident wave. 
This result is important for short-wave direc¬ 
tion finding since it means that no matter what 
the transmitted polarization may be, the down¬ 
coming ionospheric waves will, in general, be 
elliptically polarized and can therefore be re¬ 
solved into two plane-wave components polar¬ 
ized parallel and perpendicular respectively to 
the plane of incidence and having a suitable 
phase difference. The difference in azimuthal 
response of a direction finder to the several 
components of such a field will therefore result 
in inaccuracies of the bearing. 


2 - 2 - 5 Effects of Fading 

These inaccuracies will vary as the state of 
polarization of the downcoming wave varies, 
particularly since both the phase and ampli¬ 
tude of the parallel and perpendicular compo¬ 
nents varies. Norton 5 has treated this problem 
quantitatively by considering the effect of fad¬ 
ing of the ordinary and extraordinary waves 
on the resultant downcoming wave (equal to 
their vector sum). 

There are two causes for the fading of iono¬ 
spheric waves: phase interference between 
waves traveling along slightly different paths 
in the ionosphere, and changes in the absorption 
of radio waves caused by variations in the ioni¬ 
zation distribution in the ionosphere. The phase 
interference is responsible for the rapid 
changes in intensity which occur from minute 
to minute, while the changes of absorption are 
responsible for slower changes in the average 
level of the received fields which occur from 
hour to hour and from day to day. These latter 
changes can be neglected for this work. 


The fading caused by phase interference is 
a result of the irregular nature of the iono¬ 
sphere. The ionosphere probably consists of 
clouds of ions distributed in such a manner 
that a single downcoming ionospheric wave 
actually consists of a large number of compo¬ 
nent waves, each of which has been reflected 
at a slightly different place in the ionosphere. 
These separate wave components, since they 
have traveled along slightly different paths 
through the ionosphere, will arrive at the 
ground with random relative phases. This fad¬ 
ing has been found experimentally 6 to follow 
a distribution law first derived by Rayleigh 
which gives the resultant of a large number of 
waves of the same frequency but of arbitrary 
phase. Using this distribution law, the fading 
of the ordinary and extraordinary downcoming 
waves can individually be determined and 
therefore also the fading of the resultant down- 
coming wave. Clearly this fading of the ordi¬ 
nary and extraordinary waves gives rise to 
variations in the state of polarization of the 
resultant downcoming wave with concomi¬ 
tant variations in the d-f polarization error. 
Such variations in state of polarization have 
been observed experimentally and account for 
the swinging of bearings observed in practice. 
The complete analysis of the fading problem 
leads to the general conclusion that, except for 
the two special cases given below, the relative 
amplitude and relative phase of the parallel 
and perpendicular components of a downcom¬ 
ing ionospheric wave will have a random dis¬ 
tribution, the distribution being more nearly 
random the higher the frequency. The average 
of a series of swinging bearings will then be 
close to the true bearing (excluding lateral 
deviation) except for frequencies near the 
magneto-ionic frequency or near the maximum 
usable frequency. 

Several conclusions relative to direction find¬ 
ing may be drawn from the preceding discus¬ 
sion. It is clear that the state of polarization 
of downcoming ionospheric waves will vary 
over wide limits, there being times when the 
parallel component only is present and times 
when the perpendicular component only is 
present. The phase between these two compo¬ 
nents also can have any value. These variations 
have been observed experimentally by direct 





ANALYSIS 


27 


measurements of the state of polarization of 
downcoming waves. Busignies 7 has proposed 
that some of the large bearing errors previ¬ 
ously ascribed to lateral deviation may perhaps 
be accounted for as polarization errors occur¬ 
ring when the desired component of the wave 
is practically zero. Clearly, for these cases even 
a direction finder having a very low standard 
wave error would exhibit a large bearing error. 
It is therefore evident that the reduction of the 
standard wave error alone will not prevent the 
occurrence of a certain percentage of large 
bearing errors. Since the period when this hap¬ 
pens will usually be short, NBS and Busignies 


2.2.6 E£f ec t of Ground on Total Field 
at the Direction Finder 

The preceding discussion of the nature of the 
downcoming waves must be supplemented by a 
consideration of the effect of the ground reflec¬ 
tion on the resultant field at the direction 
finder. The response of the antenna system in 
this resultant field can then be calculated and 
therefore also the polarization errors. 

The wave coming down from the ionosphere 
will be effectively a plane wave since it will 
have come from a great distance. It will be re¬ 
flected from the ground, assumed here to be flat 


x* 



have independently proposed a direction finder 
in which the taking of bearings is automati¬ 
cally prevented unless the state of polarization 
of the incident radio wave is favorable for ac¬ 
curate bearings. The antenna system in this 
method is used not only to take bearings but 
also to measure the relative polarization of the 
downcoming wave and thus to control the in¬ 
dications of the direction finder. 

A direction finder using spaced, horizontal 
loop-antenna elements has been suggested by 
NBS 3 and others as having favorable proper¬ 
ties for accurate direction finding. The opera¬ 
tion of such a direction finder requires a per¬ 
pendicular component in the downcoming wave. 
The preceding analysis has shown that such 
components will be present approximately 
equally with the parallel components, so that 
direction finders designed for either component 
are feasible. 


and homogeneous, so that the total field will be 
given by the vector sum of the direct and 
ground-reflected waves. The reflected wave can 
be calculated by using Fresnel’s equations; some 
typical cases will be given here to illustrate the 
large magnitude of the effect on the resultant 
field. This will have an important bearing on 
the selection of a suitable figure of merit for 
polarization error in direction finders. Figure 1 
shows the geometry of the problem for the case 
in which the electric vector lies in the plane of 
incidence. The electric vector E Pid of the inci¬ 
dent downcoming wave is shown as a dotted 
line while the electric vector E p>r of the corre¬ 
sponding ground-reflected wave is shown as a 
solid line. Since the vertical component E p>z of 
the resultant field is in general out of phase 
with the component parallel to the ground, E P)X> 
the resultant vector will rotate in an ellipse 
in the i-k plane (the plane of incidence). 















28 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


Figure 1 has been drawn for the particular 
case of a downcoming wave on a frequency of 
1 me arriving at an angle of elevation ip = 4° 
over land of average conductivity <r = 5 X 10 -14 
emu and with a dielectric constant K — 15; the 
ellipse represents the resultant field at a height 
z = 10 feet. 

The general equations for the reflected wave 
are given by Fresnel’s formulas as follows: 

E p , r = RpE p> d (5) 

E n , r = R n E M (6) 


where the plane-wave reflection coefficients are 
given by 

n~ sin ip — V n 2 — cos 2 ip 

R v = ~~z —:-;- , == v*) 

n sin ip + v n ~ — cos " t 
^ _ sin yp ~ Vn 2 — cos 2 ip ^ 

sin ip + \/n 2 — cos2 ’A 

In these equations n is the complex index of 
refraction of the earth and n 2 = K + iX where 
X = 1.797 X 10 15 <r//; a is the conductivity of 
the ground in emu, / is frequency in megacycles 
per second, and K is the dielectric constant of 




" <? 

<// p = 8° ^p'12* 8 20* *Pp s 30* ^p = 90* 


t--90" 


INCIDENT WAVE 


46 MC OVER LAND : Ksl5, <T *5X10 emu 
X = 1.9536 




Figure 2. Vector representation of ground-reflected wave. E,, parallel to plane of incidence; E ft normal to 
plane of incidence; dotted lines indicate incident plane; full lines indicate ground-reflected wave; = 
Brewster’s angle. 























ANALYSIS 


29 







FIELD AT THE SURFACE FIELD */4 ABOVE THE SURFACE 


(A) X * 179,731, K = 80 

(B) X * 89.8655; K= 15 

(C) X * 1.95360; K= 15 

Figure 3. Vertical and horizontal components of total field near earth’s surface when plane wave of unit 
intensity is incident at angle M'. 


















30 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


the ground. The numerical magnitude of the 
quantity X is of fundamental importance in 
connection with the effect of the ground on the 
radio waves, the nature of the reflection being 
radically different in the two cases when X is 
very much larger than the dielectric constant 
and when X is very much smaller than the di¬ 
electric constant. 


Table 1. Values of X for various frequencies and 
for the ground conductivities normally encountered 
in practice. 


Description < 

x(emu) 


/(me) 

X 

Land, low conductivity 

1 

X 

10- 

-14 

1.6 

11.23 

Land, average conductivity 

5 

X 

10- 

-14 

1 

89.86 

Land, average conductivity 

5 

X 

10- 

-14 

5 

17.97 

Land, average conductivity 

5 

X 

10- 

-14 

46 

1.95 

Sea water 

5 

X 

10- 

-11 

5 

1.79 X 10 4 

Sea water 

5 

X 

10- 

-11 

46 

1.95 X 103 

Copper 5 

.8 

X 

10- 

-4 

10 

1.04 X 10 11 


The dielectric constant varies over a much 
more limited range, from unity for air to 80 for 
water. The dielectric constant for land varies 
from about 5 up to about 30. In many of the 
calculations on specific direction finders given 
later on in this report, computations were de¬ 
sired for typical cases; accordingly average 
ground constants of K = 15 and o- = 5X10- 14 
emu were assumed. 

Figure 2 illustrates the intensity and phase 
relationship (as given by R p or R n ) between 
the incident and the ground-reflected waves for 
the case z — 0. The vector diagrams give the 
cases in which the electric vector is either par¬ 
allel or perpendicular to the plane of incidence 
for three different frequencies and sets of 
ground constants. The upper diagram corre¬ 
sponds to an ultra-high frequency and ground 
constants such that X < < K ; the middle dia¬ 
gram corresponds to a frequency in the stand¬ 
ard broadcast band and average ground con¬ 
stants; and the bottom diagram corresponds 
to reflection at 500 kc over sea water. The two 
upper diagrams therefore show the effect of 
different frequencies over average land while 
the two bottom diagrams show the effect of 
changing the index of refraction at one end of 
the band. On these diagrams the intensities of 
the direct and ground-reflected waves are rep¬ 
resented by the length of the arrows while the 
phase of the ground-reflected wave is repre¬ 
sented by the angle at which the solid arrow is 


drawn; the phase of the incident wave is zero 
in each case. Each of the solid arrows repre¬ 
sents the ground-reflected wave for a different 
angle of incidence of the incident wave. 

The resultant wave at a d-f antenna will be 
the vector sum of the incident and ground-re¬ 
flected waves. These must be added in proper 
phase, the phase difference being caused in part 
by a phase shift introduced on reflection and in 
part by the difference in path length. When 
this is done, the total field intensity is found to 
be given by the following equations: 

E P , X = E P , d sin + 11 — R p e*™( z/X) sin * | (9) 

E p ,z = Ep, d cos \f/ 11 + R P e^ z/X) sin * | (10) 

En = E n , d 11 + R n e*™W ) 8in * |. (11) 

Figure 3 shows these three components for 
the case when E Ptd = E n>d = 1 for heights z 
= 0 and A/4 above ground and for the same 
frequencies and sets of ground constants as for 
Figure 2. Notice that the vertical component 
E p ~ increases, at z = 0, with increasing values 
of X (increasing conductivity or decreasing 
frequency) while the horizontal components 
E VjX and E n decrease with increasing values of 
X. On the other hand, at a height A/4 above the 
ground, E p>x and E n increase with increasing 
values of X. These figures indicate that a direc¬ 
tion finder designed to respond to the E p com¬ 
ponent of the wave should be located over 
ground with the highest possible conductivity 
to increase E p>z and decrease E n which is usual¬ 
ly responsible for polarization errors. 

The relative values of the total field compo¬ 
nents will depend on the height z at which the 
fields are considered. Since E p>z is usually the 
desired field component while E n is the un¬ 
desired component, the ratio \E n /E PjZ \ is shown 
in Figures 4 and 5 for frequencies of 2 and 20 
me as a function of height z of the receiving 
point. These curves are drawn for equal parallel 
and normal components in the downcoming 
wave and are given for several ground con¬ 
stants and angles of elevation. The important 
result to be noted is that, under most condi¬ 
tions, the effect of the ground is to suppress the 
horizontal component in the resultant wave in 
comparison to the vertical component. When E n 
is the undesired field component, it should be 
clear from these figures that the d-f antenna 






ANALYSIS 


31 


system should be located as near as possible to 
ground and over ground with the highest pos¬ 
sible conductivity. At 20 me, Figure 5 shows 
that for large angles of elevation the vertical 
component rather than the horizontal compo¬ 
nent is often suppressed. From these two fig¬ 
ures it is clear that in the case of a direction 



O 10 20 30 40 50 60 70 

HEIGHT ABOVE THE GROUND Z IN FEET 


Figure 4. Ratio of resultant horizontal to vertical 
electric field components when plane wave with 
equal parallel and normal components is incident 
on ground at angle of elevation 4'. 


finder designed to respond to E n , such as the 
spaced horizontal loop-antenna type, the an¬ 
tenna system should not be placed too close to 
the ground. In fact an optimum height of about 
A/3 to a/4 is indicated on Figure 5. 

Figure 6 indicates the maximum height for 
direction finders designed to reject E n . This 
figure shows the height above perfect ground 
(<r — oo) at which \E n /E p , g \ equals \E n>d /E v , d \. 
Below this height the ground acts to suppress 
E n so that the direction finder should be kept 
below the limit. Over imperfect ground this 
limiting height will be even less. 

So far expressions for the electric field com¬ 
ponents only have been considered. In the case 


of loop-antenna direction finders the magnetic 
field components are also needed and so are 
given below for the resultant field of an inci¬ 
dent and ground-reflected wave using Heavi- 
side-Lorentz units as before. 


Hp ,x 

= E n , d sin \P 1 1 — R n e < 2 ^ sin * | 

(12) 

H P ,z 

= E n ,d cos \f/ 11 + R n e* Ti sin * | 

(13) 

Hn 

= E p ,d 1 1+ Rpe 8in *| . 

(14) 


A consideration of these equations indicates 
the effect of the height above ground when the 
direction finder uses loop-antenna elements, that 



z 


HEIGHT IN FEET ABOVE THE GROUND 

Figure 5. Data similar to that in Figure 4 except 
at frequency of 20 me. 

is, magnetic dipole elements. In particular, for 
the spaced, vertical, coaxial loop-antenna sys¬ 
tem, the undesired field component is E n , while 
the desired component is H n . Since 

| En/Hn | = — | E n /Ep,z | COS \J/ , (15) 

Figures 4 and 5 may be used in connection 
with the spaced coaxial loop-antenna system 
simply by multiplying the values given in the 
curves by the factor cos 















































































32 


nbs high-frequency direction-finder research 


227 The Calculation of Polarization Errors 

The preceding sections have shown how the 
total field components at a direction finder are 
determined for downcoming ionospheric waves 
for any frequency or values of ground con¬ 
stants. The response of the antenna system in 
such a field must next be calculated, including 
the effect of both the desired and undesired 



FREQUENCY f IN MEGACYLES PER SECOND 

Figure 6. Maximum height above perfect ground 
at which | EJE p>z \ = | E n>d /E pd |. 

field components on the d-f azimuthal direc¬ 
tional pattern. The departure of this directional 
pattern from the ideal pattern obtained for the 
desired field component alone is the cause of 
the polarization error of the antenna system. 
The difference in bearing given by the ideal 
pattern and the actual pattern is equal to the 
polarization error. When the incident wave is 
assumed to have equal parallel and perpendic¬ 
ular components of such a phase relation as to 
cause maximum bearing error and to have an 
angle of elevation of 45°, the calculated error 
will be Barfield’s standard wave error. 


The response of the antenna system to any 
field component will be proportional to that 
component and will have a certain functional 
dependence on the azimuthal angle <j> and angle 
of elevation ^ of the incident wave. The azi¬ 
muth angle <f> is the angle between the plane of 
propagation and the vertical plane passing 
through the centers of the two spaced antenna 
elements. The output voltage V of the antenna 
system can therefore be written as follows, 
where the voltages induced in the antenna 
elements and in the feeders are arbitrarily 
separated. 

V = V antenna “1“ Ffeeders (lb) 

V an t = h x E v , x F x (<t>d) + lizEp^F 

+ hyE n Fy((j),\J/) (17) 

V feed = k x Ep t xfx(<t>,'l') 4" &z-Fp,z/z($,l/') 

+ k y Enfy(<t>,lp) . (18) 

In these equations the proportionality constants 
h and k , corresponding to desired and un¬ 
desired pickup respectively, are to be deter¬ 
mined experimentally. The feeder voltage here 
is meant to include all undesired voltages. The 
National Bureau of Standards has adopted a 
standard value of i p as zero in this work so that 
the values of h and k may be determined by 
measurements on the ground, that is, at hori¬ 
zontal incidence; this seems to be possible for 
most direction finders. The functions 
and /(</>,i/0 give the directional dependence of 
each term in the response and are complex 
quantities including the phase of each term. 
The functions are dimensionless, while V is to 
be measured in volts and the field intensities 
in volts per meter. In this case the constants h 
will be measured in meters. These functions 
will depend on the particular antenna system 
being considered and are used in the preceding 
equations as holding for a single pair of spaced 
antenna elements, that is, for a rotatable type 
of direction finder. Fixed direction finders will 
be considered later. 

Equations (16) to (18) include the effect of 
the ground reflection. In most cases the func¬ 
tions and can be accurately writ¬ 

ten down a priori from a knowledge of the an¬ 
tenna structure. The NBS procedure usually 
used is to assume a priori the dependence on ^ 
while the dependence on <j> can he determined 
by measurements on the ground. The total field 






























































ANALYSIS 


33 


components H p>x , H v>z , and H n could be used in 
equations (17) and (18) instead of the electric 
field components with equal generality. The 
fields can induce voltages in the antenna sys¬ 
tem directly or indirectly through pickup and 
reradiation of wires, supporting posts, etc. In 
any case, the total output voltage can be found 
as a function of <£; that is, the azimuthal direc¬ 
tivity pattern will be given by equation (16) 
and can be rewritten as follows: 


V 


- E P ,z$y 


(h x F x + k x f x 


-j- (hzFz -f- kzfz) -f~ QlyFy + k y f % 




(19) 


Here the shape of the directional pattern is 
seen to be determined simply by the ratio of 
the total field components, which in turn are 
fixed by the ratio of the parallel to perpendic¬ 
ular components in the downcoming wave. 
Therefore the units used for E or H can be 
arbitrary when calculating polarization errors 
since only the ratios of the field components are 
needed. By setting E n>d = E Pyd and $ = 45°, the 
directional pattern for standard waves can be 
found. In most direction finders many of the h 
and k constants appearing in equation (19) 
are zero or negligibly small; also the different 
h or k constants are sometimes equal. This sim¬ 
plifies the problem considerably. 

Equation (19) gives the phase and amplitude 
of the output voltage as a function of <f>. This 
directional pattern therefore can be compared 
with the ideal, desired, directional pattern for 
either the phase-comparison or amplitude-com¬ 
parison d-f type. The h constants correspond¬ 
ing to the wanted response should be large 
compared to the k constants corresponding to 
the unwanted response in order that the dis¬ 
tortion of the directional pattern and therefore 
the polarization error be a minimum. The ratios 
of h to k can therefore be used as figures of 
merit for judging the freedom from polariza¬ 
tion error of a given direction finder. The use 
of these ratios for this purpose has the advan¬ 
tage that the ratios are independent of the 
ground constants and height of the direction 
finder above the ground. Each type of direction 
finder will in general require a different proce¬ 
dure to be used in determining the polarization 
error from equation (19). For example, in 


those direction finders which determine a bear¬ 
ing by rotating the antenna system until a mini¬ 
mum in the output voltage is found, the bear¬ 
ing, </>, will be given by the equation 

,if F i* 0 ' <2o) 

Since in most rotatable systems the true bear¬ 
ing is given b y <f> = 90°, the bearing error or 
polarization error e will then be given by 
90° — $ where </> is the azimuth angle satisfying 
equation (20). The value of e obtained from 
equation (20), or other determining equations 
depending on the direction finder, will be a 
function of the phase angle of the various 
terms in V and these in turn will be variable 
since the phase of the components in the down¬ 
coming wave are random. The maximum value 
of e is usually the value desired. This is then 
determined by letting the relative phase of E p>d 
and E n>d be varied until the maximum value of 
e is found. 

For the case of fixed-type direction finders, 
the directional pattern is found for each pair 
of antennas and the bearing determined from 
these patterns in a manner depending upon the 
particular d-f type. A common example is the 
type using a goniometer or similar principle 
with the antenna pairs at right angles to each 
other, such as the Western Electric-Civil Aero¬ 
nautics Administration and International Tele¬ 
phone and Radio fixed direction finders. For 
this type the observed bearing 6 relative to the 
plane through one of the fixed pairs of anten¬ 
nas will be given by 

tan0=j^| (21) 

where V 1 and V 2 are the output voltages of the 
two pairs of antennas. The correct bearing </> 
relative to the same plane is given by 

tan <f) = j ~ J = tan 0 (22) 

only when V 1 andF 2 follow the ideal directional 
patterns for which the antenna systems were 
theoretically designed. The polarization error, 
e, is in this case given by <f> — 0. The maximum 
value must again be determined by varying the 
phase of the field components in the down¬ 
coming wave. 



34 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


Typical Calculation 

To illustrate the method of calculating polar¬ 
ization errors the case of a rotatable, balanced 
H antenna will be worked out. 3 

The rotatable balanced H-antenna direction 
finder is a spaced, electric dipole system of the 
Adcock variety in which the dipoles are dif¬ 
ferentially connected by means of horizontal 
transmission lines. These lines are sometimes 
enclosed by a metal shield and sometimes not. 
For vertical dipoles, the antenna elements will 
respond directly only to the vertical electric 
component of the field of the radio wave. Volt¬ 
age may be induced in the dipoles by the hori¬ 
zontal component of the field, indirectly, if the 
coupling to the dipoles of some other part of 
the system excited by the horizontal field com¬ 
ponents is not negligible. The polar response 
pattern of a vertical dipole is nondirectional 
in azimuth, while it is a figure eight having cir¬ 
cular lobes in the vertical plane when the 
length of the dipole is small compared to the 
wavelength A. In general the vertical directional 
pattern of a dipole depends on its length and 
height above ground. In the case of the direc¬ 
tion finders measured by NBS, the antenna was 
always short enough to be considered as a pure 
doublet with a figure eight response pattern, at 
least for elevation angles if/ up to 45 to 60 de¬ 
grees. The response of the antenna elements 
will therefore be taken to be proportional to E p , z 
alone and the pattern will thus be a figure eight 
in the vertical plane. The function, F z (<f>,if/), 
however, includes not only the directivity 
function for a single dipole but also for the two 
dipoles differentially connected together. The 
total dipole response will be the vector differ¬ 
ence between the voltages induced in the indi¬ 
vidual dipoles and so will be proportional to 
twice the sine of half the phase difference be¬ 
tween these voltages. This phase difference will 
be 2tt(cZ/A) cos if/ cos <f> where d is the spacing 
between the dipoles. Accordingly the total out¬ 
put voltage is proportional to 2 E v>z sin [?r(d/A) 
cos if/ cos <f>\ or, when d /A is small as is generally 
the case, to 2irE p>z (d/\) cos if/ cos <f>. It there¬ 
fore follows for this case that 

\F 8 ((f), if/) | = cos if/ cos <f>, 
or if e = 90° — (f>, 

| F z ( c, if/) | = cos if/ sin e. 


The balanced H antenna also has an unde¬ 
sired response to the horizontal components of 
the field E n and E p , x . The mechanism of this 
response is not completely understood, except 
that it is caused by the voltage induced in the 
horizontal feeders or the shield surrounding 
the feeders. Clearly the proportionality con¬ 
stant for this pickup will be the same for re¬ 
sponse to both E n and E p>x . Also the response 
will be nondirectional in the vertical plane 
since the feeders can be considered to act as a 
horizontal antenna (loaded by the dipoles, un¬ 
less separated, for example, by cathode follow¬ 
ers). The following explanations have been 
proposed for this unwanted response: (1) The 
system is unbalanced because the lower halves 
of the dipoles are closer to the ground than the 
upper halves. (2) The feeders or feeder shield 
have unbalanced coupling to the dipoles. In 
either case the directional pattern for this re¬ 
sponse would be expected to be the same as 
that of a horizontal doublet (since the length 
of the feeders is short compared to A) . Finally, 
the azimuthal response pattern can be deter¬ 
mined by measurements and has been found to 
be that of a horizontal doublet. It follows there¬ 
fore that 


and 

1 = COSe 

(24) 


| fz(*,<l>)\ = sine. 

(25) 

Since 

for this antenna system h x 

II 

cS* 

II 


= 0 and k x = k y , the total output voltage, drop¬ 
ping the subscripts, is 
V = hE PyZ cos if/ sin e + kE p>x e iA sin e 

-f kE n e l & cos e. (26) 

Here /3 is the phase angle between the output 
voltage induced by E n and that induced by E p>z . 
It can have any value in practice as already ex¬ 
plained. The phase angle A is in part made up 
of a phase shift between E p>x and E PjZ intro¬ 
duced by the ground reflection and in part 
caused by phase shift in the antenna circuit 
depending upon the differential antenna con¬ 
nection and the exact mechanism of horizontal 
response. 

Equation (26) gives the d-f azimuthal direc¬ 
tional pattern. The ideal, desired pattern is the 
pattern obtained when either k or E n is zero. 
This will be a figure eight on a polar diagram 
with a null at e = 0, the true bearing. In equa- 


( 23 ) 




MEASUREMENT OF POLARIZATION ERROR 


35 


tion (26) there will not be a null but simply a 
minimum response unless p = 0. For this case 
(p = 0) the figure eight will be rotated so that 
the null does not occur for e = 0. Since the 
bearing is taken as that value of e for which 
| V | is a minimum, an incorrect bearing will be 
obtained. In general, the response pattern will 
not have a null but a broad minimum and in 
addition will be rotated. By solving the equation 
d\v\/de = 0, the bearing error or polarization 
error can be obtained (e equals the bearing 
error since the true bearing is given by e = 0). 
It is found that the error is a maximum when 
p = 0, that is, for cophased output voltages. 
The maximum polarization error is given by 


tan 


\kE n \ _ 

\hE PtZl cos if/ + kE PsX c iA |* 


(27) 


For downcoming ionospheric waves, the val¬ 
ues of E P}X , E PjZ , and E n can be obtained from 
equations (9), (10), and (11). It is clear that 
only the ratios of the field components need be 
known. Also only the pickup ratio h/k need be 
known as seen by rewriting equation (27) as 


tan < 


I hEp z . E p a 


(28) 


In general, the h and k constants only need be 
measured to within a constant factor since this 
constant can always be factored out of the ex¬ 
pression for V and so will not affect the shape 
of the directivity pattern. 

By taking E Pjd = E n>d and if/ = 45°, equation 
(27) gives the value of the standard wave 
error of the balanced H antenna. However, the 
polarization error for all values of if/ can be 
obtained once h/k is known. 

In general, the polarization error is smaller, 
the larger the pickup ratio h/k, as seen from 
equation (28). The National Bureau of Stand¬ 
ards has proposed the use of the pickup ratio 
as a figure of merit for polarization error in 
direction finders. It is clear that the pickup 
ratio is independent of the ground constants 
and of the height of the direction finder above 
the ground and so lends itself to the compari¬ 
son of different direction finders of the same 
type, that is, following the same law for polar¬ 
ization error, such as equation (28). This 
comparison as to accuracy can therefore be 


separated from the complicating influence of 
the ground and height above ground. Once this 
fundamental constant is known, not only the 
standard wave error but the polarization errors 
for all values of if/ can be determined for any 
particular ground and antenna height. A curve 
of e vs if/ can be plotted; it is this complete 
curve which should be compared with similar 
curves of other types of direction finders to 
compare their accuracy relative to polarization 
errors. The pickup ratio also furnishes a figure 
of merit by which the progress of development 
work on a particular direction finder can be 
judged. The effect of changes in the design can 
thus be studied and the cause and mechanism 
of polarization errors isolated. 

A useful figure of merit somewhat similar to 
the standard wave error is the polarization er¬ 
ror for a horizontally incident wave with equal 
E v>z and E n components of such phase as to 
cause maximum bearing error. The error for 
this wave will be called the horizontal wave 
error, e 0 , while Barfield’s standard wave error 
will be denoted by e 45 . For the cases of the 
rotatable H-antenna direction finder, tan e 0 — 
k/h. The e 0 error is independent of the ground 
constants or height of the direction finder 
above the ground. 

23 MEASUREMENT OF POLARIZATION 
ERROR 

2.3.i Plane Wave Measurements 

The preceding section has shown that the 
problem of the measurement of polarization 
error can be reduced to that of measuring the 
pickup factors, the constants h and k. In most 
direction finders the response of the antenna 
system can be reduced to a single term by plac¬ 
ing the antenna in a suitable plane-wave radia¬ 
tion field, having only one component such as 
E n or E PjZf and orienting the antenna to the 
proper azimuth angle. By properly choosing 
the field, the antenna output voltage can then 
be made to be 


v t = hEflM) 

(29) 

V 2 = kEJ(w) 

(30) 


where E , and E , are the particular field com¬ 
ponents used. The field intensity can be meas- 





36 


nbs high-frequency direction-finder research 


ured by means of a field-intensity meter, and 
the output voltage V of the antenna system 
also determined. The value of F(<f>,\p) or f(<f>,\p) 
is known, so that equations (29) and (30) can 
be solved for the pickup factors. 


h = 


Vi 

EJF (<f>,\p) 


(31) 


k = 


EJ (<f>,ip) 

Usually the field used is one at horizontal i 


(32) 


dence so that xp = 0. This simplifies the tech¬ 
nique by allowing all measurements to be made 
close to the ground. Often the measurements 
must be made at horizontal incidence to reduce 
the response of the antenna to a single term; the 
presence of voltage corresponding to more than 
one term would require a knowledge of the 
phase of each term. Also the measurements are 
usually made at particular values of <f> in order 
to obtain various experimental advantages. 
The pickup factors can be defined as the output 
voltage per unit field intensity for azimuth and 
elevation angles such that F (<p,\p) and f(<f>,xp) 
are unity (provided they can assume such val¬ 
ues, as is usually the case). This is the reason 
for the designation pickup factor. 

Usually the azimuthal directional or re¬ 
sponse patterns of the system are determined 
by measuring V and E for the special fields al¬ 
ready indicated as a function of <f> (with xp = 0). 
If the response is defined as the ratio of V to E , 
that is, the output voltage per unit field inten¬ 
sity, these curves will be given by 

= AF(*,0) 


or 

= fc/4,0). 

The response is equal to the pickup factors if 
and when F or / is equal to unity. 

To illustrate the procedure just outlined, the 
case of the balanced H antenna will again be 
treated. When the antenna is placed in the field 
of a plane wave polarized parallel to the plane 
of incidence so that E v>x = E n = 0, the output 
voltage will be 

V = E PjZ cos ip sin e. (33) 

When ip = 0 and e = 90°, V and E PjZ are mea¬ 
sured, giving h = V/E PtZ in meters if V is 
measured in volts and E p>z in volts per meter. 
Similarly if a plane wave polarized perpendicu¬ 


lar to the plane of incidence is used so that 
E p x = E p>z = 0, the output voltage will be 

V = kEn cos C, (34) 

so that k = V/E n when e = 0. In this case the 
pickup factors are equal to the output voltages 
per unit field intensity at maximum response. 

2 3 2 Application to Buried U Direction 
Finders 

A difficulty arises when applying the method 
just outlined to the buried U-antenna direction 
finder. As its name indicates, the antenna con¬ 
sists of vertical electric elements connected by 
horizontal feeders or transmission lines which 
are buried below the surface of the earth. By 
this means the field intensity at the feeders is 
greatly reduced both because of the partial re¬ 
flection of the incident wave by the ground and 
the attenuation of the transmitted wave by 
absorption in the ground. The expression for 
the output voltage of this antenna system in 
the field of a downcoming ionospheric wave 
will consist of terms for the voltage induced in 
the antenna elements and terms for the voltage 
induced in the feeders. The voltage induced 
in the antenna elements will involve the de¬ 
sired pickup factor h and the field intensity at 
the antenna elements, while that induced in the 
feeders will involve the undesired pickup factor 
k and the field intensity at the feeders. Since 
the pickup factor k is the proportionality con¬ 
stant relating the output voltage of the antenna 
as a result of voltage induced in the feeder to 
the field intensity at the feeders, both the out¬ 
put voltage and field intensity must be meas¬ 
ured to determine k. Because the feeders are 
buried, the difficulty then arises of measuring 
the field intensity below the ground. This diffi¬ 
culty can be met, however, by a procedure to be 
described in which expressions are used for the 
field intensity at any depth A below the surface 
of ground having arbitrary constants. 

The procedure for avoiding the above diffi¬ 
culty in measuring k is as follows. In all meth¬ 
ods of studying polarization errors, the errors 
must be determined from a knowledge of the 
field intensities in an incident , downcoming 
wave. In the NBS method the output voltage 
of the antenna system must be calculated when 
these field intensities are given. This involves 






MEASUREMENT OF POLARIZATION ERROR 


37 


the calculation of the field intensity at the an¬ 
tenna components when the field intensity in 
the incident wave is given. However, the field 
intensity at the antenna components can be 
specified in terms of the field intensity at any 
other point. If this reference point is taken to 
be the same for the vertical antenna elements 
and for the buried feeders and to be above 
ground, both h and k can be measured by pro¬ 
cedures quite similar to those already given. 
The reference point could be taken at the cen¬ 
ter of the d-f antenna system at a height, z, 
above the ground. 

Before proceeding to a development of this 
procedure some specific points must be con¬ 
sidered concerning the expression for the out¬ 
put voltage of the vertical antenna elements 
and concerning the effect of a ground mat. In 
the other direction finders which have been con¬ 
sidered, namely those using elevated antennas, 
the field intensity was taken to be the same 
over the entire region occupied by the antenna 
elements. This assumption is probably not a 
good enough approximation for the buried 
U antenna, so that an integration over the an¬ 
tenna elements would be required to obtain a 
more accurate expression for the induced volt¬ 
age. However, in the simplified analysis to be 
given, this will not be done, the assumption be¬ 
ing made that the field intensity at the center 
of the antenna system can be used for comput¬ 
ing the voltage induced in the antenna elements. 
Some buried U-antenna systems use a large 
ground mat which must be considered when 
computing the field intensities above and below 
the surface of the ground. However, in the 
analysis to be given it will be assumed that no 
ground mat is present or that it is so small 
compared to the wavelength that its effect on 
the field intensity can be neglected. 

A derivation will now be given of the expres¬ 
sion for the total output voltage of the buried 
U antenna without ground mat as a function 
of incident angle as the principal parameter. 
The polarization errors can then be derived ac¬ 
cording to the particular indicating method 
used and so will not be given here. The voltage 
induced in the feeders by E PjX can be safely 
neglected since E PjX will be very small except, 
perhaps, for very large angles of elevation of 
the incident wave. The output voltage as a re¬ 


sult of voltage induced in the antenna elements 
will be hE PiZ cos ^ sin e, where E PiZ is the verti¬ 
cal electric field intensity at the center of the 
antenna system (a height 2 above the ground). 
The output voltage as a result of voltage in¬ 
duced in the feeders will be kE n>t cos e, where 
E n>t is the transmitted horizontal electric field 
intensity at the depth, A, below the ground 
where the feeders are buried. From the ma¬ 
terial below on d-f sites the value of E n>t is 
taken as 

E n . t = E n ,d I (1 -\-Rn) e ( 2iti/X)(z sin <J» +A\/ w 2 -cos 2 4») | (35) 

while E v>z is given by equation (10) as 

E p , z = E P , d cos * 11 + R P e 4 ™ sin * |. (36) 

However, the field intensity E n at the height z 
above the ground was given by equation (11) as 

En = E n ,d 11 + Rne 4 ™ sin * |. (37) 

Therefore, if the direction finder is placed in 
the field of a perpendicularly polarized down¬ 
coming wave in order to measure k } E n can be 
measured at the height z and E n>d calculated 
by means of equation (37). Using this value of 
E n>d the value of E n>t at the depth A can be 
determined by using equation (35). The value 
of k can then be found from the equation 


— 71,1 v. 

where V n is the measured output voltage. It is 
clear that this procedure effectively measures 
k in terms of the field intensity at the feeders 
by measuring the field intensity above the 
ground and then calculating the field intensity 
at the depth A from this measurement. To do 
this, the ground constants must be known. 
However the constant k still is independent of 
the ground constants or the depth of the feed¬ 
ers below the ground and so is a useful figure 
of merit for measuring undesired pickup. Once 
k is measured, the output voltage of the antenna 
system for any downcoming ionospheric wave 
will be given by 

V = hE P 'Z cos i p sin e + kE n ,t cos e e % $ (39) 

with E PjZ and E n>t given by equations (36) 
and (35) respectively. Here (3 is the arbitrary 
phase angle already discussed in connection 
with equation (26). 

When measuring k in practice, a local trans¬ 
mitter is used which does not generate plane 






38 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


waves. In this case the fields E n and E n>t are 
given below in this report and in equation 
(194), page 64, of the Norton report. 5 The 
pickup factor k is then determined by the pro¬ 
cedure outlined except that these equations for 
a local transmitter must be used rather than 
equations (35) and (37). After finding k by 
using the equations for a local transmitter , 
equations (35) and (39) must still be used for 
calculating the polarization error for down¬ 
coming ionospheric waves. Equations (35) and 
(37) can be used to determine k even when a 
local transmitter is used provided that the 
transmitter is at a great enough distance from 
the direction finder and at an angle of elevation 
large enough so that equations (35) and (37) 
are valid. This point is discussed in detail be¬ 
low. When the local transmitter is used near 
the ground, however, the exact expressions for 
the field from a local transmitter must be used 
to determine k by measurements above the 
surface of the ground. 

2 * 3 * 3 Local Transmitter Measurements 

The pickup factors h and k which determine 
the response of a direction finder were defined 
for plane waves such as downcoming iono¬ 
spheric waves. The procedures in which special 
plane waves are used to make possible the 
measurement of h and k must be modified in 
practice since the only practicable means of 
generating such special fields is by the use of a 
local transmitter placed a relatively short dis¬ 
tance from the direction finder. The wave from 
such a transmitter will simulate an ionospheric 
wave only approximately, thus introducing into 
the experimental technique several difficulties 
which must now be considered. 

Two methods of determining polarization er¬ 
rors have been introduced by NBS and RCA 4 
respectively. 13 The NBS method uses a local 
transmitter near the ground while RCA uses 


b The RCA method of measuring polarization errors 
differs from the NBS method already outlined as fol¬ 
lows. The special fields used are those of downcoming 
waves, first polarized parallel to the plane of incidence, 
then perpendicular to the plane of incidence. The out¬ 
put voltage of the direction finder is measured for each 
wave and will be V v and V n respectively when oriented 
for maximum response. The polarization error e in the 
field of both waves will be given by tan e = Vn/V v . 


an elevated transmitter to generate the fields 
required in the two methods; accordingly the 
first method is a horizontal incidence method 
while the second utilizes downcoming waves. 
In both methods the two special waves gener¬ 
ated are, first, a wave polarized parallel to the 
plane of incidence so that E n — 0 and, second, 
a wave polarized perpendicular to the plane of 
incidence so that E p — 0. In the NBS method 
the wave of parallel polarization arrives at 
horizontal incidence so that E p>x = 0 also. In 
practice these conditions are only approxi¬ 
mately met, the deviations from the desired 
fields being as follows. 

The E p>x Wave Component 

In the NBS method the presence of the 
ground causes a wave tilt which gives rise to 
a small E VyX component. The wave tilt is usu¬ 
ally less than 10° so that this component can 
be neglected in practice, especially since it in¬ 
duces voltage which is small in comparison to 
that of the E PjZ component. The pickup factor 
for the E p>x component is small because it is 
usually the feeders which are responsible for 
such pickup. 

Generation of Pure Fields 

It is very difficult to generate a wave polar¬ 
ized perpendicular to the plane of incidence 
without also generating some parallel compo¬ 
nent. The stray parallel component becomes 
relatively more important, the greater the dis¬ 
tance of the local transmitter from the d-f site, 
since the ground very rapidly attenuates the 
perpendicular component in comparison to the 
low attenuation of the parallel component. The 
more accurate the direction finder, if designed 
to reject the perpendicular component and to 
respond to the parallel component, the greater 
must be the purity of the field to measure the 
smaller polarization error which such an im¬ 
proved direction finder would have. 

The Adcock type of direction finder, in which 
spaced electric monopole or dipole antenna ele¬ 
ments are balanced against each other, relaxes 
the stringent conditions for purity of the field 
since the response of the system to E n can be 
measured with the antenna oriented at the null 
position for E p (and vice versa, although the 




MEASUREMENT OF POLARIZATION ERROR 


39 


problem of generating a field with E n negligible 
is not difficult). For this reason NBS has meas¬ 
ured the pickup factors of such direction find¬ 
ers with the antenna system oriented in the 
proper null position. 

Careful design of the local transmitter helps 
to prevent the generation of undesired field 
components. The antenna should contain or be 
an extension of the shield containing the oscil¬ 
lator and batteries so that current flow will be 
possible in the desired paths only. 

Finally, a flat homogeneous site should be 
used when making measurements of polariza¬ 
tion error. 

2 3 4 Field Generated by a Local Radiator 

The Surface Wave Component 

The presence of the local transmitter near 
the ground results in the generation of a sur¬ 
face wave component in the wave in addition 
to the direct and ground-reflected waves. 5 The 
expressions for the field generated by a local 
transmitter were also obtained by Burrows, 8 - 10 
who used a somewhat different terminology 
from that used here. The surface wave termi¬ 
nology will be used in this report since much 
of the work was carried out by using the equa¬ 
tions and methods of K. A. Norton. The vector 
sum of the direct and ground-reflected waves is 
called the space wave. The space wave is the 
only wave present at a direction finder for 
downcoming ionospheric waves, so that the sur¬ 
face wave component prevents the simulation 
of such waves by the use of a local transmitter. 
The presence of the surface wave introduces no 
difficulty in the NBS method since the total 
field intensity is measured, the effect of the 
surface wave thus being allowed for. However, 
in the RCA method and in Barfield’s method, 
the response of the antenna system will not be 
the same as for an ionospheric wave arriving 
at the same angle of elevation. The magnitude 
of this effect increases as the distance to the 
transmitter decreases and the angle of eleva¬ 
tion decreases; it can be very large for the 
usual experimental setup. If it is assumed that 
the surface wave is negligible when it has an 
intensity less than 1 per cent of the space wave, 
then, considering the parallel electric field radi¬ 


ated by a vertical electric dipole, it is found 
that transmissions designed to simulate iono¬ 
spheric wave transmission must be made from 
a distance of the order of 2a when ip = 45°, 
50a when ip = 15°, and 500a when ip = 5°. 

The practical importance of the surface 
wave component for polarization measurements 
using elevated transmitters can be illustrated 
by the experience of the RCA group. In the 
RCA method, the maximum response of the 
antenna system to the parallel polarized field 
and to the perpendicularly polarized field was 
measured, the ratio being V p /V„. Clearly the 
pickup ratio h/k can be determined from this 
measurement at the angle if/, just as the re¬ 
sponse at the angle ip can be calculated from the 
measured values of the pickup ratio. If the re¬ 
sponse of the antenna system to E P)X is neg¬ 
lected it follows that 

V v = hE p>z cos ip (40) 

V n = kE n (41) 

since the maximum output voltages are meas¬ 
ured. The total field components will be propor¬ 
tional to the corresponding field components in 


the incident wave E p>d and E n>d . Thus 

V p = hE p>d f p cos if/ (42) 

Vn = kE n,dfn» (43) 

Here the functions f p and f n are given simply 
by the laws of plane-wave reflection when 
ionospheric waves are considered. Thus 

f p = cos if/ [l+#p e ( * /x) sin *] (44) 

f n = i + R n e sin * (45) 


Putting the proper values of the parameters in 
these equations and using the measured value 
of V p /V n , the pickup ratio can be solved for, 
giving 

h _ V p E n , d f n 1 

k ~ V n E n>d f p cos if/ 

In this manner RCA determined h/k for values 
of ip from near zero up to almost 45°. The pick¬ 
up ratios thus found were constant for large 
values of if/ but much greater at low angles 
than at high. However, when the functions f p 
and f n were computed using the surface wave 
component as well as the space wave, the pick¬ 
up ratios thus determined were constant for all 
values of ip. Accordingly, the field did not simu¬ 
late that of an ionospheric wave until elevation 




40 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


angles of 20° to 30° were reached. If the trans¬ 
mitter were moved further from the direction 
finder the surface wave would have been re¬ 
duced since it is attenuated faster than the 
space wave. 


TRANSMITTING 



Figure 7. Elevated dipole transmitting over 
finitely conducting ground. 


Field Generated by Vertical and Horizon¬ 
tal Electric and Magnetic Dipoles 

For purposes of reference and for use in the 
next section of this report the complete equa¬ 
tions are given for the field from vertical and 
horizontal electric and magnetic dipoles at dis¬ 
tances, d, greater than the wavelength. 5 These 
expressions refer to dipoles transmitting over 



a finitely conducting ground as shown in Fig¬ 
ures 1 and 7 and include the surface wave 
component. E oe and H oe are the values of the 
electric and magnetic radiation fields at a unit 
distance in free space in the equatorial plane 
of the electric dipole, while E om and H om are 
the corresponding values for a magnetic di¬ 
pole. The d-k plane is taken as the plane of 
incidence with k vertical; Figure 7 defines 
r u r 2 , i j/ lt and if/ 2 and the unit vectors 


iji = cos if / 1 k + sin i/^d, 

(47) 

i \ 2 = cos if / 2 k — sin if/ 2 d t 

(48) 

r 1 = cos ^jd — sin xf / 1 k, 

(49) 

r 2 = cos »/r 2 d + sin ^ 2 k. 

(50) 

expressions for the fields are 

(here k = 


27r/A) I 


E = lEoe ]cOS xpi - tjri + COS \f /2 R P - t|f: 

^ r 2 


oikr 2 


[■ 


H, 


cos i/^k + 

iH oe8 jcos if/ 


+ COS if /2 (1 — Rp)f(Pe,Be). 

V n 2 — cos 2 If/ 2 J~\ e ^ + kr2 ) \ 

n 2 J r 2 j ’ 


e ikri . . j-j e 

-- -(- cos if /2 Rp- 


r 2 

p i(<t>kr 2 ) \ 

+ cos ^ (1 - R P )m,B e ) 


> (* 


E„ = E„ m 6 | 

+ COS 

H P = H 


£> IKT 1 gftA //2 

COS Ip 1 -b COS if/ 2 Rn - 

r i r 2 

x f / 2 (1 Rn)f(Pm,Bm ) ~ [ > ( 

f g ikn pikn 

-\COS if/i - ih + COS if / 2 --t|f 2 

( r j t 2 

-|“ COS if/ 2 (1 — Rn)f{Pm,Bm )• 

[ _ *1 g *(0 + ^ 2 ) I 

cos ^ 2 k + V n 2 — cos 2 if/ 2 d I- —— b . ( 


In this case E n = U v = 0 for the vertical elec¬ 
tric dipole while E p = H„ = 0 for the vertical 
magnetic dipole. 

Horizontal Electric and Magnetic Dipoles. 

Figure 8 illustrates these cases, the electric 
dipole, and axis of the magnetic dipole (loop) 
pointing in the positive i direction. 

/ pikn pikr 2 

E« = iE oe sin 00]-h R n - 

l r i 1 2 

e i(<f>+kr 2 ) ) 

+ (1 — Rn)f(Pm,B m ) — j , 


Figure 8. Case of horizontal electric and mag¬ 
netic dipoles. 


( 55 ) 



























MEASUREMENT OF POLARIZATION ERROR 


41 


f p ikr i p ikn 

Ep = iE oe cos 0 ]sin xj/ 1 -+ sin \J/ 2 R P - ijj 2 cos xf/ 2 k + 

( T i T 2 


+ — ~ ^ (1 - Rp)f(Pe,B e ). 

[ , » i Vn 2 — cos 2 \f /2 1 ~\ e *(0+* r «)) 
cos^ 2 k +-^-dj 




n L 


J—h 




( p ikn . 

E« = Eom COS 00 \ sin \J /1 - + Sin i p 2 Rn 

l r 1 


, ikrz 
Ti 


(56) 


- p i(0+&r 2 )) . N 

> 2 ^2 (1 - Rn)f(P m ,B m ) e - -K (60) 

r 2 J 


f p ikn p ikn 

Hp = iH oe sin 0 ] -tjfi ~h R n - ^2 

( r 1 r 2 

+ (1 — Rn)f(Pm,Bm)- 


COS 


+ -\/ n 2 — cos 

l p ikr 1 p ikn 

Hm = Horn sin 00 - iH -f- sin \l/ 2 Rn — 

( 1*1 r r 2 

+ Vn 2 - cos 2 ^2 (1 -R n )f(P m ,B m )- 

i(<f>+kn ) | 


1&2 


, , . /-t -— r -- 1 piW+kn)) 

^ok + vn —cos 2 ^ 2 dj——— j, (57) cos xf/ 2 k + \/n 2 —cos 2 xJ/ 2 d — -/. (61) 


( e *^ ri 

Hn = —iH oe cos 00] sin xf/i -(- sin if/ 2 

‘ T 0 


R P 


pikn 


+ — J° S2 ^ (1 - Rp)f(Pe,B e ) 


r 2 

p i(<f>+kn) J 


n* 


r 2 


7 


,(58) 


En = Eom sin 0 


> ikn 


fl 


tfci + R p 


? ifcr2 


r 2 


t|f2 


+ (1 - Rv)f{Pe,B e )- 


In these equations, R p and R n are the plane 
wave reflection coefficients as already defined. 
The third term in each equation represents the 
surface wave and f(P,B)e i(f> is the surface wave 
attenuation function which is given graphically 
as a function of P and B in Figures 9 and 10. 
Here the angle <f> is the phase of the surface 
wave attenuation function (not to be confused 
with the azimuth angle). 



Figure 9. Surface wave attenuation function. 

















































42 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


f(P,B) e** = 1 + i v / iPie- p 'Erfc (—lV^i), 


P e e' B ‘ = 
P m e iBm = 


ikr 2 


2 cos 2 \p 2 [ 
ikr 2 

2 cos 2 ^2 


Px = Pe iB , 

V^ 2 — cos 2 \p 2 


sin \f/ 2 +- 


]: 


(62)“ 

(63) 

(64) 


^sin^ 2 + Vn 2 — cos 2 ^2 J. (65) 



NUMERICAL DISTANCE P 

Figure 10. Phase of surface wave attenuation 
function. 

P e and P m are called numerical distances. The 
preceding expressions for the field from a local 
radiator reduce to the values given for plane 
waves when r is allowed to increase without 
limit. 

To indicate the magnitude of the surface 
wave, Figure 11 shows the ratio of surface to 
space wave intensities at the surface of the 
earth radiated from a vertical electric dipole 
at a height a. It is seen that the surface wave 
falls off with increasing distance r and increas¬ 
ing elevation angle if/. For a local transmitter 
at a distance of one wavelength, very large val¬ 
ues of if/ are needed to simulate a downcoming 
plane wave. 


‘ Erfc(;c) represents the so-called error function. 11 - 14 


23 5 Radiator Parallax 

An examination of equation (56) for the 
field from a local horizontal electric dipole 
transmitter reveals that E p ^0 except in the 
equatorial plane where cos 0 = 0. When such a 
radiator is used for determinations of polari¬ 
zation errors there will be E p components at 
the d-f antenna elements since these will lie on 
either side of the equatorial plane of the trans¬ 
mitting dipole. Furthermore the phase of the 
fields at the two antenna elements will be op¬ 
posite so that the induced voltages will add up 
in the output of the system, as a result of the 
differential connection of the direction finder, 
causing an error which will be called “radiator 
parallax.” The response of the antenna system 
to these E v components is not desired when 
using such a horizontal dipole, since the re¬ 
sponse to E n alone must be measured to make 
possible an accurate determination of the cor¬ 
responding pickup factor. 

These undesired parallel components will be 
present for both horizontal incidence and ele¬ 
vated transmitter methods of measuring polar¬ 
ization error. The presence of the ground is 
responsible for this state of affairs in the case 
of the horizontal incidence method where the 
transmitting and receiving antennas are at the 
same height above ground, since there is no E p>s 
component in the direct wave (sin if/ 1 = 0) 
while there is such a component in the direct 
wave from an elevated transmitter. The sig¬ 
nificance of the undesired E p component in 
direction-finder testing was first pointed out by 
W. H. Wirkler of the Collins Radio Company. 
This component is important because, although 
small compared to E n , it is not attenuated so 
rapidly and so, at large distances from the 
transmitter, it becomes relatively large enough 
to render the measurements inaccurate. This 
holds especially for direction finders having 
low values of polarization error. If E n is gen¬ 
erated by means of a vertical magnetic dipole 
(horizontal loop-antenna), E n = 0 so that this 
error is avoided. 

When a local transmitter is used to test a 
spaced vertical loop-antenna direction finder 
for response to E ny there will be unwanted H n 
components at the two loop antennas if the 
transmitter uses a horizontal electric dipole. 








































































MEASUREMENT OF POLARIZATION ERROR 


43 


This case is similar to the one just discussed. 
This result is seen in equation (58) which 
shows that the phase of H n will be opposite at 
the two loops, thus causing an apparent re¬ 
sponse to E n which is misleading. Here also 
the remedy is to use a horizontal loop antenna 
in the transmitter. 

Situations similar to that just described arise 


23,6 Collector Parallax 

Another error which occurs with local trans¬ 
mitters was also pointed out by the Collins 
Radio Company. This error, called parallax 
error, occurs when measurements of polariza¬ 
tion error are made on spaced, vertical loop- 
antenna direction finders. 



1 2 3 4 6 8 10 20 30 40 60 80 100 200 300 400 600 

r/\ 


Figure 11. Ratio of surface to space wave intensities at surface of earth radiated from vertical electric dipole 
at height a for frequencies in megacycles for which curves are labeled. 


when generating a wave polarized parallel to 
the plane of incidence. This may also be seen 
from an examination of the preceding equa¬ 
tions (60) and (61). Accordingly the general 
rule follows that a local transmitter using a 
vertical electric dipole should be used for gen¬ 
erating the wave polarized parallel to the plane 
of incidence and one using a horizontal loop 
antenna for generating the wave polarized per¬ 
pendicular to the plane of incidence. 


When the response of the antenna system to 
E n is tested, the forward tilt of the H p compo¬ 
nent of the field results in pickup in the loop 
antennas which is not balanced out. This effect 
can be seen by an inspection of equations (57) 
and (54) which shows an H p>x component of 
the field, whether a horizontal electric dipole or 
a horizontal loop antenna is used in transmis¬ 
sion. The H p>x component induces voltage in the 
loop antennas because of the finite “parallax” 





































44 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


angle subtended by the direction finder at th.e 
transmitter. This magnetic field component is 
not parallel to the plane of the H lhX loop antennas 
because the loop antennas lie on either side of 
the line connecting the center of the direction 
finder and the center of the transmitter. Fur¬ 
thermore, the pickup in each loop antenna is of 
opposite phase so that the induced voltages add 
up. 

2 4 EXPERIMENTAL TECHNIQUE 

One of the principal results of this work 
was the development of a method for measur¬ 
ing polarization errors and of experimental 
techniques to use the method. Representative 
direction finders were examined by these tech¬ 
niques, it being necessary to find means of 
overcoming experimental difficulties when the 
method was applied to particular d-f systems. 

The d-f systems examined were: (1) an ex¬ 
perimental rotatable, balanced H antenna built 
and installed at Laurel, Md.—a direction finder 
using unshielded, horizontal feeders but other¬ 
wise practically the same as a Navy DY also 
installed at Laurel; (2) SCR-551-T1 installed 
at Fort Monmouth, N. J.; (3) a Western Elec¬ 
tric [WE] direction finder developed for the 
Civil Aeronautics Administration [CAA] and 
located at LaGuardia Airport, New York City 
—a ten-frequency, fixed, tuned, Adcock ar¬ 
rangement using balanced H antennas; (4) 
an elevated, rotatable, spaced loop-antenna sys¬ 
tem procured from United Air Lines and in¬ 
stalled at Laurel, Md.—the system examined 
with the antennas in both a vertical and a 
horizontal position; (5) a Collins CXAL di¬ 
rection finder measured at Cedar Rapids, Iowa. 

The general procedure will be given here, 
special procedures being described in the sec¬ 
tion to which they apply. la In all cases except 
the Collins measurements, a local transmitter 
employing an electric dipole was used to gen¬ 
erate the desired fields. Waves at horizontal in¬ 
cidence were used; first a wave polarized par¬ 
allel to the plane of incidence, then one polar¬ 
ized perpendicular to the plane of incidence. 
The error introduced by using a horizontal elec¬ 
tric dipole rather than a horizontal loop an¬ 
tenna was not fully appreciated at the time the 
measurements were made. However, it seems 


that the conclusions drawn from the measure¬ 
ments would not be materially altered since the 
polarization errors found were quite large ex¬ 
cept for the loop-antenna direction finders. The 
measurements of the Collins installation were 
carried out by means of a horizontal loop- 
antenna radiator. Also radiator parallax was 
negligible in the case of the measurements of 
the spaced horizontal loop-antenna system. 

The results of some preliminary calculations 
on radiator parallax using the exact equations 
already given indicate agreement with the few 
experimental observations so far available. The 
errors caused by radiator parallax can be con¬ 
siderable for both high and low elevations of 
the transmitter. The error decreases with in¬ 
creasing frequency. 

The adjustment of the horizontal dipole used 
for transmission was very critical. Any slight 
tilt from the horizontal gave rise to a vertical 
field component which assumed large relative 
proportions. The rapid attenuation of the hori¬ 
zontal component was responsible for this. It 
was found that the presence of personnel near 
the transmitter also caused the generation of 
undue amounts of unwanted vertical field com¬ 
ponent. These difficulties were solved by ar¬ 
ranging to control the tilt of the dipole by 
means of cords. For the case of the balanced 
H antenna, the antenna was oriented for maxi¬ 
mum response to E p and the transmitting dipole 
rotated until minimum output was indicated. 
The transmitting dipole was then exactly hori¬ 
zontal ; the purity of field at the direction finder 
was thus determined by the direction finder 
itself in the equatorial plane of the transmit¬ 
ting dipole. 

The output of the antenna system was meas¬ 
ured by substituting a standard voltage gen¬ 
erator for the antenna and determining the 
voltage required to give the same output as 
obtained with the antenna. 

The measurement of field intensity was sub¬ 
ject to inaccuracies resulting from the presence 
of the direction finder. However, when the an¬ 
tenna elements were disconnected, the effect on 
the field intensity was reduced to a negligible 
value in most cases as shown by the effect of 
rotating the direction finder. Before discon¬ 
necting the antenna elements, the aparent field 
intensity varied greatly as the direction finder 



EXPERIMENTAL TECHNIQUE 


45 


was rotated, but this variation become negligi¬ 
ble after disconnecting the antennas. For the 
rotatable H antennas it was found that dis¬ 
connecting the dipoles was not necessary if the 
field was measured when the direction finder 
was properly oriented. For measuring E n , this 
orientation corresponded to minimum response 
to E n . By orienting the direction finder so as 
not to respond to E n , it could not pick up and 
reradiate fields which would disturb the meas¬ 
urement of the field intensity. Such a proce¬ 
dure could not be used in the case of the spaced 
vertical and horizontal loop-antenna systems. 
For these cases the field was measured some 
distance to the side of the direction finder but 
the same distance from the transmitter. The 
assumption was then made that the attenuation 
of the wave was the same for this path as for 
the path to the direction finder. 

When using a field-intensity meter employ¬ 
ing a loop antenna, the electric field calibration 
made with plane waves does not hold when 
measurements are made in other than plane- 
wave fields, although a magnetic field calibration 
would. The field generated by the local trans¬ 
mitter is not the same as for a plane wave, the 
ground-reflected and surface waves being pres¬ 
ent as well as the direct wave. Under these 
conditions, to measure E PiZ and E n with the 
loop-antenna field-intensity meter, 5 the loop an¬ 
tenna of the field-intensity meter is oriented so 
as to respond to either H n or H p>z . The reading 
of the field-intensity meter, which will be in 
volts per meter, then refers to the related value 
of E p>z or E n which would be present if the 
field measured were that of a plane wave in 
free space. This is what is meant, in this re¬ 
port, when H is measured by means of a loop- 
antenna field-intensity meter and designated as 
volts per meter ; it is just the related value of 
E for a plane wave in free space. The reading 
of the field-intensity meter, though in volts per 
meter, will be a number proportional to H. The 
relation between E and H is given by the fol¬ 
lowing equations: 


Hn — — 

H P ,z = 


H oeEp' 


Eoe COS ’ 

H omEn COS \p 
Earn 


( 66 ) 

(67) 


Here E oe and H oe are the values of the electric 
and magnetic radiation fields at a unit distance 
in free space in the equatorial plane of the 
electric dipole, while E om and H om are the corre¬ 
sponding values for a magnetic dipole. 

Equation (67) is important because it shows 
that the loop antenna of the field-intensity 
meter should be oriented to respond to H P)Z 
when measuring E n , rather than for maximum 
reading of the meter as is done for plane waves 
in free space. The maximum reading corre¬ 
sponds to the amplitude of H,, which is often 
much larger than H p>z , so that too large a value 
for E n would be obtained. Since cos ^ is al¬ 
most unity for these measurements it follows 
that E P)Z equals the reading of the meter when 
oriented to H n , and E n equals the reading when 
oriented to respond to H p>z . In most cases in 
this work the measurements of E n were made 
by orienting the meter for the amplitude of H p 
rather than H p>z , since the correct procedure 
was not evolved until after most of the experi¬ 
ments were performed. This renders the meas¬ 
ured values of E n inaccurate for frequencies 
below about 7.5 me where the wave tilt is ap¬ 
preciable. As already stated, the direction of 
this effect is such as to make the measured 
values of E n larger than they actually are. 
Consequently the measured pickup factors, k, 
corresponding to E n were too small and the 
calculated polarization errors also too small. 
This error was not made in the case of the 
United Air Lines [UAL] direction finder when 
used either with horizontal or vertical loop 
antennas. 

After measuring the pickup factors of a di¬ 
rection finder the polarization errors were cal¬ 
culated using the method already given. In 
these calculations the response of the antenna 
system to E p>x involves an unknown phase 
angle A. An inspection of equation (27) for 
the balanced H-antenna system shows that the 
limits on the values of the maximum polariza¬ 
tion error set by the unknown value of A are 
given by 


tan e = 


_ kEn _ 

hE Pi z cos \f/ rb kEp, x ' 


( 68 ) 


These limits can therefore be calculated. How¬ 
ever, since E p>x varies as sin this unknown 
term will be small for low angles of elevation 






46 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


and also for low values of k. For a direction 
finder having low polarization error, k is small 
so that the following approximate equation is 
valid: 


tan e = 


kE n 

hE ViZ cos ^ 


(69) 


This is the equation used in practice for the 
calculation of polarization errors in this re¬ 
port, except for the loop-antenna direction 
finders. If k is so large that equation (69) is 
not a good approximation, the exact value of 
the polarization error is not needed because it 
will be very large. The equation can be used to 
indicate the superior or inferior performance 
of a direction finder. 

In the results that follow, polarization errors 
are calculated for average ground conditions, 
that is, a conductivity a = 5 X 10 14 emu and 
dielectric constant K = 15. 


2 41 Test Conclusions 

This experimental technique evolved as the 
research progressed and included special means 
to overcome experimental difficulties encoun¬ 
tered in the application of the method to meas¬ 
urement on particular direction-finder systems. 
(1) The stringent conditions for pure fields 
were relaxed for the case of Adcock direction 
finders by orienting the antenna system to the 
proper null position. (2) Remote control was 
used to control the radiator when generating 
perpendicularly polarized fields, while the di¬ 
rection finder itself was used as an indicator 
to tell when the radiator was exactly horizon¬ 
tal. (3) The influence of the direction finder 
on the measured field intensity was removed 
in many cases by properly orienting the direc¬ 
tion finder. In other cases measurements of 
field intensity were made off to one side of the 


direction finder. (4) Correction factors were 
computed and applied to allow measurement of 
electric field intensity by means of a field-in¬ 
tensity meter using a loop antenna. (5) To 
reduce errors caused by radiator parallax, a 
horizontal loop-antenna radiator was found 
necessary when generating the perpendicularly 
polarized wave field required in this work, 
while a vertical electric antenna was necessary 
for generating the wave polarized parallel to 
the plane of incidence. (6) Methods were de¬ 
veloped for reducing the experimental errors 
caused by collector parallax for the case of 
particular direction finders. 

The experimental technique finally evolved 
has important application to other methods of 
measuring polarization error. The errors 
caused by radiator and collector parallax will 
be present in the RCA and Barfield methods 
unless techniques similar to those used in this 
work are applied. Furthermore these methods 
encounter another difficulty, clarified by the 
work of this report, when measurements are 
made at angles of elevation below 20° to 30°. 
This is the error caused by the surface wave 
component of the field from the local trans¬ 
mitter and is not present in the NBS method. 

Since the experimental technique evolved 
gradually, the measurements of polarization er¬ 
rors of the various direction finders were not 
all carried out with the same accuracy. In some 
cases E n was incorrectly measured, giving 
measured polarization errors which were too 
small. This effect was unimportant above 7.5 
me. In other cases, radiator parallax was not 
avoided. Bearing in mind these limitations as 
to accuracy, Table 2 of approximate polariza¬ 
tion errors compiled from these sections may 
still be used to draw certain important con¬ 
clusions. 


Table 2. Approximate polarization errors. 


Direction finder 

2.5 me 

5.0 me 

7.5 me 

10 me 

12.5 me 

15 me 

eo 

£45 

£o 

£45 

£o 

£45 

£0 

£45 

£0 

£45 

£0 £45 

WE—CAA. 

41 

45 

15 

37 

8 

30 






Experimental H antenna. 

21 

20 

13 

23 

12 

36 

4 

20 

11.7 

51 

5 11 

SCR-551. 

22 

12 

12 

14 

11 

20 






Navy DY. 

12 

7 

7 

7 

4 

5 

9 

18 

7.8 

20 

19 50 

Collins CXAL. 

4.5 


6.5 


6.5 


5.5 


3.2 


0.7 .... 

Vertical loop antennas. 



16 

9 

10 

7 

1.2 

1.1 

1.0 

1.1 

1.1 1.6 

Horizontal loop antennas. 

1.8 

3 

1.9 

1 

1.9 

1 

1.8 

0.8 

1.6 

0.8 

2 1.8 
























EXPERIMENTAL TECHNIQUE 


47 


In Table 2 the values are given of two par¬ 
ticularly significant “wave errors” which may 
be derived to represent the performance of a 
given direction finder: (1) the value of maxi¬ 
mum bearing error for a downcoming wave 
incident at 45° with equal parallel and perpen¬ 
dicular components, e 45 , which is the “standard 
wave error” as defined by Barfield and (2) 
the value of maximum bearing error for a hori¬ 
zontally traveling wave also with equal parallel 
and perpendicular components, e 0 . The error 
e 4r , includes the effect of the height of the direc¬ 
tion finder antenna above ground and of the 
electrical properties of the ground whereas 
e 0 is independent of these effects. 

Table 2 gives the horizontal wave error, 
£q, and the standard wave error, c 45 , in de¬ 
grees. The value of e 0 is independent of the 
ground constants as already stated, while that 
of e 45 is for average ground having constants 
K = 15 and a — 5 X 10 -14 emu. The height of 
the various direction finders for which the 
values of e 45 are given is the height for which 
the direction finders were designed except in 
the case of the UAL antenna system. The 
height was taken as 10 feet for the vertical 
loop-antenna system as a practical value ap¬ 
proaching optimum results. Two different 
heights were taken for the horizontal loop- 
antenna system; 50 feet over the band from 
2.5 to 7.5 me and 30 feet above 7.5 me. In this 
way the whole frequency range was divided 
into two bands with approximately optimum 
antenna heights for each band. The values of 
c 45 for the experimental H antenna, the 
SCR-551 and the WE-CAA direction finders 
would have been lower for lower antenna 
heights. The complete data for each system are 
summed up in the graphs given in the particu¬ 
lar section for that direction finder in the final 
report. 1 ’ 19-26 

The data given in Table 2 are shown in graph¬ 
ical form in Figures 12 and 13 in which e 0 and 
e 45 are respectively plotted as a function of fre¬ 
quency for each of the direction finders. 

Inspection of Table 2 and Figures 12 and 13 
shows that the polarization errors are in gen¬ 
eral much larger for those direction finders 
using open antenna elements than for those 
using loop antennas. This indicates that the 
loop antennas are inherently easier to balance 



0 2 4 6 8 10 12 14 16 

FREQUENCY IN MEGACYCLES / SEC 

Figure 12. e G versus frequency for direction 
finders tested. 



0 2 4 6 8 10 12 14 16 

FREQUENCY IN MEGACYCLES/SEC 


Figure 13. Maximum polarization error for equal 
plane wave components parallel and normal to 
plane of incidence. 





















































48 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


and to shield properly so as to suppress un¬ 
wanted pickup. The low impedance of the loop 
antennas is helpful on this score. This conclu¬ 
sion as to the relative superiority of the spaced 
loop-antenna direction finders is one of the 
principal results of this work. 

Table 2 and Figures 12 and 13 also indicate 
that the direction finder having the least polar¬ 
ization errors of all those tested was the one 
using horizontal loop antennas. This superior 
performance was obtained without any critical 
adjustments of the antenna system. Measure¬ 
ments on this antenna system were free from 
radiator parallax errors and from errors of 
measurement of field intensity. Coupled with the 
low polarization error of this system is its low 
susceptibility to site errors. These two proper¬ 
ties together would indicate a very promising 
system for those applications where sky waves 
only are being received and where the antenna 
may be placed approximately one-fourth wave¬ 
length above the ground. 

Methods similar to those given in this report 
were used by RCA and International Telephone 
and Radio [IT & R] Laboratories to make polar¬ 
ization error measurements on the Bell Tele¬ 
phone Laboratories [BTL] buried U-antenna 
system at Holmdel, on the elevated, shielded 
U-antenna system of RCA, and on various 
IT & R systems at Great River, Long Island. It 
was found that the BTL buried U antenna had 
a performance similar to the loop-antenna sys¬ 
tems described herein; the RCA antenna sys¬ 
tem had errors about the same as those of the 
electric antenna systems of this report, while 
IT & R reported an increase in accuracy for 
electric antenna systems achieved by using 
cathode followers to couple the antenna ele¬ 
ments to the transmission line. 

Critical consideration of the NBS method of 
measuring polarization errors as applied to the 
several direction finders shows that the method 
has several advantages. First may be men¬ 
tioned the convenience and speed with which 
measurements are made since, in general, all 
measurements are made with the equipment 
near the ground and because waves polarized 
parallel and perpendicular respectively to the 
plane of incidence are used separately. This 
avoids the need for adjusting the phase of these 
two components as is necessary when both 


waves are used simultaneously. The fact that 
these waves are used separately also results 
in another important advantage, namely, that 
maximum polarization errors are measured. 

This result is one of the principal results of 
the present research. Originally, Barfield de¬ 
fined the ‘‘standard wave error” to be the bear¬ 
ing error for the “standard wave” with such 
phase relation between the parallel and perpen¬ 
dicular wave components as to result in maxi¬ 
mum error. However, the experimental tech¬ 
nique employed by Barfield for many years did 
not meet the conditions required for maximum 
error; as a result the polarization errors meas¬ 
ured were much too low. The publication of 
polarization errors measured by this method 
led to the general belief that polarization er¬ 
rors were quite small. Measurements by the 
NBS method and subsequently by that of RCA 
indicated much larger polarization errors for 
existing direction finders than had generally 
been believed to be the case. 

In the Barfield technique maximum errors 
were not, in general, measured because the 
phase relation of the two components in the 
wave from the target transmitter was not ad¬ 
justed for maximum error. Some months after 
first publication of the NBS method, an account 
appeared of recent attempts to modify the Bar- 
field method so as to control the phase of the 
two components 15 but these had not yet been 
applied practically because of experimental dif¬ 
ficulties. However, further measurements 16 of 
an H-antenna system, 17 in which allowance was 
made for the proper phase relation to give 
maximum error, showecf errors two to ten 
times larger than those values given previously 
on the basis of measurements by the Barfield 
method. This result as to the extraordinarily 
large polarization errors of many present types 
of direction finders now agrees with that of the 
NBS and the RCA groups. 

The large polarization errors found as a re¬ 
sult of this work have refocused attention on 
the reduction of polarization errors. The NBS 
method has an important application to this 
problem of the reduction of polarization errors 
since it furnishes a figure of merit by which 
the progress of development work may be 
judged. After each change in design the pickup 
ratio of the antenna system may be measured 



DIRECTION-FINDER SITE PROBLEMS 


49 


in order to determine the effect of the change 
on the polarization error. The technique is 
rapid and accurate. 

The figure of merit proposed by the NBS and 
measured by the methods given in this report 
is the pickup ratio, h/k, of the direction finder. 
A practically equivalent figure of merit is the 
horizontal wave error, e 0 , as previously defined. 
The equation tan e 0 = k/h is the usual one 
given and is a means of translating the pick¬ 
up ratio into an actual bearing error for an 
incident wave. The pickup ratio allows a direct 
comparison of polarization error for all direc¬ 
tion finders following the same equation for 
polarization error and working on the same 
field components. The complete curve of polari¬ 
zation error versus angle of elevation of the 
incident wave should be used to compare the 
accuracy of antenna systems following differ¬ 
ent laws for polarization error. The pickup 
ratio is especially valuable for comparing the 
accuracy of direction finders because it is a 
fundamental d-f constant which is independent 
of the ground constants and the height of the 
antenna above the ground. In the case of buried 
U-antenna systems this constant is indepen¬ 
dent of the depth of the feeders below the 
ground even though the accuracy may be 
greater when the depth is increased, just as the 
accuracy of those systems above the ground, 
which are designed to suppress response to E n , 
is increased by lowering the height of the an¬ 
tenna. Once the pickup ratio is measured, the 
polarization errors for any downcoming wave, 
such as the “standard wave,” may be calcu¬ 
lated for any antenna height or ground con¬ 
stants. This enables a study to be made of the 
optimum antenna height and ground constants 
for lowest polarization errors. 

On the basis of such studies it was shown 
that the polarization error of a direction finder 
designed to utilize the E v component of the in¬ 
cident wave and to suppress response to E n 
components should be located over ground hav¬ 
ing the highest possible index of refraction. 
The choice of such a site requires methods of 
measuring the ground constants of proposed 
sites. A method was developed for this meas¬ 
urement which uses a field-intensity meter hav¬ 
ing a loop antenna. This method is easy to use 
and uncritical in its application. By measuring 


the ground constants at various points of the 
site, as indicated below, a test can be made of 
the subsurface homogeneity of the site and 
therefore its suitability from the standpoint of 
local site errors. Such methods and tests are 
becoming of greater importance because of the 
improved accuracy of newer types of direction 
finders. It is possible that polarization errors 
may eventually be reduced to the point where 
the bearing errors caused by the site may be of 
relatively greater importance. In this respect 
it may be important to use direction finders 
having inherently lower susceptibility to site 
errors caused by local reradiation. This re¬ 
search has shown that the spaced, horizontal 
loop-antenna direction finder should be rela¬ 
tively insensitive to reradiation errors because 
of the rapid attenuation by the ground of the 
horizontally polarized fields reradiated by sur¬ 
rounding objects. 

2.5 DIRECTION-FINDER SITE PROBLEMS 

The problems connected with d-f sites are 
numerous and complex. They may be classified, 
broadly, into two groups. The first group con¬ 
cerns the bearing errors caused by the site it¬ 
self, that is, errors caused by deviation of the 
wave front or by reradiation. The second group 
concerns the effect of the site on (1) the direc¬ 
tion finder or (2) the principal field at the di¬ 
rection finder. By (1) is meant the effect of 
the site in unbalancing the d-f antenna system, 
while by (2) is meant the effect of the site in 
suppressing undesired field components because 
of the interference between the direct and 
ground reflected waves. Furthermore, site er¬ 
rors can be classified as local or remote depend¬ 
ing upon the distance of the source of the site 
error from the direction finder. Corresponding 
to this division into groups, the following dis¬ 
cussion will take up the problems connected 
with the choice of a direction finder having the 
lowest susceptibility to site errors caused by 
reradiation, and those connected with the 
choice and testing of a suitable site. These site 
problems have recently assumed greater impor¬ 
tance as a result of the improved accuracy of 
newer types of direction finders. It is possible 
that, excluding errors caused by lateral devia¬ 
tion, d-f accuracy will no longer be limited by 




50 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


type, is believed to be relatively free from local 
site error. 

2,5,1 Site Errors 

Bearing errors caused by imperfections of 
the site have been classified as local or remote, 
although there is no sharp dividing line be¬ 
tween these two groups. Remote site errors are 
usually caused by the character of the terrain. 
Large obstacles in the path of the wave, such 
as mountains, give rise to diffraction which 
results in a deviation of the wave front. Local 
site errors are caused by reradiation or reflec¬ 
tion from nearby trees, wires, cliffs, etc. The 
random summation at the direction finder of all 
the reradiated waves gives a field which results 
in bearing errors. This distorting field can 
change rapidly with small changes of azimuth 
or frequency of the incoming wave. 18 


type, is believed to be relatively free from local 
site error. 

This direction finder is designed to take a 
bearing on the E n component of the incident 
field while ideally it should have no response 
to E p . In general, reradiated E n field compo¬ 
nents will be so severely attenuated by absorp¬ 
tion in the ground that the reradiated E n field 
intensity at the site will usually be very small. 
This is true even if the spaced horizontal loop- 
antenna direction finder is used at its optimum 
height a/ 4 above ground. Figures 4 and 5 show 
that E n will in general be equal to or greater 
than the perpendicular component in the inci¬ 
dent wave, E n ,a• At this height, therefore, the 
effect of ground reflection on E n will not be one 
of suppression. However, the field intensity at 
smaller heights will be decreased by the ground 
reflection so that reradiating objects at these 
smaller heights will have their effectiveness as 



Figure 14. Absorption of plane radio waves in earth. Solid lines, land; dotted lines, water. 


Local site errors can be reduced by choosing 
a clear, flat homogeneous tract. However, it is 
also clear that certain types of direction finders 
will be less susceptible to local site errors than 
others. The spaced, horizontal loop-antenna di¬ 
rection finder, either of the fixed or rotatable 


sources of site error reduced. Furthermore, for 
reradiating sources at a distance of approxi¬ 
mately 300 feet, the E n component of the 
field will be attenuated two to ten times as 
much as reradiated E p components. Therefore 
in comparing the horizontal loop-antenna di- 



















































A METHOD OF MEASURING GROUND CONSTANTS 


51 


rection finder with those types designed to re¬ 
spond to the E p component, it is clear that for 
equal field intensities in a downcoming wave, 
the total desired field will be approximately 
the same for the two types (used at optimum 
heights), while reradiated field intensities ca¬ 
pable of causing bearing errors will be much 
less for the horizontal loop-antenna system. 
This comparison is made on the basis of equal 
amounts of energy reradiated from the dis¬ 
turbing source for the two cases. 


252 Required Depth for Buried Cables 

It is very convenient in many direction find¬ 
ers to be able to run cables, power lines, or tele¬ 
phone lines near the d-f antenna system. To 
avoid site errors caused by reradiation from 
such lines it is necessary to bury them an ade¬ 
quate distance below the surface of the ground. 
Figure 14 shows the absorption of plane radio 
waves in earth for various ground constants 
and for frequencies up to 1,000 me. The dis¬ 
tance in feet required for an attenuation of ten 
to one is shown on this diagram. It is interesting 
to note that the ultra-high frequencies are ab¬ 
sorbed only slightly more than frequencies in 
the standard broadcast band in passing 
through media of average conductivity. 

The field intensity at a depth A below the 
ground for a plane wave incident on the sur¬ 
face will be determined not only by the absorp¬ 
tion of the wave but by the reflection at the 
surface. Previously, equations were given for 
the total field intensity at a height z above the 
surface. The field intensity at a depth A directly 
below the field point considered previously will 
be given by d 


Ep t z,t— ^-(1 -\-Rp) (cos \p) e (2ici'/x) ( zsin <^ + A Vra 2 — cos*<J< ) 

; „ <■«; 

77? Ap'd 

&V,X,t ~ -«T* 

n 2 


(1 -f- Rp ) y/n 2 —COS 2 ^ e (2ici/X) (zsinty + A Vw 2 —cos 2 i]> ) , 

(143) 

En,t=En,d (!+/?«) e (2 *t'/x) (z sin 4* + A Vw 2 -cos 2 ) . (144) 


“Equations (70) to (141) inclusive of the final re¬ 
port 1 are not given in this summary. The original 
equation numbers are retained here, however, for 
ease in referring to the original. 


Here the subscript t indicates the transmitted 
wave. The term 2 ?rz (sin xj/) J\ in the above 
equations relates the phase of the transmitted 
wave to that of the incident wave at the height 
z above the ground. The attenuation factors in 
equations (142) to (144) may be determined 
from Figure 14 by identifying d with A and K 
with K — cos 2 if/. This follows from the fact 
that the attenuation factor of a plane wave is 
given by e~ Ad where 


27 m 

B + 

iA 

(145) 


2t 

l~JT . a 

(146) 

A = 


J -sin—, 

f COS a 2 * 

tan a = 

X 

K' 


(147) 


The absorption coefficient A may be determined 
for the case where d is expressed in feet simply 
by dividing the constant 2.303 by the distance 
in feet as given in Figure 14. 


Table 3. Recommended depth to which under¬ 
ground lines must be buried to avoid reradiation 
errors. 


Ground conductivity 
in emu 

Recommended depth 
for buried lines 

Sandy soil 

10- 14 

100 feet 

2 X 10- 14 

50 feet 

Average land 

5 X 10- 14 

20 feet 

10- 13 

10 feet 

Good conducting land 

2 X 10- 13 

5 feet 

5 X 10- 13 

2 feet 

Sea water 

5 X 10-n 

6 inches 


2 6 A METHOD OF MEASURING GROUND 
CONSTANTS 

In choosing a site, it is desirable to have 
available quick and sensitive methods for test¬ 
ing its suitability without actually setting up 
the equipment and making bearing tests. For 
this purpose, visual observation of the flatness 
and freedom from reradiating objects of the 
proposed site is not sufficient, because the site 
must also be electrically homogeneous below 
the surface and must also have electrical con¬ 
stants falling within certain limits for best 












52 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


results. A downcoming ionospheric wave may 
penetrate a considerable distance into the 
ground so that inhomogeneities located below 
the surface may reflect the waves strongly 
enough to cause bearing errors. To reduce this 
effect a site should be chosen over ground hav¬ 
ing as large an index of refraction as possible, 
such as a salt marsh. However, high conduc¬ 
tivity and dielectric constant are also desirable 
from another standpoint. The site has a strong 
effect on the principal field at the direction 
finder because the total field is the vector sum 
of the incident wave and the ground reflected 
wave. This vector sum is termed the principal 
field in order to exclude fields generated by re¬ 
radiating objects near the site. If the direction 
finder is designed to take a bearing through its 
response to E v field components while its re¬ 
sponse to E n is suppressed, then it is desirable 
to locate the direction finder over a site having 
ground constants such as to suppress E n as 
much as possible. It has been shown in preced¬ 
ing sections that ground having a high index 
of refraction suppresses E n at points near the 
surface. Such ground also helps to screen 
buried lines and cable so that reradiation errors 
are reduced. All these considerations indicate 
that a quick method of testing a proposed site 
for subsurface electrical inhomogeneities and 
of measuring its electrical constants would be 
useful. NBS has considered a method of testing 
for inhomogeneities which uses a local oscil¬ 
lator and antenna near the ground. The varia¬ 
tion over the site of the impedance reflected 
into the oscillator circuit by the ground reflec¬ 
tion gives a test of the homogeneity of the site. 
Either the variations in the plate current of the 
oscillator or of the oscillator frequency could 
be used as an indication of homogeneity in 
these tests. 

An alternative method of testing a site is to 
measure the ground conductivity and dielectric 
constant at various points of the proposed site. 
This method would not only determine the 
ground constants but also give a measure of 
the subsurface homogeneity of the site. Previ¬ 
ous methods of measuring ground constants 
made use of electric antennas with their at¬ 
tendant difficulties. A new method of measur¬ 
ing ground constants by means of a standard 
field intensity set using a loop antenna, such as 


the RCA 308-A, will now be described 5 because 
of its application to the problem here consid¬ 
ered. 

In the previous discussion expressions for 
the fields generated by electric and magnetic 
dipoles near the ground were given for both 
vertical and horizontal dipoles. Equation (54) 
for the magnetic vector for the case of a ra¬ 
diating vertical magnetic dipole can be written 
as follows for the field intensity at the surface 
of the ground (z = 0) ; in this case r t = r 2 = r 
and = xp 2 — xfj and we obtain 
H„ Hom COS if/ • 

[(1 + R n ) + (1 - R n )f(P m , 

--=-r o ibr 

(cos xf/k + \/n 2 “ cos 2 ^d) z — (148) 

r 

where z = 0 and d > X. 


Equation (148) shows that at the surface 
of the ground both the space and surface wave 
components have the same polarization. The 
particular polarization of the magnetic vector 
and its forward tilt will depend on the ground 
constants and, if determined, give a means of 
measuring the ground constants. This is also 
true of the polarization and forward tilt of the 
electric vector E p in the field of a radiating 
vertical electric dipole. However, in this case 
measurements of the electric vector would have 
to be made using electric dipole receiving an¬ 
tennas. Such measurements are difficult to 
make accurately because of disturbances to the 
electric field by the field-intensity meter and 
operating personnel. Therefore the new method 
based on measurements of H p gives a prefer¬ 
able method of determining the ground con¬ 
stants. 

Equation (148) may be written as follows: 

( cos if/ . \ 
d +Vn*-cos*/> K (U9) 

If we write 


cos \f/ 


1 


ae l $ = y /— cos 2 if/ y/ K' — 1 — iX 

where 

K 

COS 2 \p ’ 

X 


K' = 

I 

x'= 


=-,(150) 

(151) 

(152) 


COS 2 ’ 

and K is the dielectric constant of the ground 












A METHOD OF MEASURING GROUND CONSTANTS 


53 


while X = 1.797 X 10 15 o- em u//mc, then equation 
(149) becomes 

U p = H p d [d cos iot + kacos (o >t + /3)]. (153) 

The above equation shows that the vector mag¬ 
netic field from a vertical magnetic dipole ro¬ 
tates in an ellipse in the plane of incidence with 
its major axis tilted a few degrees above the 
horizontal. The magnetic vector reaches its 


Figure 15 shows 0 and r as a function of X ' 
for K' = 5, 10, 20, and 80. 

The procedure for determining the ground 
constants from the-above results is as follows. 
A small transmitter is used with a loop antenna 
which is set up with its axis in the plane of 
incidence at a distance greater than A from the 
point at which the ground constants are to be 
determined and at a height such that an easily 



Figure 15. Polarization of vector magnetic field from vertical magnetic dipole. 


maximum extension when <ot = —8 and its 
minimum extension when o)t = — 8 + (tt/ 2) 
where 


tanS = i(Vl + 4r 2 -l), 

(154) 

a 2 sin 2/3 

T 2 (1 + a 2 cos 2/3) 

(155) 


The measurable properties of the ellipses are 0, 
the tilt of the major axis above the horizontal, 
and r, the ratio of the minor to the major axis. 

tan 6 = a(cos /3 + sin ji tan 8 ), (156) 

r = tan 8 cot 6. (157) 


measurable field intensity is obtained at the 
receiving point. Figure 15 also shows a dia¬ 
gram of the experimental setup. Using a field- 
intensity meter with a loop antenna set up in 
such a manner that it can be rotated about an 
axis perpendicular to the plane of incidence 
but with the loop axis always lying in the plane 
of incidence, measurements are made of 6 and 
r. The loop antenna of the field-intensity meter 
is to be placed as near the ground as possible 
because this procedure is based on equations 
derived for the case z == 0. Having measured 6 
and r, a corresponding set of values of K and 
X' may be determined from the curves given in 
Figure 15. Finally K and a are determined by 






























































54 


NBS HIGH-FREQUENCY DIRECTION-FINDER RESEARCH 


means of equations (151) and (152) as fol- either curve allows a determination of X' and 
lows: therefore of a. For this case 


K = K r cos 2 + (158) 

a = X' cos 2 xp • / mc • 5.564 X 10“ 1G emu. (159) 


At very high frequencies, X' will be small and 
we may write 


K' = 


sin 2 0 


(160) 


X' = 


2 r 

sin 2 6 tan 0 9 


(161) 


where X' << (A' —1). 

Equations (160) and (161) show that the 
dielectric constant K may be determined at very 
high frequencies simply by measuring 0 while 
a determination of o- requires a measurement 
of both 6 and r. This is also evident from Fig¬ 
ure 15. At very low frequencies where X' is 
large, Figure 15 shows that the curves of 0 and 
r are independent of the dielectric constant but 




1 

2 tan 2 6 


2r 2 ’ 


( 162 ) 


where X' << (K' — 1). Equation (162) also 
shows that the dielectric constant can not be 
measured at very low frequencies. However, this 
is no defect of the method since the dielectric 
constant has no appreciable effect on wave prop¬ 
agation at these low frequencies. As shown in 
Figure 15 the measurement of the ratio r of the 
minor to major axis of the polarization ellipse 
can be accurately made by means of a loop 
antenna and field-intensity meter provided only 
that the loop antenna is free from “antenna 
effect.” This may be stated quantitatively as 
follows: the ratio of minimum to maximum 
reading in a linearly polarized magnetic field, 
such as is generated by a vertical electric an¬ 
tenna, must be much less than r. 





Chapter 3 


STUDY OF RADIO PULSE PROPAGATION 


Pulses were transmitted from Puerto Rico and re¬ 
ceived at Holmdel, N. J., on a highly directional Musa 
system. Measurements indicated that direction finding 
on the first pulse of a pulse group gave significantly 
more accurate results than ordinary direction-finding 
methods, a fact of considerable value in long-range 
loran systems. Practically all the contractor’s final 
report 1 is contained in this summary. 

31 OBJECTIVE 

T his project a had as its objective the confir¬ 
mation of certain ideas concerning the pos¬ 
sibilities of long-distance short-wave direction 
finding and in particular the idea that there 
were times when measurements made on the 
first pulse of a pulse group would give a more 
accurate determination of the bearing of a sta¬ 
tion than would be obtained by ordinary d-f 
means. 

Another object of the project was to obtain 
evidence as to what percentage of the time 
during which energy arrived over paths devi¬ 
ated from the great circle, energy also arrived 
over great-circle paths in sufficient amounts to 
operate a d-f system. For the period covered 
by the observations this condition existed for 
80 per cent of the time. 

32 DIRECTION FINDING 

SOURCES OF ERRORS 

When energy is received over two or more 
paths, errors can be produced in certain types 
of short-wave direction finders even if the 
paths are all confined to the plane of the great 
circle passing through the transmitter and d-f 
locations. 2 These errors result from the fact 
that interference of the different components 
with one another produces instantaneous fields 
at each element of the antenna system, the 
phases and amplitudes of which are not deter¬ 
mined solely by the wave direction and the 
geometry of the antenna system. 

a Project C-35, Contract No. OEMsr-310, Western 
Electric Company. 


Furthermore, if one or more of the paths is 
deviated from the great circle, then practically 
all direction finders will give erroneous bear¬ 
ings. The extent of the errors and the per¬ 
centage of time that they exist will depend 
upon the relative intensities of the components 
arriving over the various paths. Appreciable 
errors can be obtained even when the great- 
circle energy is greater than that arriving over 
the deviated paths. 

Studies of short-wave radio transmission 
across the North Atlantic have disclosed two 
types of transmission phenomena which would 
produce such errors. During severe magnetic 
storms large amounts of energy have been ob¬ 
served arriving from the transmitter over 
paths which were widely deviated from the 
great-circle plane between the transmitter and 
receiver. At such times it has occasionally been 
observed that small amounts of energy arrive 
over a great circle path. At other times, during 
more or less normal transmission periods and 
on relatively low frequencies when energy ar¬ 
rives over several different paths, it has been 
observed that energy which has suffered sev¬ 
eral reflections at the ionosphere may be devi¬ 
ated appreciably from the great circle, whereas 
that which has suffered only a very few reflec¬ 
tions will be deviated only very slightly if at 
all. Where d-f methods provide no opportunity 
of separately identifying the great circle and 
deviated path components, errors might there¬ 
fore be anticipated during undisturbed as well 
as disturbed transmission periods. 

By the use of short-pulse transmissions it is 
generally possible to separate the components 
transmitted over different paths on a time 
basis and accordingly to measure the direction 
of each path. When the different paths are all 
confined to the great-circle plane, direction 
finding on any of the pulses should therefore 
result in an improvement since those errors are 
eliminated which are caused by the interfer¬ 
ence of the various components with one an¬ 
other. However, if all of the paths are not con- 


55 



56 


STUDY OF RADIO PULSE PROPAGATION 


fined to the great circle, and if the pulse upon 
which the measurements are made is chosen 
at random or because it is the strongest, errors 
in bearing would still be obtained. On the other 
hand, if the first pulse of a group is chosen it 
should generally give the most accurate bear¬ 
ings since it will have traveled over the most 
direct path. The work covered by Project C-35 
was undertaken to verify this conclusion. 

33 EXPERIMENTAL PROCEDURE 

Pulses were transmitted from the University 
of Puerto Rico with equipment made available 
and maintained in operation through the efforts 
of G. W. Kenrick. A small rhombic antenna 
directed towards New York City was used. The 
bearing of the transmitter from Holmdel is 
160° measured clockwise from true north and 
1,581 miles distant. The transmitted pulses 
were 100 microseconds long and had a peak 
power of about 1 kw. The recurrence rate was 
60 per second except for some of the prelimi¬ 
nary experiments when rates of 20 and 30 per 
second were used. 

Measurements on the direction of arrival of 
the individual pulses were made with the 
Holmdel Musa receiving equipment in accord¬ 
ance with a procedure described in a previously 
published paper. 3 As pointed out in that paper, 
two sets of antennas with different axes of 
orientation can be used in connection with the 
Musa equipment to determine the actual direc¬ 
tion of arrival of the waves in space. For these 
experiments only two antennas of each set 
were used instead of the usual six and the 
phase shifters were adjusted for cancellation 
instead of addition. This permitted the use of 
two widely spaced antennas of each set thereby 
giving greater accuracy in the bearings. Check 
measurements were made occasionally with 
closer antenna spacings (adjacent antennas) 
in order to avoid ambiguous results. The band 
width of the receiving equipment was sufficient 
to pass the pulses as transmitted without ap¬ 
preciable alteration in their shape. 

Pulses were transmitted on three different 
frequencies; 17,310, 7,175, and 6,425 kc. Dur¬ 
ing the first month pulses were transmitted on 
17,310 kc during the daytime and on 6,425 kc 
during the evening and nighttime hours. Ob¬ 
servations were made during two hours in the 


morning and two hours in the evening for three 
days a week. During the last two months the 
schedule was changed. Pulses were transmitted 
continuously on 17,310 kc for the first half of 
each week and on the lower frequency during 
the second half of each week, thus permitting 
observations to be made on either frequency 
during any desired hour of the day or night. 
During these last two months special attention 
was paid to the transmission conditions exist¬ 
ing during the sunset period since it had been 
observed that rather wide deviations in the 
direction of arrival occurred at that time. 

The 7,175-kc frequency was substituted for 
6,425 kc during the last few weeks of the tests 
because the interference on the latter fre¬ 
quency became so severe that reliable measure¬ 
ments could not be obtained. 

3 4 RESULTS 

Measurements were made between January 
12, and March 23, 1942, inclusive, on a total 
of 185 pulse groups on 17,310 kc and on 87 
pulse groups on 6,425 and 7,175 kc, the data 
taken on these last two frequencies being 
grouped together. Some of the pulse groups 
consisted of only one or two distinct pulses 
while others consisted of five or six or more 
pulses, some of which overlapped to such an 
extent that the individual pulses were indis¬ 
tinguishable. Of the 185 groups measured on 
17,310 kc only 5, or 2.7 per cent, contained 
pulses which arrived over paths deviated by 
more than 2° from the great-circle path to the 
transmitter and the maximum deviation was 
only 3°. Of the 87 groups measured on 6,425 
and 7,175 kc, 35 or 40 per cent contained pulses 
which arrived by paths deviated by more than 
2° from the great-circle path. The greatest 
deviation measured was 12.5°. These results 
are shown graphically in Figures 1A and IB 
where the deviations from the true bearing are 
plotted as abscissas and the number of pulse 
groups containing pulses with a given devia¬ 
tion are plotted as ordinates. Since the experi¬ 
mental error varied from 1.5° to 2°, depending 
upon the frequency used, observed deviations 
of 2° or less are not considered as significant 
and accordingly are not shown on these graphs. 

Four of the five pulse groups on 17,310 kc 
and twenty-nine of the thirty-five on 6,425 and 



DISCUSSION 


57 


7,175 kc which were observed to contain pulses 
which arrived over deviated paths also con¬ 
tained pulses which arrived earlier and over 
paths deviated by not more than 2°. In some 
of these cases the energy arriving over the 
deviated paths was appreciable so that bearing 


1 

1 

1 

1 

1 

1 

1 

1 

1 

n 


17310 KC 

185 PULSE GROUPS 

8 4 0 


4 8 


EAST WEST 


<r 

o 

UJ 

CO 

_l 

CL 



DEPARTURE FROM TRUE BEARING 
IN DEGREES 
B 

Figure 1. Percentage of pulse groups containing 
pulses deviated from great circle. 

measurements on the first pulse of the group 
would have given significantly more accurate 
bearings than ordinary d-f measurements. In 
the remaining one of the five 17,310-kc groups 
and in the remaining six of the thirty-five 
lower frequency groups, no earlier true bear¬ 
ing pulses were observed within the time limit 
of thirty minutes allowed for the measure¬ 
ments to be considered as including only a 
single pulse group. It is entirely possible that 
even in these few cases there were less deviated 
paths that would have become evident had 
higher-powered pulses been available. 

as DISCUSSION 

Aside from the improvement to be gained 
by direction finding on pulses in general, it was 
found that, under similar conditions as to path 
length and location, direction finding on the 


initial pulse on frequencies around 17 me would 
result in only a very slight improvement in 
accuracy, but on the lower frequencies the im¬ 
provement would at times be appreciable. 

This lack of expected improvement on the 
high frequency results from the fact that the 
transmission on these frequencies is generally 
confined very closely to the great-circle plane. 
This is in accord with previous experience 
that, in general, the higher frequencies are 
better than the lower frequencies for d-f pur¬ 
poses. This seems to be true, not only during 
normal undisturbed days, but also during mag¬ 
netic storms, the reason probably being that 
h-f transmission takes place by lower angle 
paths with fewer reflections at the ionosphere 
so that it is less adversely affected by varia¬ 
tions in that medium. 

During the period over which these experi¬ 
ments were conducted there was only one short 
severe magnetic storm and no measurements 
were taken during the height of that storm. 
Observations made during the following days 
and during other slightly disturbed periods in¬ 
dicated that for this particular path the only 
effects were a decrease in field strengths and a 
very slight increase in the number and extent 
of the deviations observed on the lower fre¬ 
quencies. This lack of a pronounced magnetic 
storm effect is not inconsistent with previous 
observations, for it has been observed that 
radio paths which pass near the magnetic poles 
are in general much more severely affected by 
magnetic disturbances than those paths which 
are distant from the poles. If more conclusive 
evidence of the improvement to be expected 
during disturbed periods by initial pulse meas¬ 
urements is desired, it is believed that pulse 
transmissions over a path much nearer the 
magnetic pole than the one used for these ex¬ 
periments will have to be studied. 

In the light of past experience with continu¬ 
ous-wave transmission over the North Atlantic 
path and present experience with the pulse 
transmissions from Puerto Rico, it is felt that 
it can safely be predicted that direction finding 
on the first pulse will give a significant im¬ 
provement in accuracy for a large percentage 
of the time during magnetic storms for trans¬ 
mission paths near the magnetic poles. 

Those engaged in short-wave d-f research 


















58 


STUDY OF RADIO PULSE PROPAGATION 


recognize that one of the most severe conditions 
under which direction finders must operate oc¬ 
curs when the d-f location is just outside the 
ground-wave range of the transmitter and still 
so close to it that the ionospheric waves arrive 
at very high angles of incidence. Under such 
conditions the sensitivity of most direction 
finders to the desired polarization is very low 
so that any errors caused by spurious pickup 
are greatly accentuated. Furthermore any 
slight irregularities in the ionosphere can cause 
the path of the waves to be deviated consid¬ 
erably from the great-circle plane. If the 


ground-wave range were extended considerably 
for such cases by increasing the peak power 
of the transmitted signal, as can be done using 
the modern pulse technique, and if bearing 
measurements are taken on the first or ground- 
wave pulse, considerable improvement in ac¬ 
curacy would be expected. To test these conclu¬ 
sions would require a high-powered pulse 
transmitter located relatively close to the re¬ 
ceiver location. The experiments discussed 
above do not apply to this case at all since the 
distance was entirely too great for the ground 
wave to be effective with the power used. 



Chapter 4 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


Study of theoretical and practical aspects of wide¬ 
band directive antennas for direction-finding [d-f] use 
in 150- to 300-mc region led to development of two an¬ 
tennas—a corner reflector arrangement and a flat 
reflector, in each case with the array before the re¬ 
flector. Proper phasing of array elements before the 
flat reflector rotated the directivity in azimuth. De¬ 
sign of a transformer for converting a balanced to an 
unbalanced system, use of new methods for evaluating 
polarization errors and for measuring electrical char¬ 
acteristics of the ground, and studies of the impedance 
characteristics of cylindrical dipoles of large trans¬ 
verse dimensions, formed a part of this study. The 
project a is reported rather fully here, the chief abridge¬ 
ment of the contractor’s final report 1 lying in the 
omission of certain photographs of the equipment and 
certain charts that resembled closely those reproduced 
herein. 

41 INTRODUCTION 

P rior to World War II, d-f systems operating 
in the u-h-f region were generally of the 
elevated H, fixed or rotatable, Adcock type. 
Their properties had been extensively studied 
and were well known. On the other hand, later 
work with certain types of arrays and their 
application to radar and closely related fields 
indicated that improved systems having con¬ 
siderably higher gain and broad-band response, 
particularly where portability was an impor¬ 
tant factor, could be devised for d-f use. 

The studies in this project, therefore, con¬ 
sisted of the design of reflectors and arrays of 
the corner- and flat-reflector types; of means 
for rotating the directivity of the flat reflector 
by phase adjustment of the array elements; of 
the use of cathode-ray oscilloscopes for visual 
indication of bearing including electrical cir¬ 
cuits for obtaining CRO patterns easily inter¬ 
preted; and, finally, some comparative studies 
of a differentially connected V array and the 
conventional elevated H Adcock. 


a Project No. 13.1-82, Contract OEMsr-1009, Radio 
Corporation of America. 


42 RESEARCH FACILITIES 

The site selected for this study is located on 
flat farm land near Medford, New Jersey, in 
an area known geologically as the Middle Marie 
Beds. The land is chiefly soil with small pro¬ 
portions of sand and clay and is known to be 
homogeneous to a considerable depth. A 10x12- 
foot building was erected to house the equip¬ 
ment and a 90-foot pole and rigging was in¬ 
stalled for making the polarization error meas¬ 
urements. So far as possible the building was 
constructed of nonconducting materials. Wood 
and masonite were used as the basic materials 
and, with the exception of removable metallic 
window screens, metallic reflecting surfaces 
were kept to a minimum. The pole for support¬ 
ing the polarization test transmitters was 
equipped with a removable carriage raised or 
lowered by means of a windlass. Wooden 
dowels were used in place of nails or bolts in 
all the structure above a fixed platform sur¬ 
rounding the pole and located at the same eleva¬ 
tion as the roof of the test house to permit mea¬ 
surements at horizontal incidence of the array 
located on the roof. 

It was found later that complete elimination 
of metallic objects in the construction of the 
equipment on the pole was not necessary at the 
frequencies used, and that metal could have 
been employed in limited amounts in the wind¬ 
lass and possibly in the pulleys. The effects of 
the metallic window screens were negligible 
since the screens were not in the line of the 
direct or ground-reflected waves at the trans¬ 
mitter. Presence or positions of persons or ob¬ 
jects in the test house had almost no influence 
on bearings from the two types of arrays 
tested. 

Power obtained from lines 600 feet away 
came to the pole in a shielded conduit buried to 
a depth of 18 inches and to a depth of two feet 
between pole and house. A six-conductor line 


59 



60 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


between house and pole (to provide meter out¬ 
lets at the house for circuits located at the 
pole) was buried to a depth of two feet in a 
trench containing the telephone circuit. The 



Figure 1. Elevation view of test site. Pole and 
measurement house were 100.33 ft apart. 



Figure 2. Carriage employed for hoisting trans¬ 
mitter for making polarization measurements. 


discontinuity in the ground characteristics 
caused by the buried cables was not serious as 
subsequent site error measurements proved. 


421 Receiving Equipment 

The receiver was an experimental model 
SCR-616 supplied by the Eatontown Signal 
Corps Laboratory. It covered the bands 150-300 
and 300-600 me. It consisted of a superhetero¬ 
dyne receiver with one r-f stage on the lower 
frequency band and no preselector on the 
higher band. The input was designed to operate 
from a 95-ohm balanced line. Satisfactory re¬ 
sults were obtained in most cases by operating 
the receiver from an unbalanced line. 

The receiver had good performance charac¬ 
teristics for the service required. It was fitted 
with an a-f injection oscillator to modulate the 
intermediate frequency to produce an audio 
component when receiving pure c-w. The re¬ 
ceiver sensitivity, expressed in microvolts to a 
95-ohm dummy antenna, modulated 30 per cent 
at 400 cycles, required to produce a change in 
output voltage of two to one with carrier on 
and with modulation on and off respectively, 
was approximately 5 microvolts on the lower 
frequency band and 15 microvolts on the higher 
band. 

422 Measuring Equipment 

A slotted transmission line was employed for 
impedance measurements and for a wide range 
of other necessary measurements. A microam¬ 
meter equipped with a tilted mirror was found 
convenient for making observations when it 
was mounted on the elevated carriage and read 
with the aid of high-powered prism binoculars 
from the ground. A General Radio power os¬ 
cillator covering the range of 150-600 me fur¬ 
nished signals. 

4.3 v-l ARRAY (1 DIPOLE PER SCREEN) 

The V type of antenna system consists of 
two similar linear cophasal broadside arrays, 
each placed in front of one side of an angled 
reflector. The two arrays have mirror-image 
response patterns, each nearly symmetrical, ex¬ 
cepting that one is rotated in azimuth with ref¬ 
erence to the other by an angle equal to the 
angular displacement of its reflector from the 
plane of the other. The direction of maximum 
response of each array is normal to the corre¬ 
sponding reflector. See Figures 3A and B. 




















V-l ARRAY (l DIPOLE PER SCREEN) 


61 


It may be observed that the patterns of the 
two arrays can be made to intersect at some 
desired point, depending on the angle of the 
reflector. This intersection represents equal re¬ 
sponse of the two arrays, and may be used as 
a bearing indication if the amplitudes are suit- 


BEARIN6 LINE 

* 



Figure 3. A shows typical V array. B shows 
typical response patterns of individual antennas. 


ably compared in associated indicating equip¬ 
ment. A number of indicating methods are 
available and are discussed later. 

Two collector systems of this general type 
were studied, the first consisting of one dipole 
before each reflector (V-l), and the second 
having two dipoles in front of each reflector 
(V-2). Larger numbers of cophasal antennas 
per screen are possible, but were not studied 
because the resulting size in the low-frequency 
band was considered excessive in view of porta¬ 
bility requirements. On the higher-frequency 
band, twice the number of elements may be 
used without exceeding the size of the low- 
frequency array. 

A large portion of the experimental work in 
this project was done on the V array having 
one dipole per reflector, and while this array 
has the least favorable performance of all con¬ 
sidered, most of the information obtained was 
useful in carrying out the examination of the 
other arrays. 


4,31 Reflectors 

The first problem presented was the deter¬ 
mination of reflector size and mesh. The experi¬ 
ence of other groups engaged in antenna re¬ 
search indicated that a reflector approximates 
a perfect plane conductor of infinite extent if 
the dimensions are such as to exceed by one- 
eighth wavelength in all directions the maxi¬ 


mum dimensions of the array with which it is 
used at the lowest frequency of operation, pro¬ 
viding that the smallest dimension of the array 
is at least one-half wavelength (a/ 2) at this 
frequency. A maximum spacing of A/20 be¬ 
tween members making up the screen at the 
highest frequency to be used was indicated, 
with the length of the elements oriented along 
the direction of desired polarization. The pres¬ 
ent study verified the adequacy of these limits. 
An increase in the screen size above this figure 
resulted in small performance change. Substi¬ 
tution of high-conductivity fine mesh screen 
also resulted in no material improvement in 
performance. 



Figure 4. V-l array in front of screen. Coupling 
transformers are behind screen at angle to an¬ 
tennas. 


The screens used were fabricated of 3/16- 
inch stainless steel tubing, spaced two inches 
apart, and made in sections so that the overall 
size was easily adjustable. The members sup¬ 
porting the dipole assemblies were designed to 
permit adjustment of the spacing between 
dipoles, and the spacing to the screen. The 
assembly was copper plated and protected by a 
coating of enamel. See Figure 5 for dimensional 
drawing. 





















62 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


i. 3.2 Dipole Dimensions and Impedance 
Characteristics 

The determination of proper dipole dimen¬ 
sions required considerable attention. The 
parameters chiefly affected by the dipole dimen¬ 
sions are the radiation impedance and its re¬ 


sistive and reactive components. The efficiency 
of energy transfer from the dipoles to the 
utilization circuits depends on the impedance 
match through the system (aside from line 
losses) ; it was therefore necessary to establish 
some criteria to guide the work toward obtain¬ 
ing a desirable characteristic. 



DIMENSIONS IN INCHES 


Figure 5. Essential physical dimensions of dipole and its reflector. 







































































































































































































































































V-l ARRAY (l DIPOLE PER SCREEN) 


63 


The input circuits of receiving equipments 
in the range of frequencies covered, 150-600 
me, differ markedly from lower frequency 
equipment in that power is generally consumed 
in the former due to an input impedance char¬ 
acteristic which may reach low values. This is 
due primarily to finite electron transit time 
effects which depend on the size of the first 
amplifier tube, its geometry, and electrode volt¬ 
ages. The input conductance varies directly 
with the square of the frequency. As a con¬ 
sequence receiving equipments may have widely 
different input impedance characteristics, and 
the input impedance of a receiver will generally 
show a large variation over a two-to-one fre¬ 
quency range. This impedance is primarily re¬ 
sistive since the circuit is usually tuned. With 
an antenna and receiver whose impedances are 
different functions of frequency, a matched con¬ 
dition can not be realized with a fixed trans¬ 
mission line over a wide band of frequencies. 
In most cases the condition for best signal-to- 
noise ratio corresponds to the condition of 
maximum power transfer. This latter condition 
requires that at any point in the system the 
impedance in one direction must be the complex 
conjugate of the impedance in the other direc¬ 
tion. 

In view of these facts it is considered im¬ 
possible to set up absolute criteria for the im¬ 
pedance characteristics of an antenna array 
without a complete knowledge of the equipment 
with which it is to be used. An alternative 
which is thought to be satisfactory is to ap¬ 
proximate a uniform resistive impedance 
through the range. The mismatch between this 
and a suitable transmission line should not be 
too severe. The input circuits of receivers ap¬ 
pear to offer a greater degree of flexibility for 
applying means to manipulate impedance char¬ 
acteristics,* and an attack of the problem in this 
direction should more readily yield the desired 
results. It is not unlikely that incomplete in¬ 
formation in the hands of receiver designers on 
wide-band antenna impedance characteristics 
is one of the chief reasons why such large 
variation is encountered in the input circuits 
of receivers. In comparison two of the systems 
developed in this project, the V-2 and flat 
arrays, show much smaller variations, while 
the variation of the V-l is of the same order as 


a representative receiver covering a similar 
frequency range. 

The preliminary design work on the corner 
array envisioned the use of dipoles mounted by 
metallic tubes supporting each half, the two 
tubes forming in effect a parallel wire trans¬ 
mission line shorted at the reflector surface. 
Electrically this represents an almost pure re¬ 
actance in shunt with the radiation impedance 
of the dipole. This shunt reactance was ex¬ 
pected to cancel partially the radiation reac¬ 
tance of the dipole; the length and charac¬ 
teristic impedance were chosen so as to accom¬ 
plish this. 

Preliminary measurements showed large dis¬ 
crepancies between actual and expected results, 
due largely to insufficient information on the 
characteristics of cylindrical dipoles of large 
transverse dimensions, the design work having 
been based on prolate spheroidal dipoles. Also, 
the effect of the reflector was not fully ac¬ 
counted for. The results of subsequent theo¬ 
retical investigations on these points are given 
below. 

As a result of information obtained in these 
measurements, it was decided to change the 
design to a single-support, insulated dipole, 
using twin coaxial cables for interconnection. 
This eliminates the shunt reactance of the 
double support, and replaces it with the much 
higher reactance of a single insulated support. 
This balanced configuration was expected to be 
Jess susceptible to response from fields of un¬ 
desired polarization. 

Dipole Considerations 

The frequency range which a dipole must 
cover restricts the choice of its length; this 
should be a half wavelength near the center of 
the band. The other constants which may be 
adjusted are the ratio of diameter to length, 
and the spacing before the reflector. A large 
ratio of diameter to length gives a low antenna 
characteristic impedance and resulting low Q. 
The limitations in this respect are chiefly me¬ 
chanical, and depend on the portability re¬ 
quired in the equipment. Weight and mechani¬ 
cal strength and rigidity are more favorable in 
the smaller diameter dipoles, and they are more 
easily supported, particularly in the balanced 
type of structure requiring members insulated 



64 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


from the support. The only practical limita¬ 
tions on the spacing before the screen are the 
effect on the impedance variation and the 
change in the response patterns. With a single 
dipole before a screen, the response pattern 
begins to break up into two lobes as a quarter- 
wavelength spacing is exceeded. This is most 
pronounced at the high-frequency end of the 
range, where the fractional wavelength spacing 
is the greatest. When this condition is encoun¬ 
tered, the antenna gain drops, and internal 
screen angles much less than 90° are required 
to obtain satisfactory overlapping of lobes. 
With arrays of more than one element per 
screen, this limitation is less serious, since the 
gain is higher and the response patterns are 
sharper. 

Impedance Considerations 

The effect on the impedance due to the spac¬ 
ing from a reflector may be examined most 
readily by replacing the screen by the negative 
image of the dipole. The mutual radiation im¬ 
pedance between the two dipoles modifies the 
impedance of the original dipole. The mutual 
reactance may be either positive or negative, 
and the mutual resistance may also be of either 



FREQUENCY-MEGACYCLES / SEC 

Figure 6. Impedance characteristics of V-l array. 
Antenna-to-reflector space is 28.5 cm (X/4 at 263 
me). 

sign, depending on the spacing in terms of 
wavelength. The resistance decreases when the 
mutual resistance is positive, and increases 
when it is negative. When the self and mutual 
reactances are of the same sign, the reactance 
decreases, while if of opposite sign, an increase 
takes place. Since these mutual effects are 


dependent on the proximity of the dipole and 
its image, the impedance variations through 
a wide frequency range are generally greater 
when the spacing is small. The impedance char¬ 
acteristics, as measured at the dipole termi¬ 
nals, of the final V-l array are shown in Figure 6. 


30 ° 20 ° 10 ° 0 ° 350 ° 340 ° 330 ° 



Figure 7. V-l array, relative response in azimuth, 
0° elevation, 150 me. 


43 3 Directivity in Azimuth 

The directivity of this array in azimuth re¬ 
sults from the use of the reflector, since the 
pattern in the equatorial plane of a dipole in 
free space is circular. As the frequency is in¬ 
creased so that the spacing from the reflector 
exceeds A/4, the pattern begins to break up 


30 ® 20 ° 10 ® 0 ® 350 ® 340 ® 330 ® 



Figure 8. V-l array, relative response in azimuth, 
0° elevation, 300 me. 


into two lobes, with a minimum normal to the 
reflector. This minimum reaches zero at a spac¬ 
ing of a/2. As a result, the spacing may not 















V-l ARRAY (l DIPOLE PER SCREEN) 


65 


appreciably exceed A/4 at the high-frequency 
limit. The patterns exhibit a broadening with 
increasing frequency because of this phenome¬ 
non, but are seen to be usable through the two- 
to-one frequency range. 

The general shape of azimuthal curves at 150 
and 300 me for elevations from 0° to 30° at 10° 
intervals changes somewhat through this range, 
but not sufficiently to affect the performance 
adversely (see Figures 7 and 8). The change 
in relative size of the patterns of the two an¬ 
tennas is due to the unsymmetrical effect of the 
horizontal electric field component lying in the 
plane of propagation; the shift of the lobe in¬ 
tersection from 0° azimuth is, therefore, a 
polarization effect. 


able. In an actual receiving system the inher¬ 
ent noise limits the useful amplification. If an 
actual receiver is considered operating with the 
antenna, a certain amount of noise will be pres¬ 
ent in the output. If a signal now arrives at 
the antenna, the output of the receiver will be 
proportional to the field intensity, and to the 
relative response r of the antenna at the azi¬ 
muth of wave arrival. (The gain of the antenna 
need not be considered at this point since its 
effect is merely to change the factor of propor¬ 
tionality.) For a fixed field intensity of the 
signal, therefore, the signal-to-noise ratio of 
the output is proportional to r, and from this 
standpoint best operation is had when r is a 
maximum. 


434 Determination of Lobe Intersection 

In d-f systems using amplitude comparison, 
the question arises as to the best point to use 
in intersecting the lobes, where such choice 
exists. The lobe intersection in the V array is 
determined by the angular position of the two 
reflectors, and can be adjusted to any desired 
point. It is, therefore, useful for this as well 
as other similar systems to consider a number 
of factors which enter into the selection of the 
intersection point, and if possible, to define an 
optimum point. 

Consider a simple array such as the corner 
type with one element per reflector. If an 
idealized point antenna is placed before a 
screen, the relative response as a function of 
the azimuth angle measured from the normal 
to the reflector is given by 



Here r is the relative response, <£ the azimuth 
angle, s the spacing from the reflector to the 
point antenna, and A the wavelength. This 
function is plotted in Figure 9 with s taken 
equal to A/4. If a receiving equipment having 
an ideal noise characteristic, that is, one pro¬ 
ducing no internally generated noise, is thought 
of as being used with this antenna, then an 
output of any desired magnitude may be ob¬ 
tained from a signal arriving from any direc¬ 
tion where the antenna response is not zero, 
assuming unlimited amplification to be avail¬ 



Figure 9. Plot of functions r and rdr/d<f> for deter¬ 
mining optimum lobe intersection, V-l array, 
single-point antenna A/4 before screen. 

The operation of indicating systems used in 
conjunction with switched lobe antennas usual¬ 
ly depends on the difference in output when the 
signal is sampled successively on the two an¬ 
tennas. For example, a differential rectifier 
might be used to actuate a zero-center micro¬ 
ammeter, or to control the input circuits of a 
servoamplifier for automatic tracking. 

Sensitivity Factors 

The sensitivity of such a system depends on 
the magnitude of the difference response for an 
increment of azimuth angle in the vicinity of 
the equisignal point, that is, the intersection 
point of the two lobes. The magnitude of the 
difference in turn depends upon two factors— 
the first of which is the slope of the antenna 









66 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


response curve at the operating point; this 
quantity might be appropriately termed “dif¬ 
ferential sensitivity.” The second is the scale 
factor, which depends on the amplification and 
signal intensity. If we assume that the maxi¬ 
mum available linear amplification is used, then 
for a fixed signal intensity the differential out¬ 
put is proportional to dr/dcf>. As previously 
indicated, the signal-to-noise ratio is propor¬ 
tional to r. The conditions of maximum signal- 
to-noise ratio and differential sensitivity can¬ 
not be satisfied simultaneously, since drjd<\> is 
zero when r is a maximum. If equal importance 
is assigned to r and dr/d<f> the maximum value 
of the product may be defined as the optimum 
intersection point. That is, 



The product rdr/d<f> may be most conveniently 
maximized graphically, especially when experi¬ 
mental response curves are used. 

For the case of a single-point antenna placed 
a distance A/4 before a screen the functions 



are plotted in Figure 9. The maximum value of 
the latter is seen to occur at 0 of approxi¬ 
mately 60°. 

For the case of two-point antennas placed 
before a screen, the response pattern is given 
by 


/ 2tt d . 

I - SI 


V X 


sin 0 1 sin I cos 0 I 


and 

dr 


its . „ (2tt d . \ . 0 /2irs \ 

d<t>=-\ sm * cos Vx~ sin V sin2 Vx C0S V 

— ~ cos <t> sin 2 ^cos sin 2 s * n ^ ■ 


Here the quantity d represents half the dis¬ 
tance between the two dipoles. These two func¬ 
tions are plotted for d — s = A/4 in Figure 
10. The maximum of the second occurs at 0 of 
approximately 28°. 


This figure of merit fails under extreme con¬ 
ditions of signal intensity. If the signal is on 
the threshold of noise, r becomes more impor¬ 
tant than its derivative, while at the other ex¬ 
treme, where the receiving system is overload¬ 
ed, operation at lower values of r is indicated. 
The first of these extremes is more likely to 
occur in practice; however, deviation from the 
above criteria should be based on statistical 
data obtained in actual use in the field. The 
data should include the noise characteristics of 
the receiving equipment and the field intensi¬ 
ties encountered. The speed of response of the 
indicating circuits, or the minimum time in 


30 ° 20 ° 10 ° 0 ° 350 ° 340 ° 330 ° 



Figure 10. Data similar to that given in Figure 9 
except for two-point antenna. In both illustrations, 
rdr/d<f> is product of azimuthal response and rate 
of change of this response with respect to azi¬ 
muthal angle. 


which integration is substantially complete, 
coupled with the signal-to-noise ratio data 
should indicate the direction and extent of the 
departure required from the above criteria. 

In the two cases considered, the rdr/d<f> 
curves are fairly broad near the maxima; for 
one doublet, the width of the curve is 20° at 
90 per cent of maximum, while for two doublets 
the width is 16°. This width affords some 
latitude of choice without departing far from 
the optimum. When an array is used over a 
two-to-one frequency range, the shape of the 
response pattern changes with frequency. The 
intersection point can be selected at the mean 
frequency, and performance will generally be 
satisfactory throughout the range. 









V-l ARRAY (l DIPOLE PER SCREEN) 


67 


4 ' 3 ' 5 Relative Response in Elevation 

Because of interference effects between the 
direct and reflected waves at the receiving an¬ 
tenna, the direct measurement of the directive 
pattern in elevation holds only for a specified 
antenna height above ground and fixed ground 
constants. For this reason an indirect method 
was adopted for the measurement, and the pat¬ 
terns so obtained are assumed to hold for all 
the arrays studied in this project. 

The method of measurement consists of de¬ 
termining the response in azimuth of one re¬ 
flector to horizontally polarized waves, with the 
receiving dipole oriented horizontally. Since 
the reflector used is very nearly square, this 
procedure, in effect, is equivalent to turning 
the entire receiving and transmitting system 
90° about a horizontal line connecting the 
transmitting and receiving points. The original 
vertical polarization now corresponds to hori¬ 
zontal, and the angle of elevation corresponds 



VERTICAL POLARIZATION 
0U= FREQUENCY 150 MC 
b = FREQUENCY 225 MC 
C =FREQUENCY 300 MC 

Figure 11. V-l and V-2 relative response in ele¬ 
vation, i.e., in half-space above the earth in absence 
of ground reflections. 

to azimuth. Ground reflections are thus con¬ 
stant, since all measurements are made at 0° 
elevation, and may be neglected. The measure¬ 
ment was made out to ±90° from the normal 
to the screen; the two halves showed slight 
dissymmetry, and the average was taken. The 
response diagrams for three frequencies, 150, 
225, and 300 me are given in Figure 11, and 
may be taken to represent the relative response 
in the half space above the earth, in the ab¬ 


sence of ground reflections. It appears to serve 
no purpose to give specific diagrams including 
the effect of ground reflections, since these 
would vary widely depending on the height, fre¬ 
quency, and ground constants. 

4 ‘ 3 ' 6 Polarization Errors 

One of the major problems encountered in 
the course of this project, and one which arises 
in most research connected with the study of 
collector systems for direction finders, is the 
investigation of polarization errors. Although 
most of the important characteristics of such 
systems can be readily determined theoretically 
or experimentally, this is not true in the case 
of polarization errors. The theoretical predic¬ 
tion is generally not possible, except in the 
case of certain elementary collectors such as 
balanced shielded loops, where the response to 
fields of any polarization is known. In the case 
of some other antennas of simple geometrical 
configuration, the shape of the response pattern 
to fields of various polarizations may be as¬ 
sumed to a good degree of accuracy; the errors 
may be predicted if the scale factors between 
them are known. The latter cannot be pre¬ 
dicted theoretically, and therefore are mea¬ 
sured; the complete performance can then be 
stated in terms of the theoretical assumptions 
and the measured values of these parameters. 

The difficulties underlying the evaluation of 
polarization errors are due basically to the lack 
of an adequate and readily measured standard 
of performance. Until recently, the standard 
wave error of Barfield 2 was widely used. This 
is defined as the error of a system when obtain¬ 
ing a bearing on a wave arriving at an angle 
of elevation of 45°, and having equal compo¬ 
nents polarized in, and perpendicular to, the 
plane of incidence, with the two components 
so phased as to produce the maximum error. 
This standard of performance is open to two 
objections. The first is that it is defined with¬ 
out consideration of the effect of the ground in 
modifying the wave arriving at the collector. 
Although the omission is justifiable in the case 
of collectors located near the ground in terms 
of wavelength, in the case of elevated systems 
the difference in reinforcement or cancellation 
of the parallel and perpendicular field compo- 







68 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


nents caused by the ground-reflected wave may 
give rise to a resultant that differs widely from 
the condition of the downcoming wave. Hence, 
large variations of the standard wave error of a 
system may be observed, depending on the ele¬ 
vation of the receiving antenna above ground 
and the electrical characteristics of the ground. 
The second objection is that a knowledge of the 
standard wave error of a system in itself is 
generally not sufficient to determine its per¬ 
formance under other conditions of wave ar¬ 
rival. The additional information necessary for 
this determination is the law of error which 
the system follows. The general law which a 
system obeys is, of course, known from the 
theory of its operation; to reduce it to a quan¬ 
titative form usable in extrapolating errors, 
other data would appear to be necessary. This 
same objection put in another form applies 
when systems following different laws are in- 
tercompared. Evidently one system may have a 
very large error at an angle of elevation of 45° 
(for example, if the vertical response is a mini¬ 
mum at this angle and the horizontal response 
is large), and have low errors at other ele¬ 
vations. A second system may respond in an 
opposite manner, and have low errors at 45° 
and high elsewhere. Comparison of these two 
on the basis of the standard wave error would 
appear favorable to the second, while the first 
may actually be superior at most other ele¬ 
vations. Therefore it is seen that two arrays 
having the same standard wave error may have 
considerably different performance under other 
conditions. 

To some extent the situation has been clari¬ 
fied by the work of the National Bureau of 
Standards [NBS] as summarized in the final 
report on Project C-18. 8 The report is con¬ 
densed in Chapter 1 of this volume. The meth¬ 
ods and criteria developed by NBS for the eval¬ 
uation of collector performance with regard to 
polarization errors overcome these objections 
to a certain extent and should be applicable, in 
theory at least, to most collector systems. These 
methods specify performance in terms of cer¬ 
tain parameters of the system analogous to 
effective heights, measured at ground level, and 
independent of the ground constants. A knowl¬ 
edge of the law of error of the system enables 
the complete performance to be stated. It was 


thought desirable to apply these methods in the 
present project, subject to verification of the 
results by other methods, principally, the direct 
measurement of the errors. Unfortunately, the 
attempt to adapt these methods in the present 
case did not result in any marked degree of 
success. 

The NBS Method 

Essentially, the NBS method is based on the 
statement of the response of the antenna sys¬ 
tem to an arriving field in terms of the desired 
response of the true antenna elements and the 
undesired response due to extraneous elements 
such as feeders, etc. The response of the an¬ 
tenna is analyzed on the basis of three resolved 
field components, with a directivity function 
associated with each, and a parameter analo¬ 
gous to effective height, called the pickup factor, 
also associated with each. These latter relate 
the voltages induced by a component to the 
intensity of the component producing it, and 
therefore have the dimensions of effective 
height. The response of the feeders is similarly 
stated for the three field components. The equa¬ 
tions may be written as follows: 

U an t = KEF(M) (1) 

feeders = W(<W) (2) 

Here V is the voltage induced in the element 
indicated in the subscript, h and k the pickup 
factors, E the electric field intensity (for sim¬ 
plicity the magnetic field components will not 
be considered), and F and / the directivity 
functions, dependent on the azimuth <£, and the 
angle of elevation if/. The field terms on the 
right hand side of the two equations are re¬ 
solved into three components, and the other two 
factors are likewise resolved to correspond. 

The resolution of the field at the collector is 
indicated in Figure 12, where the two primary 
components E n and E p , shown at A, are respec¬ 
tively perpendicular to and in the plane of 
incidence. 

The parallel component is further resolved at 
B into a vertical component E v> ~ and a hori¬ 
zontal component E v>x \ the direction of propa¬ 
gation associated with each vector is indicated 



Y-l ARRAY (l DIPOLE PER SCREEN) 


69 


in the figure. Following this resolution, equa¬ 
tions (1) and (2) can be written 

^ant = + KE p ,f z (M) 

+ h y E n F y (M ) (3) 

feeders = WpM*’*) + WpJAM) 

+ k y E n f y (M) (4) 

The sum of these two voltages is the total re¬ 
sponse of the system to the existing field. 



Figure 12. Resolution of electric field at collector 
into components parallel with and perpendicular to 
plane of incidence. 


The factors h and k for the various compo¬ 
nents are to be obtained empirically, while the 
directivity functions are deduced theoretically 
from a knowledge of the configuration of the 
system. One or two of the h factors in the an¬ 
tenna response may be zero or negligible; for 
vertical dipoles, for example, h x and h y are 
zero, simplifying the situation. For the un¬ 
desired response all of the Afs may be present; 
often two are sufficient to describe conditions. 

With respect to the directivity functions, the 
NBS procedure is to determine the dependence 
on <f> by measurements at horizontal incidence, 
while the relation to if/ is determined from a 
knowledge of the configuration of the systems. 
These directivity functions are quite general 
and may be expressed in complex form to ac¬ 
count for the phase of each term. They are 
sufficiently general to permit inclusion of the 
effect of the field set up by reflection from the 
ground. The equations, therefore, when ex¬ 
panded to include these factors, describe com¬ 


pletely the response of a system under any con¬ 
dition of wave arrival, or ground conditions. 
This response will depart from the ideal de¬ 
sired response because of the undesired pickup 
present; an analytical comparison of the actual 
with the ideal response enables the determina¬ 
tion of the polarization errors of the system. 
Generally, the phase modifications undergone 
by the various induced voltages through the 
mechanisms whereby they are induced and 
transferred from the responding elements to 
the utilization circuits are not known, nor are 
they readily determinable, and as a result, the 
complete equations may not be written explic¬ 
itly to include them. Nevertheless, a knowledge 
of the law which the system follows enables the 
assignment of values to these unknown phase 
angles such as to make the polarization error a 
maximum, and thus set an upper limit to the 
polarization error possible for a particular con¬ 
dition of the downcoming wave. A plot of these 
maxima over a representative range of ele¬ 
vation angles at an appropriate ratio, say one- 
to-one, of the parallel and perpendicular field 
components, gives a complete picture of the 
performance in this range, and may be used for 
comparison with other systems of the same or 
different type. 

Application to Adcock Antenna 

The first step in applying this method, 
namely, the determination of the directive pat¬ 
terns for the three field components, must now 
be subjected to further examination. For the 
purpose at hand, this may best be accomplished 
in conjunction with an illustrative example. A 
differential Adcock pair of the elevated H type 
will be considered, since the NBS report re¬ 
ferred to treats a number of this general type. 

The H Adcock consists of two vertical 
dipoles differentially connected by horizontal 
feeders. The response of this system can be 
conveniently considered as resulting from a 
combination of the desired response of the two 
vertical dipoles and the undesired response of 
the horizontal feeders. The directivity function 
for the two dipoles is known accurately on 
theoretical grounds for fields of any polariza¬ 
tion. Obviously E p>z is the only component 
capable of inducing a voltage in either dipole, 
since E v>x and E n are always directed at right 







70 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


angles to the length of the dipoles. Therefore 
h x and h y are each zero, and terms containing 
them are eliminated from the response equa¬ 
tion. 

The horizontal feeders may be replaced, as a 
first approximation, by a short dipole along the 
line of the feeders. The directivity function 
for this dipole is known from theoretical con¬ 
siderations. E p>z is always normal to the direc¬ 
tion of this dipole, and can induce no voltage 
in it; k z is therefore zero. Further, the maxi¬ 
mum response to a unit field of E n (at <f> = 0°), 
must equal the maximum response to a unit 
field of E PjX (at cf> = 90°, ^ = 90°), since in 
each case the direction of the electric field is 
parallel to the dipole in question. Therefore the 
response coefficients k x and k y are equal, and 
the measurement of one is sufficient to establish 
the other. Evidently both h z and k y may be 
separately determined by measurements at 
horizontal incidence, and their ratio obtained. 
Since the complete directive pattern is known, 
it need not be measured; the NBS procedure is 
to measure the patterns due to E n and E p>z at 
ground level, probably as a verification of the 
assumptions. The method is thus seen to be 
substantially an indirect one, in that the polari¬ 
zation error is not measured directly, but is 
deduced from theoretical considerations and 
observed data. 

It is interesting to consider the situation re¬ 
sulting if in the preceding example it were not 
possible to assign on theoretical grounds a di¬ 
rectivity characteristic to the element respond¬ 
ing to the undesired field components E n and 
E p>xy that is, if knowledge of the behavior of 
the feeders is insufficient to permit the valid 
substitution of a simple dipole. It would then 
become necessary to establish this directivity 
by empirical methods. A series of measure¬ 
ments would be made, starting for example 
with E n fields. The response through 360° in 
azimuth could be measured for an angle of 
elevation equal to zero. In carrying these mea¬ 
surements to elevated angles, however, a fun¬ 
damental difficulty would arise due to reflec¬ 
tions from the ground. When a ground-reflec¬ 
ted wave exists (and this may even apply in 
an elevated system to the measurements at zero 
angle of elevation), there are in effect two 
waves present, differing in magnitude, phase, 


and direction of arrival. Moreover, the re¬ 
sponse of the system to waves from the two 
directions may introduce additional phase and 
magnitude changes. Obviously a single figure, 
the resultant output voltage, is not sufficient to 
determine uniquely the response in the desired 
direction. Nor would it be valid to take the 
downcoming wave, compute the magnitude and 
phase of the reflected wave, and add the two 
vectorially in time and space at a point of the 
system, unless it can be assumed that the point 
adequately represents the system for the two 
waves in question, i.e., that waves do in effect 
act on the system at the point, and nowhere 
else. For example, if, instead of a single hori¬ 
zontal dipole representing the feeders, it were 
necessary to substitute two parallel dipoles 
lying in the same horizontal plane the addition 
of the direct and reflected waves at one would, 
in general, not hold for the other, since the 
path differences in the two cases are not the 
same for a nearby signal source. This assump¬ 
tion concerning the configuration could not be 
made, as the configuration itself is to be deter¬ 
mined by the measurements. Should an attempt 
be made to carry the investigation on for the 
other two field components, difficulties of the 
same nature would exist and, in addition, other 
complications would be found. The E p>x and 
E p>z components are not separable; the plane 
of incidence and direction of propagation deter¬ 
mine uniquely the direction of the E p vector 
Its resolution is useful for analysis, but cannot 
be accomplished physically so as to eliminate 
one or the other of the components. As an 
alternative, the method might be modified to 
measure the total response to the E p field, rather 
than the response to its two components. The 
effect of the components could be deduced, if 
the phases of the resultant voltages were 
known. These, however, cannot as a rule be 
determined. Moreover, in a reasonably good 
collector, the desired response of the dipoles to 
E p>z would almost completely obscure the re¬ 
sponse to E p>x , unless a very high precision 
were attained in the techniques of the measure¬ 
ment. The desired response could not well be 
eliminated, since the dipoles, while not respond¬ 
ing to the undesired components directly, may 
be, and usually are, an element in the transfer 
system from a responding member to the util- 



V-l ARRAY (l DIPOLE PER SCREEN) 


71 


ization circuits, because of radiation or re¬ 
active coupling. For example, a member re¬ 
sponsive to E PtX may reradiate a component 
parallel to the dipoles and thus induce a volt¬ 
age; removal of the dipoles, or short circuiting 
them would remove this undesired effect; the 
total effect of E p>x could not then be deter¬ 
mined. 

Some consideration might be given to a 
method of evaluating the errors on the basis of 
the primary components alone, without at¬ 
tempting the more or less artificial resolution 
of the E p field. While this might conceivably 
be possible, the other difficulties mentioned 
would still be present; in addition to the com¬ 
plicating presence of the ground-reflected wave, 
the phase angles of the voltages induced by the 
two components of the parallel field would re¬ 
main unknown. In itself, this would be of no 
consequence, since the effect of the whole E p 
field is being investigated. However, while the 
two components are always in phase in a down¬ 
coming wave, this may not be the case when 
the combined direct and reflected waves appear 
at the collector. As a result, the analytic sepa¬ 
ration would still appear to be necessary to 
predict the behavior for any ground conditions. 

Free-Space Pattern for Systems Using 
Reflectors 

By positioning the reflector near, and paral¬ 
lel to, the ground, the latter becomes in effect 
an extension of the reflector; undesired ground 
reflections would be eliminated. The signal 
source could be placed directly above the reflec¬ 
tor at a suitable height, and either moved 
through arcs corresponding to azimuth and ele¬ 
vation, or left stationary, and the reflector ro¬ 
tated as required. The objections to this method 
are that the array would have to be dismantled 
from its normal operating position, and special 
gear constructed to obtain the required rota¬ 
tion of the screen, or motion of the source. 
Further, a legitimate doubt would always exist 
concerning the equivalence of the undesired 
response in the operating and measuring posi¬ 
tions, since the feeders could not be identically 
disposed in the two cases. The uncertainty 
caused by the change in phase of the E p>x and 
E P3 s fields in the presence of the ground would 
still remain. 


These possible procedures have been touched 
upon not so much to examine their merits, but 
rather to bring out the difficulties of the 
method when recourse must be had to an ex¬ 
perimental determination of the three dimen¬ 
sional response of an array to certain field com¬ 
ponents, when the directivities involved cannot 
be assumed a priori from the configuration of 
the array. Even in the case of so simple a col¬ 
lector as an H Adcock, it is open to some ques¬ 
tion whether a successful experimental deter¬ 
mination can be made. On arrays of the type 
studied in this project, the configuration of the 
elements which may respond to undesired field 
components is considerably more complex, and 
the difficulties increase accordingly. The de¬ 
sired response of the dipoles cannot be pre¬ 
dicted to a high degree of accuracy because of 
the presence of an imperfect reflector; to this 
time a reasonably accurate expression for the 
current distribution in a cylindrical dipole of 
large transverse dimensions has not been ob¬ 
tained. Any assumption as to the total response 
of the system was out of the question. 

If consideration be given to the NBS method, 
its elegance and utility are seen to reside in its 
ability to assess in terms of simple and readily 
determinable parameters, a phenomenon which 
is at best very complex. One of its outstanding 
advantages in practice is that the measure¬ 
ments are so made as to avoid the complicating 
influence of the ground, that is, at horizontal 
incidence. In any of the procedures mentioned 
above, this advantage would be lost; the meas¬ 
urements would be time consuming and diffi¬ 
cult, if at all possible, requiring, in the case of 
arrays studied in this project, fields of high 
purity of polarization; the final results would 
be indirect and subject to question on this 
ground. 

As mentioned previously, an attempt was 
made to apply the NBS method during the 
course of the project. Some of the earlier re¬ 
sults of measurements on the response of the 
V-l array to horizontally polarized waves in¬ 
dicated that no simple space pattern could be 
presumed. The lack of a reliable field-intensity 
meter hampered the work considerably. Resort 
had to be made to the array under test for field- 
intensity comparisons. This was done by orient¬ 
ing the dipole either vertically or horizontally 



72 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


as required, and assuming the same effective 
height for the two conditions. Rejection ratios 
were specified in the bearing direction and are 
given below. These ratios were found to be of 
considerable value as an indication of the prog¬ 
ress of the work, since any substantial im¬ 
provement was usually accompanied by smaller 
measured errors. 

Poast Method of Measuring Polarization 
Errors 

The final method adopted for the measure¬ 
ment of polarization errors was one originally 
intended to verify results of the indirect meth¬ 
od. It was originated by L. M. Poast of the 
National Bureau of Standards, and consists of 
a means of producing a field polarized so as to 
have equal components in, and perpendicular to 
the plane of incidence, with a continuously 
variable phase adjustment between these com¬ 
ponents. This is accomplished by exciting three 
mutually perpendicular electric dipoles from 
the same shielded source. One dipole, used as 
the axis of rotation, is horizontal and at right 
angles to the direction of propagation of inter¬ 
est. The plane through the remaining two di¬ 
poles is vertical, and coincides with the plane 
of incidence at the receiving antenna. The 
horizontal dipole and one of the dipoles in the 
vertical plane are fed in phase, and the remain¬ 
ing vertical dipole feed is displaced 90° in 
time phase by an artificial quarter-wave line. 
The two dipoles in the vertical plane produce 
a uniform field in that plane. The phase varies 
uniformly with angular position in that plane, 
and the magnitude of the resultant field is 
equal to the maximum field due to either dipole. 
There results, therefore, a field as specified 
above, with a parallel component E p whose 
phase may be varied uniformly by rotating the 
system about the horizontal dipole as an axis. 
Figure 13 shows the unit which was designed 
to operate over the 150- to 300-mc band. The di¬ 
pole extensions are interchangeable with units 
of other lengths and telescope into the oscilla¬ 
tor housing for accurate adjustment to fre¬ 
quency. 

When this system is used for the measure¬ 
ment of polarization errors, it is supported so 
as to permit rotation about the horizontal di¬ 
pole, then elevated to the desired height. The 


errors are observed as the assembly is rotated 
through 360° by means of control cords. The 
maximum error is noted, as well as errors at 
uniform angular intervals. The maximum error 
then is the maximum possible for a one-to-one 
downcoming field at that elevation angle, re¬ 
ceiving antenna elevation, and ground con¬ 
stants. The measurement of the polarization 
errors is direct, and is, therefore, not open to 
those objections which are based on the in¬ 
directness of a method. Three factors never¬ 
theless may be questioned. The first is the va¬ 
lidity of the results based on radiation from a 
nearby transmitting system. At the lowest fre- 



Figure 13. Variable phase polarization transmit¬ 
ter with removable dipole extensions which tele¬ 
scope into oscillator housing for accurate adjust¬ 
ment to frequency. 

quency in question, the distance between the 
receiving and transmitting points is about 15 a 
along the ground and 18 a at the maximum 
elevation of 34°. It is believed that this is ade¬ 
quate to produce a substantially plane wave 
front at the receiver. The effect of the surface 
wave is greatest at low angles of elevation; it 
may be neglected at the higher angles where 
the polarization errors reach their maximum 







73 


V-l ARRAY (l DIPOLE PER SCREEN) 


values. The second objection is that measure¬ 
ments are made with the receiving antenna 
array at a fixed height above ground, and may 
not represent the worst point of operation at 
all frequencies. It was not feasible to construct 
elevating gear for this work. To overcome this 
objection to some extent, each curve of polari¬ 
zation error reproduced in this report has a 
section showing the ratio of E n to the E PjZ 
components at the receiving antenna through 
the range of elevation used. From this it may 
be determined whether a specific error was ob¬ 
tained under favorable or unfavorable ground 
reflection conditions, and the extent of dis- 



+ 5 

o 

-5 


o 5 i 
-5 < 


+ 5 < 

o 9 

-5°- 


0 12 3 0 90 180 270 360 

E r/ E p,Z PHASE OF E p REFERRED TO^ 


Figure 14. V-l polarization error measurements 
at 150 me. 


crimination against one or the other field com¬ 
ponent. The third objection is that the informa¬ 
tion obtained covers only a limited range of 
conditions, and does not specify the complete 
performance. 

Although it is true that complete perform¬ 
ance cannot be specified on the basis of the in¬ 
formation obtained, it is considered that the 
range of elevation angles up to 34° covered by 
the data is wide enough to include most of the 


practical conditions of operation likely to be 
encountered. In the upper end of the v-h-f and 
the u-h-f b bands, high-angle waves originate 
generally from elevated sources—aircraft 
transmitters primarily. In homing operations 
of friendly aircraft, for example, angles of ele¬ 
vation over 34° will rarely be found beyond a 
horizontal distance of the order of one mile. 



Figure 15. V-l array, polarization error meas¬ 
urements at 300 me. 

The use of this rotating phase arrangement 
requires eight or more observations at each 
angle of elevation. To expedite measurements 
for day-to-day comparison of results, the sim¬ 
ple elevated dipole, tilted ±45° from the verti¬ 
cal, was resorted to. This method, while not 
giving the maximum error, yielded results of 
sufficient significance to be quite adequate for 
the purpose, and the measurements were readi¬ 
ly repeatable after several days’ lapse. It is in¬ 
teresting to note that the rotating phase meth¬ 
od rather consistently gives a maximum error 


b By rather common agreement the various bands 
are regarded as including the following frequencies: 
v-l-f, 3-30 kc; 1-f, 30-300 kc; m-f, 300-3,000 kc; h-f, 
3-30 me; v-h-f, 30-300 me; u-h-f, 300-3,000 me; s-h-f, 
3,000-30,000 me. 





































74 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


through the range of elevations that is about 
50 per cent in excess of the errors measured 
with the tilted dipole. The latter results are 
given in Tables 1 and 2. The errors obtained 
using the variable phase method at 150 me are 
shown in Figure 14, and at 300 me are shown 
in Figure 15. 

4 - 3,7 Selection of Optimum Height 

The selection of a suitable height for a d-f 
array should be guided by two performance 
considerations, aside from the purely mechani¬ 
cal ones involved in the design of a satisfactory 
elevated rotating mount. 

In the upper end of the v-h-f range, the ten¬ 
dency of electromagnetic waves to propagate 
along optical paths becomes evident, and this 
tendency becomes more marked as the fre¬ 
quency is increased. In the u-h-f range the 
paths are essentially optical. The curvature of 
the earth therefore limits the distance which 
may be covered, since the optical path is a 
straight line. The phenomenon of refraction in 
the atmosphere modifies this condition some¬ 
what, as the path followed curves back toward 
the earth relative to a straight line tangent to 
the earth. Quantitatively the effect may be ac¬ 
counted for by assuming a radius for the earth 
in excess of its actual radius. On this basis and 
the geometry involved, the distance to the ef¬ 
fective horizon is given in terms of the height 
by _ 

wherein the effective radius of the earth is 
taken as 1.32 times the physical radius. The 
obvious conclusion to be drawn is that to obtain 
maximum range, as great a height as practica¬ 
ble should be used for the direction-finder 
array. 

The second consideration influencing the 
choice of height is the effect on polarization 
errors. Due to interference phenomena be¬ 
tween the direct and ground-reflected waves, a 
standing wave pattern of field intensities is set 
up along the vertical line over a point. This 
pattern is different for the perpendicular and 
parallel field components, so that the ratio of 
the two varies with elevation over the point 
in question; therefore relatively large sup¬ 
pression of one or the other component is pos¬ 


sible. The degree of suppression is dependent 
on the elevation, the electrical characteristics 
of the ground, and the angle of elevation of the 
downcoming wave. 

The interference pattern is a result of the 
phase difference existing at a point between 
the direct and reflected waves. This difference 
is made up in part by the phase shift occurring 
at reflection; the remainder is due to the differ¬ 
ence in the paths traveled by the two waves. 
The corresponding phase difference for the 
latter is given by 

4 ?rh sin if/ 


where A is the phase difference in radians, h 
the elevation of the point in question, if/ the 
angle of elevation of the arriving wave, and A 
the wavelength. The difference is seen to in¬ 
crease directly as the height and the sine of 
the angle of elevation. The change of phase 
with if/ is consequently more rapid as h is in¬ 
creased. 

The phase change occurring at reflection is 
given by the appropriate Fresnel plane wave 
reflection coefficient for the parallel and per¬ 
pendicular cases. These are 

^ e sin ^ — V € — 1 + sin 2 ^ 

Rp _ t 

e sin if/ + V e — 1 + sin 2 if/ 

sin if/ — V € — 1 + sin 2 ^ 

n —' . ' — - * 

sin if/ 4- V € — 1 + sm2 t 
Here the complex dielectric constant 
€ = e 0 “ 

e 0 = dielectric constant in esu; 

€i — 2c\cr ; 

c = velocity of light in cm per sec; 

A = wavelength in cm; 

<t = conductivity in esu; 

R p — parallel reflection coefficient; 

R n = perpendicular reflection coefficient. 

Both the magnitude and phase angle of the 
coefficients for the two cases vary in a different 
manner with the angle of elevation. The phase 
angle of R n remains nearly constant, while the 
phase of R p undergoes approximately a 180° 
change as the angle of elevation changes from 
0° to 90°. The overall effect is that the ratio 
of the E n resultant to the E p resultant varies 
through wide limits along a vertical line over 











V-l ARRAY (l DIPOLE PER SCREEN) 


75 


a given point, A plot of this ratio is given in 
Figure 16, against height in terms of wave¬ 
length, for three values of the parameter 
These curves are computed for a complex di¬ 
electric constant of 10 — j 1, and correspond to 
the ground constants at the Medford site for 



Figure 16. Ratio of E n to E pz versus height. 


the low-frequency end of the range (150 me). 
The imaginary component is small enough to 
be neglected, and decreases with increasing 
frequency. The dielectric constant of 10 may 
be taken to represent average ground condi¬ 
tions in the frequency range investigated. An 
examination of these curves leads to two con¬ 
clusions : first, the ratio of the horizontal to the 
vertical field intensities is consistently small 
only at elevations less than A/4. At 600 me this 
represents a height of about 5 inches, and at 
150 me, 20 inches, values too small to be usable. 
For elevations in the usable range, say over 6 
feet, the height would represent several A at 
the higher frequencies. Second, consistently 
small ratios for different elevation angles are 
not possible for a given height over A/4 even 
at one frequency. For example, all three curves 
go through a minimum in the vicinity of 6 a; 
at intermediate or other angles, this would not 
necessarily be the case. Reference to the po¬ 
larization errors of the V and flat arrays shows 
that at the low-frequency end the errors are 
the greatest, and these occur at high elevation 
angles. If one assumes that the maximum 
angles encountered in practice are in the vicini¬ 
ty of 30° to 35°, it would be possible to select 


a height giving favorable ratios near the low- 
frequency end of the band for high angles, but 
the favorable ratios would not hold elsewhere. 
In the absence of elevating gear and means for 
determining the angle of elevation of an arriv¬ 
ing wave, it would appear that a selection of 
height based on maximum range, and com¬ 
pletely random as far as the present considera¬ 
tion is concerned, is as likely to result in satis¬ 
factory operation as would a height selected 
for a particular set of conditions. 

The NBS report 3 has data similar to Figure 
16. The latter is somewhat more general in 
that elevations are given in terms of wave¬ 
length, and the ground conditions specified in 
terms of the complex dielectric constant. Fig¬ 
ure 16 is therefore usable directly at any fre¬ 
quency for a complex dielectric constant of 
10 - jl. 

43 8 Array Gain 

To measure the gain of the V-l array, a 
configuration was required which would elimi¬ 
nate ground reflection effects. A simple man¬ 
ner of achieving this consists of performing 
the measurement in the vertical direction. The 
reflector in question is placed parallel to the 
ground, with the dipole above the reflector. A 
device to indicate relative field intensities is 
placed directly above, and elevated to a height 
great enough to eliminate spurious proximity 
effects. The antenna in question is excited with 
a power oscillator, and the field intensity so 
obtained is compared to that produced by a 
resonant half-wave dipole, A/4 above the screen, 
in exactly the same position. Relative input 
powers are measured by standing-wave equip¬ 
ment for the two cases. The relative gains of 
the two arrays are then computed. 

The absolute gain of the standard antenna 
may be obtained theoretically and, together 
with the relative gain, enables the determina¬ 
tion of the absolute gain of the array being 
measured. Figure 17 shows the gain charac¬ 
teristics of the V-l array over the entire fre¬ 
quency range, the standard of comparison 
being a hypothetical isotropic or nondirec- 
tional antenna. For comparison with a half¬ 
wave dipole in free space, the values given in 
this curve should be reduced by 2.14 db. This 








76 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


figure represents the gain of a half-wave dipole 
in free space over an isotropic antenna. The 
variation of gain with frequency is seen to be 
slow for this array. 



Figure 17. Gain of V-l array compared to hypo¬ 
thetical isotropic or nondirectional antenna. 


4.3.9 y_i ^j-ay Used as Direction Finder 

Following the decision to use a balanced 
dipole system, an array was set up with bal¬ 
anced two-wire lines connecting each antenna 
to the switch, and a twin-conductor lead from 
the switch to the receiver. The performance 
of this system was fair, but it was obvious that 
there was considerable signal pickup due to the 
dipoles and feeders responding as a unit to 
fields between them and ground, and that it 
would be necessary to install the equivalent of 
a balanced and electrostatically shielded trans¬ 
former. A suitable design was selected, utiliz¬ 
ing resonant lines; the principles of operation 
and design formulas are given below. One 
transformer was placed behind each screen at 
the point where the dipole transmission lines 
pass through the screens. Rejection ratios (cor¬ 
rected for curvature of receiver input/output 
characteristics) measured prior to the installa¬ 
tion of these transformers were approximately 
10/1 at 150 me, and 5/1 at 300 me. The use of 
the transformers improved the ratios to about 
40/1 through the frequency range. The mea¬ 
sured maximum polarization errors before in¬ 
stallation were approximately 50°, and these 
were reduced by a factor of two through the 
use of the transformers. The remaining errors 
were considered too high, and further studies 
were undertaken in an attempt to obtain a re¬ 
duction. 

The errors mentioned were noted when the 
system operated as a switched-lobe direction 
finder, that is, one in which amplitude com¬ 


parison is made by successive observations on 
each screen. In order to check the electrical 
balance between the screens simultaneously, the 
two arrays were differentially connected, in 
which case phase and amplitude balance are 
indicated by a null. The errors noted with this 
arrangement were lower by a factor of perhaps 
three-to-one, indicating good electrical balance. 
The reason for this wide discrepancy is not 
completely understood at this time, but is 
mostly likely due to the inherently balanced 
nature of the differential system as compared 
to the dissymmetry existing when only the left 
or right half of the array is observed at one 
time, which condition holds in lobe switching. 

Further investigations tended to confirm this 
explanation. Measurements were made to com¬ 
pare the response of the two halves when the 
polarization of a horizontally incident wave was 
varied. With vertical polarization, the response 
patterns of the two antennas were nearly 
identical; slight differences were attributed to 
the outputs resulting from the E p>x component 
of the ground-reflected wave, adding at differ¬ 
ent phase angles to the respective E p>z voltages 
in the two antennas. When the plane of polari¬ 
zation was rotated clockwise as viewed from 
the receiver, the response of one increased, and 
the other decreased; counterclockwise polari¬ 
zation produced the opposite effect. This effect 
was found to depend on the angle between the 
line of propagation and the normal to the 
screen. When the screens faced the source, the 
effect was a minimum. 

Effect of Support Pole 

An element of dissymmetry appeared to be 
the support pole, and its effect was next in¬ 
vestigated. The transmitter was polarized hori¬ 
zontally, and its output increased sufficiently 
to produce an output in the receiver. With this 
condition, standing waves were noted along all 
the edges of the screen, except in the vicinity 
of the support pole. The edges of both screens 
were then insulated from the pole to obtain a 
more symmetrical potential distribution. A 
decided improvement resulted; the response 
patterns were more nearly alike, and the polar¬ 
ization errors reduced. The same effect was 
observed with the differential connection be¬ 
fore insulating the screens, but while the two 






V-2 ARRAY (2 DIPOLES PER REFLECTOR) 


77 


lobes of the pattern using this connection 
changed in relative size, the position of the 
null remained substantially unchanged. With 
the screen insulated, the resulting pattern 
was symmetrical regardless of the polarization 
of the transmitter. 

Similar observations were made on the V-2 
array, but the effect of insulating the screens 
was much less pronounced; first, because in 
the bearing position the normals to the screen 
lie more nearly along the plane of propagation; 
and second, because the higher gain of the 
array provides better discrimination against 
reradiation effects due to horizontally polarized 
components. 



Figure 18. V-2 array, showing insulators between 

screen and shaft. 


Figure 4 shows the V-l array before the 
screens were insulated from the shaft. The in¬ 
sulating blocks may be seen in Figure 18 which 
shows the final V-2 array. 

44 V-2 ARRAY (2 DIPOLES PER 

REFLECTOR) 

The V-2 array is similar to the V-l in prin¬ 
ciples of operation, the major point of depar¬ 
ture being the use of a broadside array of two 
dipoles on each reflector. Consequently the 
general discussions covering the V-l array are 
applicable here. 

The use of two dipoles as compared with one 
per screen is advantageous in a number of 


respects. The gain is increased through im¬ 
proved directivity in azimuth, while the verti¬ 
cal directivity remains unchanged; the differ¬ 
ential sensitivity is higher; polarization errors 
are reduced; the size of the array is substan¬ 
tially unchanged. Figure 18 is a view of the 
V-2 array with the edges of the screens insu¬ 
lated from the shaft. 

441 Experimental Work 

The first array studied had a spacing be¬ 
tween the line of dipoles and screen equal to 
28.5 cm, the same as was used for the V-l 
array. This spacing was maintained through 
the tests and was considered to be an optimum 
from the standpoint of gain and impedance 
characteristics, although more latitude is avail¬ 
able in this array than in the V-l array. The 
distance between the two dipoles of a screen 
was made 66 cm, or approximately one-half 
wave near the middle (225 me) of the frequen¬ 
cy range. A set of operational data was ob¬ 
tained including polar patterns, polarization 
errors, and gain. The data indicated that this 
array was considerably superior to the V-l 
array primarily because of the improved direc¬ 
tivity in azimuth. To increase the directivity 
further, the spacing between dipoles was in¬ 
creased to 86 cm, representing a half wave¬ 
length at 175 me, without changing the screen 
dimensions. Observations were made with this 
spacing, the maximum that the screen will ac¬ 
commodate and still have the required one- 
eighth wave projection beyond the dipoles. 
Further increase is not usable, since the spuri¬ 
ous side lobes at the high-frequency end be¬ 
come troublesome. Comparative data on the 
V-l, V-2 (66 cm), and V-2 (86 cm) arrays are 
considered in Tables 1 and 2. As in the V-l 
array, both switched-lobe and differential oper¬ 
ation were investigated. 

4 - 4,2 Relative Response in Azimuth 

Polar diagrams showing the relative azi¬ 
muthal response for one-half of the V-2 array 
with 86-cm spacing are given in Figures 19 
and 20 for 150 and 300 me, respectively, at 0° 
elevation. The increased directivity over the 
V-l array is clearly evident in these diagrams. 











78 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


Table 1. Comparative polarization errors, V-l and V-2 arrays. 





Switched-lobe connection 


Differential connection 





(Error in 

degrees) 




(Error in 

degrees) 


Angle of 












elevation 












in degrees 

Array 

V-l 

V-2 (66 cm) 

V-2 (86 cm) 

V- 

-1 

V-2 (66 cm) 


Polarization 

-45° 

+45° 

-45° 

+45° 

-45° 

+45° 

-45° 

+45° 

-45° 

+45° 

0 


-3 

-3 

+ 10 

-3 

+2 

-3 

+0.75 

+ 1.5 

0 

+ 1 

5 


-3 

—4 

+ 1 

0 

+ 1 

-2.5 

+0.5 

+3.0 

-0.5 

+2 

10 


-6 

-1 

-3 

+5 

-4 

+2 

+0.5 

+3.0 

+ 1 

+ 1 

15 

150 me 

—5 

+2 

+ 12 

-4 

+4 

-5 

+0.5 

+0.5 

0 

0 

20 


-11 

+ 1 

+2 

+ 1 

+ 1 

-0.5 

+0.5 

+2.5 

-1 

+ 1 

25 


-15 

+3 

-7 

+7 

-4 

+4 

+ 1.0 

+4.0 

0 

+2 

30 


-16 

+ 13 

+ 12 

+4 

+2 

0 

+4.5 

+2.0 

+3 

0 

34 


-10 

+7 

+ 12 

0 

+5 

-3 

+ 1.5 

+2.0 

-0.5 

0 

0 


-0.5 

+ 1 

-0.5 

0 

+0.25 

+0.5 

0 

+0.5 

+0.25 

+0.25 

5 


-3 

+2 

-0.25 

-0.25 

-0.75 

0 

+0.2 

+0.2 

0 

0 

10 


-1 

+2 

-1.0 

+0.5 

-0.5 

+ 1 

0 

-0.2 

0 

+0.5 

15 

300 me 

-0.5 

+3 

-0.5 

+0.25 

-0.75 

+ 1 

0 

0 

0 

+0.5 

20 


+0.5 

+ 1 

0 

0 

— 1 

+0.75 

+0.2 

-0.2 

0 

0 

25 


+ 1 

+2.5 

0 

-0.25 

-0.5 

+ 1.25 

+0.5 

-0.2 

-0.5 

0 

30 


-3 

+4 

-4 

+0.5 

— 1.5 

+2 

+0.6 

+0.5 

-1.0 

+0.25 

34 


-3 

+9 

-2 

+ 1.0 

— 1.5 

+2 

-2.5 

-3.5 

0 

0 


Table 2. Summary of maximum errors. 



Switched-lobe connection 

Differential connection 



(Error in degrees) 


(Error in degrees) 

Frequency 






V-l 

V-2 (66 cm) 

V-2 (86 cm) 

V-l 

V-2 (66 cm) 

150 me 

16 

12 

5 

4.5 

3 

300 me 

9 

4 

2 

3.5 

1 


The directivity increases with frequency over 
the range illustrated; this is in opposition to 
the behavior of the V-l antenna, where maxi¬ 


mum directivity occurs at the low-frequency 
end. Response diagrams for 66-cm spacing are 
not given; these are very closely similar to the 


30° 20° 10° 0° 350° 340° 330° 



300° 


270° 


120° »50° 210® 240° 

Figure 19. V-2 array, relative response in azi¬ 

muth, 0° elevation, 150 me. 


30° 20° 10° 0° 350° 340° 330° 



Figure 20. V-2 array, relative response in azi¬ 

muth, 0° elevation, 300 me. 















































































V-2 ARRAY (2 DIPOLES PER REFLECTOR) 


79 


ones shown, but are slightly broader. The pres¬ 
ent diagrams were obtained before the screen 
angles were adjusted for the optimum position. 
This may be noted in the 300-mc diagram 
where the response on bearing is too low. The 
patterns should, therefore, be rotated approxi¬ 
mately 6° toward the zero azimuth line to cor¬ 
respond to optimum setting. The relative re¬ 
sponse in elevation for the lobe-switching con¬ 
nection is given in Figure 11. 

443 Impedance Characteristics 

Figure 21 is a plot of the impedance charac¬ 
teristics of the V-2 array at the balanced- 
unbalanced transformer, and includes the effect 
of the latter, as well as of the transmission 



140 180 220 260 300 340 380 


FREQUENCY-MEGACYCLES/SEC 

Figure 21r V-2 array impedance characteristics. 

Dipoles spaced 86 cm (X/2 at 175 me) ; spacing to 

reflector is 28.5 cm (X/4 at 263 me). 

lines between it and the dipoles. The impe¬ 
dance is comparatively uniform through the 
140- to 300-mc frequency range, with the reac¬ 
tive remaining less than the resistive component 
through the range. The geometric mean of the 
minimum and maximum points is approxi¬ 
mately 57 ohms; therefore, standard 60-ohm 
cable may be used without additional matching 
transformers. The impedance mismatch when 
using 60-ohm cable does not exceed two-to-one, 
and is considerably less through most of the 
range. 

4 - 4 - 4 Polarization Errors 

Comparative data on polarization errors of 
the V-l and V-2 arrays are given in Table 1 for 
both the switched-lobe and differential connec¬ 


tions. (The latter connection is more fully 
discussed later in this chapter.) These were 
obtained with a tilted dipole transmitter. The 
tilt in this test is always about a horizontal 
line lying in the plane of incidence, i.e., the 
dipole always lies in a vertical plane normal to 
the plane of incidence, and, therefore, the E p 
component of the downcoming wave is less 
than the E n component at elevated angles. The 
ratio of the two is nearly proportional to the 
cosine of the angle of elevation; i.e., unity at 
horizontal incidence, and dropping to 0.83 at 
34° elevation. It is to be noted that the differ¬ 
ential connection is better by a factor of three 
or four to one compared to the corresponding 
switched-lobe array in regard to polarization 
errors. Nevertheless, the errors of the V-2 ar¬ 
ray using 86-cm spacing between the dipoles 
are quite low for lobe-switching operation. For 
rapid comparison, Table 2 lists the maximum 
errors found in Table 1. 

It is evident from these tables that there is a 
progressive improvement in polarization error 
performance as the azimuthal directivity is in¬ 
creased. Consequently it is reasonably safe to 
predict that still greater improvement is pos¬ 
sible if arrays of greater directivity are used. 
Because of size, their use would probably be 
limited to permanent or semipermanent instal¬ 
lations. It is pertinent to observe that as a 
result of increased directivity, searching be¬ 
comes more difficult, since high response is 
limited to a narrower azimuthal sector. Exces¬ 
sively directive arrays may require the use of 
subsidiary searching equipment. 



Figure 22. Gain of V-2 array. 

445 Gain of V-2 Array 

Measurements of gain on this array were 
made at both the 66-cm and 86-cm spacing, and 
are given graphically in Figure 22. The im- 








80 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


provement obtained by the wider spacing is 
greatest at the low-frequency end; at the high- 
frequency end the appearance of side lobes 
limits the possible improvement. As in the case 
of the V-l array, the standard of comparison 
in the curves is a nondirectional or isotropic 
antenna. For comparison with a half-wave 
dipole in free space, the gain taken from these 
curves should be decreased by 2.14 db. 

4.5 FLAT array 

451 Theory of Operation 

The principle of operation of the flat array 
in which the directivity pattern is shifted in 
azimuth by a change in phase of some of the 
elements may be readily seen from the follow¬ 
ing considerations: If in Figure 23 we have 
two electric doublets, 1 and 2, in a radiation 

A 

* 

/ 

/ 

/ 
s 

/ 




Figure 23. Representation of doublet in radiation 
field. 


field propagated from a direction A, at an angle 
0 from the normal to the line joining the cen¬ 
ters of the two doublets, the voltage induced 
in each is in phase with the field at the doublet. 
If we consider the wave front to be plane, the 
arrival of the wave front at doublet 2 occurs 
later than the time of arrival at doublet 1, be¬ 
cause of the finite velocity of propagation of 
electromagnetic waves. Hence, the phase of the 


'© 

i 


induced voltage in doublet 2 lags the voltage in 
doublet 1 by fix, where x is the additional dis¬ 
tance traveled, and /3 is the phase constant of 
free space, equal to 2?r/A, A being the free space 
wavelength. It is convenient to refer phases to 
the field at 0, the center of the line joining the 
two doublets. Then if d is the distance from 
this center to either doublet, the voltage in¬ 
duced in 1 leads the field at 0 by an angle 2?r d 
sin 0/A, while that in 2 lags by the same angle. 
A vector diagram is shown in Figure 24, where 



Figure 24. Vector diagram of doublet voltages. 


E, is the field at 0, V x is the voltage induced in 
doublet 1, and V 2 the voltage in doublet 2. 

Either the vector sum or difference of these 
two voltages, which are also shown in the 
diagram as V sum and V dit f f may be utilized. The 
salient fact revealed in this diagram is that the 
sum voltage is in phase with the field at 0, 
while the difference voltage is displaced 90° in 
time-phase from both E 0 and the sum voltage, 
for equal magnitude component voltages V x 
and Vo. 


j 

( 2nd 

sin 0^ 

| + j sin 

(2 Ttd . , \ 


COS 1 



1 

j 

(2* d 

sin 0^ 


f 2 wd . , \ 

l 

cos I 


1 — J sin ( 

K — Sln +) 

1 


where A; is a proportionality factor. Therefore 
^um = 2kE » cos sin 4,) 

v dis = j2kE a sin sin <*>) • 

The sum voltage is thus real, and in phase 
with the field E 0 , while the different voltage is 
imaginary, and therefore displaced 90° from 
E 0 and V sum . 








FLAT ARRAY 


81 


Typical directional patterns for the differen¬ 
tial and additive cases are shown in Figure 25 
B and C for the case of spacing between the 
doublets of the order of a half Wavelength. 

In the differential case, the resultant voltage 
is zero when <f> is zero, while in the additive 

v sum 

v d iff 


A B C 

Figure 25. Directional patterns for doublets sepa¬ 
rated X/2; B shows differential voltage pattern; C 

shows sum voltage pattern. 

case the voltage is a maximum. In both cases, 
there is a reversal in phase where the resultant 
passes through zero. The axes of maximum 
response for the two cases are displaced from 
each other 90° in azimuth. A 90° phase change 
introduced in the output of either one or the 
other will bring corresponding lobes in phase, 
but will not change the space pattern. There¬ 
fore the voltages from two pairs, one additive, 
and the other differential, may be added, and 
the resultant space pattern will be rotated in 
azimuth. In Figure 26, A shows two such pairs, 
disposed along the same line; at B is shown 
the pattern of the differential pair, with an 
advance in phase of 90° introduced in its out¬ 
put, while at C the sum pair is shown un¬ 
changed. The resultant pattern at D has its 
line of maximum response along a line inter¬ 
mediate between the lines of the individual 
maxima. If we consider the axis of the sum 
pattern as a reference direction, then the re¬ 
sultant pattern has been rotated clockwise. 
Obviously a reversal in phase of either pair 
will rotate the resultant counterclockwise by a 
like amount. 

It is interesting to observe a close similarity 
between the action of this system and that of 
switched cardioids, obtained, for example, by a 
loop and sense antenna. In the latter case the 
voltage induced in the loop is proportional to 
the time derivative of the magnetic field, and 




since this latter is in phase with the electric 
field, the induced voltage is displaced 90° in 
phase from the electric field. The sense anten¬ 
na voltage is in phase with the electric field. 
Or, alternatively, the vertical members of the 
loop may be considered electric doublets, differ¬ 
entially connected by means of the horizontal 
members, yielding the same result. 

The use of a reflector behind the line of 
doublets removes one lobe of the response pat- 

V diff ^90° 





Figure 26. Directional patterns of two sets of 
doublets in line with voltage of differential pair 
advanced 90°. 


tern, leaving one point of intersection when the 
pattern is alternately rotated clockwise and 
counterclockwise. This intersection represents 
equal response to the same wave by each array 
and may be used as a bearing indication. 

t 

452 Physical Arrangement 

The actual collector system developed follows 
basically the scheme outlined above. Since a 
wide band is covered by the antenna system in 
question, quantities given in terms of wave- 






82 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


length refer to the wavelength at the arithmetic 
mean frequency unless otherwise specified. Fig¬ 
ure 27 is a schematic diagram of the array. 
For reasons of symmetry, one pair of dipoles 
is placed between the dipoles of the other. On 
the figure, the outer pair, spaced one wave¬ 
length, are differentially connected, while the 
inner pair, spaced a/ 2 apart, are connected 
additively. All dipoles are placed 28.5 cm, or 
approximately A/4 from the reflector at 260 
me. Balanced feeders run from each dipole to 



the screen, and through the screen to the 
switch or transformer, as the case may be. 
Switching may be accomplished in either pair; 
in the final experimental model the outer pair 
were switched. The feeders to the outer pair 
exceed in electrical length the ones to the inner 
pair by approximately A/4 to introduce the 
phase lag of 90° required. 

4.5.3 Choice of Electrical Elements 

To cover a two-to-one frequency range satis¬ 
factorily, the electrical characteristics of the 
various elements making up the array must be 
carefully chosen. Brief considerations will in¬ 
dicate the large number of parameters, each of 
which individually affects the performance, and 
many of which are interdependent. Theoreti¬ 
cally, for nondirectional “point source” ele¬ 
ments, the array will produce an ideal direc¬ 


tional pattern having no spurious response 
lobes when the phase shift introduced artificial¬ 
ly is exactly 90° and the amplitudes utilized 
from the center pair and outer pair respectively 
bear a ratio of two to one. Since it was con¬ 
sidered undesirable to control the relative am¬ 
plitudes of the two pairs, investigation indi¬ 
cated that good results could be obtained with 
a one-to-one amplitude ratio, allowing the ideal 
90° phase shift to change with frequency from 
60° at the low-frequency end, through 90° at 
the center frequency, to 120° at the upper end 
of the frequency band. Limiting the phase shift 
to this range of values and maintaining a one- 
to-one amplitude ratio through the frequency 
range, presupposes resistive dipole elements 
matched to the transmission lines, with no mis¬ 
match at the junction or other points of the sys¬ 
tem. 

Reference to dipole impedance characteris¬ 
tics, Figure 6, obtained during the development 
of the V-l antenna system will show that it is 
impossible to obtain a uniform resistive charac¬ 
teristic over the frequency range. While quite 
good standing-wave ratios are obtainable using 
one or more dipoles feeding suitable lines in the 
V-l system, where phase shift is of secondary 
consequence, in the present case, where spuri¬ 
ous phase shifts may make the system inopera¬ 
tive, attention must be given to all factors 
which can contribute to phase and amplitude 
variations. Since the presence of mutual radia¬ 
tion impedance between a dipole and its image, 
and between two dipoles tends to increase the 
variation, over a range, of the total impedance 
of a dipole, spacings between dipoles and from 
the reflector must be chosen to keep these 
mutual impedances at as low a value at the 
lowest frequency used as is consistent with 
other requirements. This means that spacings 
between dipoles and from dipole to screen 
should be large in terms of wavelength at the 
lowest frequency used. 

Conflicting with this requirement is the 
phenomenon of spurious response lobes ap¬ 
pearing at the high-frequency end of the band 
when spacings are of the order of one wave¬ 
length or more. The choice of these spacings 
must therefore be a compromise based on these 
two limiting factors. 

































FLAT ARRAY 


83 


For the same reasons, the self-impedance of 
the dipoles should be as uniform as possible 
and essentially resistive. The dipole length to 
diameter ratio (7.5/1) used in the V-l system, 
offers a fair approximation to the ideal con¬ 
dition. A better approximation is not possible 
without increasing the diameter to a size con¬ 
sidered excessive for portable use. For fixed- 
station direction finding, however, modifica¬ 
tions along these lines should produce a col¬ 
lector system capable of more uniform per¬ 
formance through a two-to-one frequency band. 

Transmission Lines 

At the center frequency, the transmission 
lines to the outer pair exceed the inner lines by 
nearly one-quarter wave in order to produce 
the required phase shift. As outlined previous¬ 
ly, the ideal phase difference is 90°. A quarter- 
wave excess in tranmission lines produces this 
phase difference when no impedance mismatch 
occurs in the system. When terminated by ac¬ 
tual dipoles, however, whose impedance varies 
through the band and is partly reactive except¬ 
ing at a few points in the band, this condition 
does not hold. Further, the quarter-wave excess 
produces a transformation in impedance in ad¬ 
dition to that occurring in the shorter line. 
Therefore, at the junction point, the impedance 
presented by one set of lines, terminated by its 
pair of dipoles, is generally different from that 
of the other. As a result, both the phase and 
amplitude of the two currents in the load are 
modified by an undesired amount. This last 
factor should, however, be qualified to this 
extent, that the amplitude modification may be 
in a direction to approach a two-to-one ratio, 
which is preferable to the one-to-one ratio, and 
also, when operating away from the center 
frequency, where the nominal phase shift is 
more or less than 90°, the change may tend 
toward the 90° value desired. Both, of course, 
may move in the wrong direction. A result 
which is unqualifiedly desirable is that at the 
center frequency, the impedance transforma¬ 
tion of the lines to one pair of dipoles is the 
inverse of the transformation in the lines to the 
other pair, referred to the characteristic im¬ 
pedance of the line. This means that the re¬ 
actances are of opposite sign and at least par¬ 
tially cancel. Off the center frequency, while 


the transformations are not exactly inverse, 
they are nearly so, and reactance cancellation 
still occurs. The impedance of the system as a 
whole consequently has small phase angles 
through most of the range, as shown in Fig- * 
ure 28. 



Figure 28. Flat array impedance characteristics. 

These considerations should make it evident 
that in addition to the excess line in one 
branch, the characteristic impedance and total 
length of lines must be properly chosen. The 
number of impedances available in standard 
solid dielectric low-loss high-frequency lines is 
limited. The lines used have an effective im¬ 
pedance that approximates the geometric mean 
of the impedance range of each dipole; this 
minimizes the variation of the transformed 
impedances. The total length of lines should 
be kept as low as possible to minimize losses 
and undesired pickup, with this reservation, 
however, that they should be so selected as to 
avoid quarter-wave transformations at points 
where the dipole impedance departs farthest 
from the characteristic impedance of the line. 
This is particularly the case at the low-fre¬ 
quency end of the band. Should a quarter-wave 
transformation occur here on one of the lines, 
the other may be near a half-wave transforma¬ 
tion point; the latter will remain substantially 
unchanged, while the former will be raised in 
impedance by a factor of perhaps three or 
more. If this happens to be the center pair, its 
output current will be reduced by a factor of 
three or more, thus departing by a factor of 
six from the ideal two-to-one ratio. 

The physical disposition of the elements of 
the array sets a lower limit on the usable 






84 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


length of transmission lines. This minimum is 
somewhat more than one meter from the trans¬ 
former, through the switch, to each outer 
dipole. Since the velocity of propagation in the 
50-ohm polyethylene cable used is approximately 
64 per cent of the velocity of light in free 
space, the actual electrical length is greater 
than the mechanical length by a factor of ap¬ 
proximately 1.6. For this reason, air dielectric 
lines could possibly be used to advantage. 
Furthermore, the attenuation factor of air di¬ 
electric lines is generally lower, and more lati¬ 
tude is available in the choice of characteristic 
impedance. Due chiefly to the ease of adjust¬ 
ment to length, the experimental work on this 
array was completed using only the solid di¬ 
electric cable mentioned. The actual lengths 



LENGTHS OF SHIELDED 50 OHM 
POLYETHYLENE CABLE 
d = 9 CM 

b = 105 CM 
C = 104 CM 

Figure 29. Cable lengths employed in connecting 
flat array. 

used are given in Figure 29. The electrical dif¬ 
ference in length is approximately 77°, this 
figure producing the best performance through 
the range as determined experimentally. A 
word of caution is appropriate at this point 
concerning line lengths. Should it be desired 
to duplicate this array, the electrical length of 
the lines must be accurately set. While not 
critical the adjustments should be made to 
within i /2 cm or less. The velocity of com¬ 
mercial cable varies between different runs. All 
commercial cable should, therefore, be meas¬ 
ured, and cable preferably from the same run 
be used on one array. 


BALANCED-TO-UNBALANCED TRANSFORMER 

The transformer for converting the balanced 
system to unbalanced feed is similar to those 
used in the V-l array. A single transformer is 
quite satisfactory. While reactance cancella¬ 
tion by means of a half-wave series line is pos¬ 
sible as in the case of the V-l array, it may be 
omitted here with very little change in overall 
performance. The reason for this is that in 
effect four dipoles are paralleled (after impe¬ 
dance transformation by their individual lines) 
at the transformers, resulting generally in a 
lower effective impedance than the individual 
dipoles have; the characteristic impedance of 
the quarter-wave transformer lines is high in 
comparison with this, and is increased by a 
factor equal to the tangent of the phase length, 
so that the effective shunt reactance is high, 
resulting in a low equivalent residual series 
reactance. 

Reflector Dimensions 

The dimensions of the screen used for this 
array are 120 cm high and 188 cm wide. The 
spacing between adjacent vertical elements is 
the same as used in the V-l array, namely, 
about a/20 at the highest frequency covered. 
By substituting fine mesh high-conductivity 
screen, this spacing was found to be adequate 
in that array. The overall size is about the 
minimum that can be satisfactorily used. Some 
improvement in gain at the low-frequency end 
is possible by increasing the reflector size. For 
sizes smaller than used, the pattern broadens 
considerably, resulting in lowered gain. 

Relative Response in Azimuth 

The performance of this system compares 
favorably with that of the corner type using 
an array of two dipoles per screen. Response 
patterns for the flat array, at 140 and 300 me, 
are given in Figures 30 and 31. While the pat¬ 
terns exhibit considerable variation through 
the band, as compared to the corner type, the 
intersection points of overlapping lobes are 
satisfactory. The adjustment of the system in 
this respect is very much more restricted than 
in the V type, where a mere change in the 
screen angle changes the intersection point. 
While at any one frequency the pattern may 
be changed over wide limits by adjustment of 























SWITCHING AND INDICATING DEVICES 


85 


the line lengths, this process also changes the 
pattern through the rest of the range in a dif¬ 
ferent manner. The intersection points should 


3 0“ 20® 10“ 0° 350“ 340“ 330“ 



Figure 30. Flat array, relative response in azi¬ 
muth at 140 me. 


be considered fixed, unless variable controls are 
incorporated in the system, and this was ruled 
out in setting the preliminary scope of work. 


30“ 20“ 10“ 0“ 350“ 340° 330“ 



Figure 31. Flat array, relative response in azi¬ 
muth at 300 me. 


Polarization Errors 

The polarization errors of the flat array are 
quite low. The method of presenting polariza¬ 
tion error data is the same as used in the V-l 
array. On each graph showing the error under 
various conditions is placed a curve showing the 
horizontal-to-vertical field ratios through the 
range of elevations used. The maximum error 
observed at 150 me is 6.5°; at 800 me the maxi¬ 
mum error is 3.5°. 


Impedance Characteristics 

As indicated above, the approximately in¬ 
verse transformations occurring in the trans¬ 
mission lines to the outer and inner dipoles, 
produce a comparatively high degree of reac¬ 
tance cancellation. The resulting impedance of 
the system, as seen at the transformer, and in¬ 
cluding the effect of the latter, is quite uni¬ 
form, and has small phase angles through most 
of the range. Reference may be made to Fig¬ 
ure 28, which gives the impedance and the 
resistive and reactive components through the 
frequency range. 

Gain of the Flat Array 

As may be inferred from an examination of 
the relative response patterns, the gain of the 
flat array is less uniform through the band 
than the gain of either the V-l or V-2 arrays. 
The variation is cyclic, but its magnitude is 
not large enough to be serious. Figure 32 gives 



Figure 32. Flat-array gain characteristic com¬ 
pared to that for nondirective antenna. 


the gain over the range, compared, as in the V 
arrays, to a nondirective, or isotropic antenna. 
For comparison with a half-wave dipole in free 
space, figures obtained from this curve should 
be reduced by 2.14 db. 

4-6 SWITCHING AND INDICATING DEVICES 

461 Switches 

To switch between antennas of the V arrays 
and to obtain the required phase reversal in 
the flat array, a motor-driven switch was 
developed that has electrical characteristics 
similar to the transmission lines so that dis¬ 
continuity of impedance and the resulting re¬ 
flections are minimized. The desired charac¬ 
teristics are obtained by adjustment of capaci¬ 
tance per unit length to the required value. 














86 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


The type of switch employed is shown in 
Figure 33 and consists of two moving contact 
members and four fixed contacts. The moving 
contacts are driven through an eccentric ball¬ 
bearing race. The fixed contacts are adjustable 
so that adjustments may be made which permit 



Figure 33. Motor-driven r-f switch used to obtain 
phase reversal in flat array. 


closing the r-f section before closing the meter 
contacts. Adjustments are also possible which 
permit opening the r-f circuit before the indi¬ 
cator circuit. This arrangement was found nec¬ 
essary to eliminate transients in the meter be¬ 
cause of the antenna make-and-break. The bear¬ 
ings of each moving contact are clamped in 
rubber pads between bakelite blocks in order to 
minimize chatter. 

Previous experience in the construction of a 
switch for similar functions showed that the 
selection of the correct contact material was 
important. Silver, gold, iron, and several other 
metals and alloys proved unsatisfactory where 
extremely low r-f currents were to be broken, 
even though fair contact pressure was avail¬ 
able and the contacts were mechanically wiping. 
The most satisfactory material found, and one 


which operates for long periods without trouble 
from varying resistance, is rhodium. 

As used on the V array the switch consists 
of two sections. The first section switches the 
leads from each side of the array to the receiver 
line and simultaneously disconnects the unused 
half of the array and grounds it by means of 
back contacts. The second section of the switch 
consists of a mechanically similar unit con¬ 
nected to operate as a single-pole single-throw 
switch to couple the receiver output to the indi¬ 
cator bridge. 

When used with the flat array, the first 
section of the switch was modified to become a 
double-pole double-throw unit with the back 
contacts insulated from ground and utilized as 
shown schematically in Figure 27. 

The motor used to operate the switch has a 
12-volt universal winding, coupled to the 
switches through a ten-to-one reduction gear. 
Speed is controlled by a Variac in series with 
the primary of the supply transformer. Nor¬ 
mally the switch is operated at a speed between 
five and ten cycles per second. Limitations in 
the maximum speed are purely mechanical. The 
indicator damping and desired responsiveness 
set the lower limit to the usable speed. 

The shafts of the switch sections are linked 
together through Oldham couplings. These 
allow removal of individual switch sections for 
repair or adjustment and permit proper re¬ 
placement of the switch without the necessity 
of resynchronizing, or the reorientation of the 
shafts, since the latter can be reassembled only 
in the desired position. In addition, this type 
of coupling takes up misalignment of shafts. 

The motor leads are unshielded and run 
through the hollow aluminum antenna drive 
shaft without r-f filtering. The r-f interference 
caused by sparking motor brushes appeared to 
be entirely absent at the frequencies used, and 
no trouble was encountered from this source. 

462 Indicators 

The indicator used in the majority of tests 
consists of a simple zero-centered, 100-micro¬ 
ampere d-c meter having rather high electrical 
damping. The scale is marked off with the 
letters L-O-R, indicating the direction in which 



COMPARISON BETWEEN V AND FLAT ARRAYS 


87 


the array should be rotated in order to obtain 
the bearing. 

The connections to the indicator from the re¬ 
ceiver and switch are shown schematically in 
Figure 34. The use of the capacitors C x and C 2 
in place of a resistor network is advisable as it 
allows considerable latitude in the adjustment 
of the switch contacts. The dwell periods do 
not need to be equal with this arrangement 
since it operates similarly to a peak voltmeter. 



Figure 34. Connection of indicator to receiver 
and switch. 


Capacitor C a is used to stabilize the indicator 
and prevents the pointer from responding to 
the low-frequency switching rate. With this 
type of indicator it is necessary that the re¬ 
ceiver furnish an audio-frequency output that 
in turn is rectified by the rectox unit. A beat- 
frequency oscillator in the receiver would be 
desirable to furnish the audio frequency, but 
due to the inherent instability of the receiver 
r-f oscillator and many of the transmitter 
carriers operating at these frequencies, the use 
of a beat-frequency oscillator is limited. The 
particular receiver used is equipped with an 
audio oscillator that modulates the inter¬ 
mediate frequency and furnishes a tone output 
from a c-w carrier input. 

When receiving radar, pulsed at audio fre¬ 
quencies, it is not necessary to use the a-f 
heterodyne oscillator, providing the repetition 
rate of the transmitter is sufficient to produce 
a fair amount of a-f output. The direct current 
to operate the indicator could have been direct¬ 
ly obtained from the a-c line. In this case it 
would not have been necessary to modulate lo¬ 
cally the c-w carrier but it would have been 
inconvenient in using the receiver for other 
measurements requiring fixed gain. Although 
methods of indication were not a part of the 
problem, there are several others which may 
be used advantageously with these arrays. 
These are described below. 


47 COMPARISON BETWEEN V AND 
FLAT ARRAYS 

The advantages and comparison of the two 
types of arrays as observed during their devel¬ 
opment may be summarized as follows: 

The V array, using two dipoles (spaced 86 
cm) per screen, is electrically a satisfactory 
unit possessing good directivity and reasonably 
low polarization errors which may be further 
improved by careful balance. The directivity is 
not confined to a narrow sector so that there is 
little possibility of losing a desired signal 
located within a known sector of at least 120°. 

The construction of the dipoles and trans¬ 
mission line system is such that an accurate 
balance between the two halves of the array 
may be readily obtained. This balance can be 
maintained for long periods of time without re¬ 
adjustment. 

The intersection point of the switched lobes 
may be chosen by setting the screens to the 
desired angle, and this point remains reason¬ 
ably constant throughout the frequency range 
as the polar patterns are not subject to sudden 
changes with frequency. 

The bearings are sharp, and with interlock¬ 
ing of the lobes at the angle of 17.5° from the 
lobe maximum, i.e., with the internal angle be¬ 
tween screens set at 145°, the bearings are 
repeatable to approximately V 2 0 throughout the 
frequency range down to a signal input equal 
to one-half the receiver noise, measured at a lobe 
maximum. 

There are no reversals in bearing throughout 
the frequency range and “sense” is therefore 
unmistakable. If a bearing should be obtained 
using the back of the screen, i.e., at 180°, it is 
readily noticed for two reasons: 

1. The action of the indicator is reversed. 

2. The amplitude of the received signal is 
greatly reduced. 

Mechanically this array appears to be best 
suited to locations which are semifixed in 
nature and where space is not of great impor¬ 
tance. This is due to the wide turning radius 
required for the 150- to 300-mc array. If de¬ 
signed for higher frequencies, the array size is 
proportionately decreased throughout and be¬ 
comes suitable for portable use. 

The large array as used during the tests 















88 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


proved somewhat awkward to handle in a high 
wind. This could have been remedied by plac¬ 
ing the apex of the screens somewhat ahead 
of the supporting rotating member, thereby 
improving the dynamic balance. 

The addition of a 300- to 600-mc array at¬ 
tached to the back of the large array, forming a 
diamond-shaped section as viewed from the 
top, would also tend to improve the balance and 
decrease the weather-vane action. 

The advantages of the flat array lie in its 
smaller size, greatly improved rotational bal¬ 
ance, and ease of operation. 

The bearings are sharp throughout the band 
and may readily be repeated to better than one- 
half a degree, a slight improvement existing 
between the sharpness of this array at cer¬ 
tain frequencies and that of the V array. 

The average polarization error is slightly 
lower than that obtained with the V array, and 
in general, this system appears to be a prefer¬ 
able type for operation on signals which have 
a reasonable length of transmission period. The 
reason for this latter qualification is that at 
some frequencies the lobes are quite sharp, and 
unless the array is oriented within a few 
degrees of one of the lobe maxima, the signal 
may not be picked up. In addition, at a num¬ 
ber of frequencies there are reversals of indi¬ 
cation that are symmetrically located on either 
side of the true bearing. The reversals are 
caused by the way in which the lobes overlap, 
or in some instances do not fully overlap, a 
spurious side lobe. In general, a false indica¬ 
tion is readily detected either by the amplitude 
of output, which is relatively weak at the re¬ 
versal point, or more accurately by reversal of 
indication. The use of the cathode-ray indica¬ 
tor removes all ambiguity. 

Since the polar patterns change rapidly with 
changes in frequency due to a multiplicity of 
effects resulting from phase shift caused by 
the electrical changes in spacings and trans¬ 
mission-line linkage, it is not possible to locate 
the optimum cross-over point of lobe intersec¬ 
tions at more than a few frequencies. At the 
remaining frequencies the intersections fall 
where they may, although with the antenna 
spacings and lines cut to the dimensions shown, 
the performance approaches the maximum ob¬ 
tainable over the frequency range and does not 


depart greatly from the optimum or desired 
performance except at the higher end of the 
frequency range, where the intersection of the 
lobe drops to an amplitude somewhat below 
that desirable for optimum signal-to-noise 
ratio. The effect on bearing sensitivity in this 
case is to increase the angular sensitivity to 
the detriment of the radio frequency sensitivity 
as indicated by the calculations for optimum 
performance. 

In either type of array the use of two coaxial 
shielded flexible cables appears to have an ad¬ 
vantage over the more commonly used spaced 
air-dielectric twin pairs. This is apparent elec¬ 
trically from the good degree of balance that 
may be obtained by simply cutting to the same 
mechanical length leads that are to be matched. 
The uniformity of cable, of reliable manufac¬ 
ture, obtained from the same reel, is sufficient 
in most cases for one to be reasonably sure of 
better than passable matching. This was elec¬ 
trically measured and checked several times in 
the course of changes and development. It is 
believed that such r-f cable is adaptable to 
feeders for the elevated H Adcock-type anten¬ 
na, where strict symmetry and balance are re¬ 
quired. 

COMPARISON BETWEEN DIFFER¬ 
ENTIALLY CONNECTED SCREEN ARRAYS 
AND H ADCOCKS 

Some data were obtained using the V-l ar¬ 
ray differentially connected, and the V-2 array 
with each pair differentially connected to the 
opposite pair, the dipoles of each reflector 
being connected in phase. The latter connection 
is indicated in Figure 35. This construction is 
interesting because, while it resembles an H 
Adcock, the screen angle is such that the spuri¬ 
ous side lobes normally obtained with multiple 
dipoles on a flat array are absent due to the 
fact that at wide angles from the null, one an¬ 
tenna is shielded by the screen from the signal 
source and hence the system no longer acts as 
a differential system. 

It is difficult to make quantitative comparison 
between this differentially connected screen ar¬ 
ray and the more common elevated H Adcock 
which it resembles without a side-by-side check 
using field-intensity equipment. However, it is 




COMPARISON BETWEEN SCREEN ARRAYS AND H ADCOCKS 


89 


possible to indicate certain generalities and 
limitations. 

The type of Adcock to be considered as a 
reference is of the balanced elevated H design 
most commonly used on these frequencies. The 
selection of dipole length and spacing would 



Figure 35. Differential connection of elements of 
V-2 array. 


vary slightly with the designer’s choice, but a 
normal unit would have a dipole spacing such 
that, at the minimum wavelength, the spacing 
would not exceed A/2. In any case, the choice 
of spacing would be such that the polar pat¬ 
terns would not divide into more than two 
lobes. Four lobes, such as would appear at a 
spacing of A would, of course, be unusable as 
there would be no rapid way to distinguish 
which of the four bilateral minima would be 
correct. Therefore the conventional spacing 
would be such as to give a pattern approaching 
a cosine curve, and might be in the order of 
A/6 to a/2, giving at the smaller spacing a 
maximum response equivalent to that of a 
single dipole in free space, and at the greater 
spacing a maximum response double this value. 

If, however, single dipoles of length equal to 
the above Adcock but of suitable diameter are 
arranged at an appropriate distance in front 
of a V screen and these dipoles are differential¬ 
ly connected, there are three important results. 
First, the gain of the dipoles is increased in a 
direction normal to the screen by a factor of 
approximately 5 db; second, the response pat¬ 
tern becomes unidirectional; third, the pres¬ 
ence of surrounding objects outside of the field 


of the lobes does not materially affect the bear¬ 
ings, and therefore the effect of reradiating 
objects, located behind the screen, can be 
tolerated to an increased degree. 



Figure 36. V-2 array, differential connection, 

relative response at 150 me. 


It becomes possible to utilize a spacing of 
greater than one wavelength between dipoles 
placed in front of an angle screen and still ob¬ 
tain only two lobes. There are no lobes on the 
back of the screen, and hence, no “null” or 
balance between lobes in a direction parallel to 


30° 20® «0° 0® 350® 340® 330® 



Figure 37. V-2 array, differential connection, 

relative response at 300 me. 


the plane of the screen. Minima exist along the 
plane of the screen, but these cannot be con¬ 
fused with a null as rotation beyond the line 
parallel with the screen does not increase the 
output. The advantage of the increased dipole 
separation up to one wavelength or more is 
that the angular sensitivity is increased. 
















































90 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


The polar patterns resulting when the V 
array, using two dipoles per screen, is con¬ 
nected as a balanced system using a differential 
connection between the two pairs are indicated 
in Figures 36 and 37. The spacing between 
dipoles in this case was 1.22a between the 
midpoint of each pair at the highest frequen¬ 
cy, 300 me. The angle between screens was not 
optimum for this use, but the polar patterns 
illustrate at three frequencies the forward gain 
and indicate the extreme sharpness of nulls. 

The forward gain of each pair of two dipoles 
in front of a screen, at an angle normal to the 
screen, is approximately 8 db over the two-to- 
one frequency band when compared to a single 
dipole in free space, a/2 long at each frequency 
of comparison. The gain measurements for the 
pair of antennas in front of a screen are given 
in Figure 22. 

The measured polarization errors are low 
and are given in Table 1. The tilted-dipole 
method was used in measuring these errors, 
and this may be roughly correlated with mea¬ 
surements made with the variable-phase polari¬ 
zation transmitter by reference to the measure¬ 
ments made on the lobe-switched V-l array 
where both methods were used. In general, it 
appeared that the tilted-dipole method was 
quite satisfactory at these frequencies, particu¬ 
larly when the errors were low. When properly 
used it is indicative of the general performance 
to be expected. 

The V-2 array mentioned above is that dis¬ 
cussed in preceding sections as the V-2 array, 
wherein it was connected as a lobe-switched 
device. The edges of the screens were insulated 
from the supporting pole. Separate balanced- 
to-unbalanced transformers were used at the 
back of each screen and grounding to the sup¬ 
port pole was made through the shield of the 
coaxial cable leading from the transformers to 
the central support shaft. 

49 DESIGN OF BALANCED-TO- 

UNBALANCED TRANSFORMERS 

The transformers used in both the V and flat 
arrays for converting the balanced dipole sys¬ 
tems to an unbalanced line are designed along 
the same lines. Each consists of two short-cir¬ 
cuited coaxial sections, A/4 long at the mean 


frequency, one connected across each half of 
the balanced line. Where reactance cancellation 
is desired, a shorted half-wave section may be 
inserted in series with the grounded side of the 
unbalanced line. The circuit is shown schemat¬ 
ically in Figure 38. 



Figure 38. Schematic of transformers for con¬ 
necting balanced doublets to unbalanced lines. 


If the dipole impedance is taken to be re¬ 
sistive, each half may be represented by R a . 
The a/ 4 lines have a characteristic impedance 
Z x and input impedance Z a , while the corre¬ 
sponding impedances for the half-wave line are 
Z 2 and Z b . The impedance looking toward the 
dipoles at Z is 


Z = 


2 Z a Ra 
Z n + R, 


(1) 


For lossless lines, Z a — jZ x tan </>, 0 being the 
phase length. 


Then 


2jZi tan 0 R a 
jZ 1 tan 0 + R a * 


( 2 ) 


2Zi 2 tan 2 0 R a . . 2Zi tan 0 R a 2 
Z* tan 2 </> + R 2 + J Z 2 tan 2 <j> + R 2 ( ’ 


2 R a . 2Zi tan 0 

, R? +J , Zi 2 tan 2 0 

+ Z i 2 tan 2 <j> + R 2 


(4) 


If Z x is made greater than R a , and 0 is in the 
vicinity of 90°: 

Zi 2 tan 2 0 » R a 2 . (5) 


Then 


- R a 2 , 

1 + V 2 ' I. — 1> aI1Cl 

Zi 2 tan 2 0 

Zi 2 tan 2 0 _ Z] 2 tan 2 0 

+ R a % ” R a 2 


( 6 ) 





































DESIGN OF BALANCED-TO-UNBALANCED TRANSFORMERS 


91 


hence z = 2R a+j ^^. (7) 

Also Z b = jZ 2 tan 20 (8) 


and for reactance cancellation— 

jZ 2 tan 20 + Im (Z) = 0 

v X n, 2 Ra 2 

Z 2 tan 20 = — 77—- - 

Zi tan 0 

or 2 R a 2 = — ZiZ 2 tan (f> tan 20 

2 tan 2 0 

ZiZ, 1 _ tan2 ^ * 

Again, for 0 in the vicinity of 90°, 

2 tan 2 0 _ _ 9 

1 — tan 2 0 

so that f^a 2 = Z t Z o. If this condition is satisfied, 
the residual series reactance introduced by the 
transformer is minimized through a frequency 
range over which the approximations made are 
valid. 

The error due to the approximation 

Z x 2 tan 2 0 » R & 2 (14) 

may be made negligible by making Z x much 
greater than R a . The practical limitation is the 
large ratio of diameters required in the coaxial 
elements for high Z x ; also, Z x increases much 
more slowly than this ratio (as the logarithm 
of the ratio). 

The other approximation: 


(9) 

( 10 ) 

( 11 ) 

( 12 ) 

(13) 


tan 2 0 
tan 2 0 — 1 


(15) 


when made over an effective phase length of 
60° to 120°, corresponding to a two-to-one fre¬ 
quency range, introduces an error of 33% 
per cent at the two extremes, and, if desired, 
may be taken into account. 

If an open-circuited quarter-wave line is sub¬ 
stituted for the half-wave line, the condition 
for cancellation becomes 


Z 2 cot 0 


2R g 2 
Zi tan 0 


(16) 


or 2R a 2 = ZiZ 2 tan 0 cot 0 

= Z X Z 2 . (17) 

This is exact, and eliminates the second ap¬ 
proximation, but requires twice the previous 


value for Z^Z. 2 . The shorted half-wave section 
also assists in keeping current from traveling 
down the outer conductor of the coaxial down 
lead and may be more desirable for direction¬ 
finding use, where stray fields must be kept to 
a minimum. 

The transformers may be seen in Figure 4 
mounted on the back of the V-l array. The me¬ 
chanical arrangement of the transformer is 
shown in Figure 39. 

BALANCED 

INPUT 



BALANCED 

INPUT 

Figure 39. Mechanical arrangement of trans¬ 
formers. 

A transformer of this type is the equivalent 
of an electrostatically screened transformer, 
and prevents a balanced system from acting as 
a grounded antenna, that is, it prevents a 
dipole from responding as a unit to a potential 
gradient between it and the ground. It enables 
the system to be balanced by merely establish¬ 
ing balance from the transformer to the dipole; 
the unbalanced line from the transformer to 
the receiving equipment does not, of course, 
require such treatment. 

The reactance cancellation line may be omit¬ 
ted under certain circumstances. If the resul¬ 
tant impedance of the antenna circuits, as seen 
at the transformer, is low, the transformer 
characteristic impedance may be made suffi¬ 
ciently high with reasonable diameters of the 
quarter-wave elements so that the residual 
series reactance introduced may be negligibly 








































92 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


small. Compensation under these circum¬ 
stances would hardly be justified, in view of 
the fact that the antenna impedance is itself 
partially reactive, and may contribute an ap¬ 
preciably larger reactive component than the 
transformer. 

*•10 DETERMINATION OF GROUND 
CONSTANTS 

To obtain the magnitude of ground reflection 
effects required for the correlation of polariza¬ 
tion error data, a number of methods of mea¬ 
surements were reviewed in the literature. The 
normal incidence method developed by Mc- 
Petrie 4 appeared to be the most likely to yield 
accurate results. Essentially it consists of set¬ 
ting up a field from an elevated source, and 
sampling the standing wave pattern set up by 
the direct and ground-reflected waves at nor¬ 
mal incidence. The ground constants may be 
deduced from the data so obtained. 

Primarily because of the special setup re¬ 
quired for this method, a comparatively simple 
laboratory method was developed, more suita¬ 
ble for the available facilities. The results ob¬ 
tained show good agreement with published 
data on the ground constants in the vicinity of 
the test site, as well as with oblique incidence 
field-intensity measurements made during po¬ 
larization error investigations. The degree of 
correlation may be observed in Figures 14 and 
15, where the measured standing wave pattern 
is shown with the pattern calculated on the 
basis of the measured ground constants. 

The method employs a short section of coaxial 
line as an extension to a slotted coaxial mea¬ 
suring line, both having the same transverse 
dimensions, and consequently the same charac¬ 
teristic impedance with air as dielectric. Pro¬ 
vision is made for either open or short circuit¬ 
ing the end of the extension. The input im¬ 
pedance of the extension is measured, when a 
sample of the ground in question is substituted 
for the air as dielectric, for the two conditions. 
The impedances so obtained may be repre¬ 
sented as follows: 


z oc — I Z oc | /floe 

(18) 

Z ac = \ Z sc \ /flsc 

(19) 


The characteristic impedance of the extension 
is then: 

7 2 — 7 7 

c ground ^oc ^sc 
= I ^OC II ^sc I /floe + flsc 

= I Z oc Z sc I [cos (floe + flsc) + 3 sin (fl oc + fl sc )] 

( 20 ) 

= r + jx (21) 

where Z oc = open-circuit impedance, 

Z sc = short-circuited impedance. 

Assume a harmonic plane wave propagated 
longitudinally along the coaxial line. The field 
components are transverse; E r is the radial 
electric field and H e the tangential magnetic 
field, as in Figure 40. The other field compo¬ 
nents are zero. 



The ratio of the electric to the magnetic field 
of a plane wave at a point is the intrinsic im¬ 
pedance of the medium for plane waves: 


A 

H, 



+ jwe 


( 22 ) 


Here /*, c, <x, are the permeability, permittivity, 
and conductivity, respectively, and to the angu¬ 
lar velocity. 

To obtain the relation between the intrinsic 
impedance of the dielectric and the characteris¬ 
tic impedance of the line, the longitudinal cur¬ 
rent and the transverse voltage are required. 







DETERMINATION OF GROUND CONSTANTS 


93 


The longitudinal current is 


IL - ds. 

(23) 

Because of circular symmetry, H e is indepen¬ 
dent of 0 ; then, along a circle of radius r, since 
Hg lies along the circle, 

II = j> H e ds. 

(24) 

Since H e is constant for a given r, 


II = Hej>ds 

(25) 

= H e (2wr). 

(26) 


If the conductivity of the inner and outer 
cylinders is much greater than that of the di¬ 
electric, the current enclosed within the path 
of integration (r 1 < r < r 2 ) may be assumed 
to flow entirely on the inner conductor. Equa¬ 
tion (26) therefore gives the longitudinal cur¬ 
rent on the inner conductor. 

The transverse voltage between the outer 
and inner conductors is defined as the line 
integral of the electric field between the con¬ 
ductors along a path lying in a transverse 
plane. A radial path is most convenient: 


Vt 



E • ds. 


(27) 


Since E and ds are both directed along a 
radius, it follows that 


Ft = J Erdr 

(28) 

from (22) E r = H»Z 

(29) 

from (26) II e = ~ 

ZttT 

(30) 

substituting (30) in (29) 


II 

toi. 

(31) 

substituting (31) in (28) 


Vt = J 2 Vr Zdl ' 

(32) 

= 4rl logr 1'; 

(33) 

hZ. , r 2 

= Iog r“ • 

(34) 


The characteristic impedance of a line is 
defined as the ratio of the tranverse voltage 
to the longitudinal current. Hence 



(35) 


This is the required relation connecting Z c 
and The characteristic impedance of a 
coaxial line is thus given by the product of a 
geometrical factor and the intrinsic impedance 
,of the dielectric medium. If two dielectrics, 
air and ground, are compared in a line of fixed 
geometry, 


Z c (ground) 
Zc (air) 


yL (ground) 
JL (air) 


(36) 


1 

1 juMff 


(Tg + ju€g 


1 ju^O 

\ 

(7 0 + j CO€ 0 


(37) 


Here the subscript g refers to ground used as 
dielectric, and 0 to air dielectric. 


or 


Z c (ground) 


'c (air) 



(38) 


If the permeability of the ground is taken to be 
equal to that of air or free space, 


Z c (ground) ‘Sj 

bl 3 

1 

o 

VD 

Zc (air) 

/ • (Tg 

\l 

^0 J 

\ 

CO 


(39) 


For free space, o- = 0, 


Zc (ground) V Co 

Zc (air) / . ag 

\ eg 3 ~ 

\ CO 


(40) 


Thus far mks units have been used. The equa¬ 
tion for converting e to electrostatic units is 


€ mks 


€ esu 

4tt X 10- 11 c 2 * 


(41) 


For converting o- to electromagnetic units the 
following equation applies: 


O’ mks — 10 11 (7 emu. 


(42) 

























94 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


Making these conversions, noting that e 0 = 1 
in esu and dropping the subscripts g and 0, 




€ esu 


/ 


me 


^c (.ground) 


The quantity under the radical is known as 
the complex dielectric constant, and may be 
represented as c' — jt". 


J _ X.n — Zc 2 (air) 
e J t 72 

c (ground) 

and from equation (21) 




Z 2 (air) 

r + jx 

tZc 1 (six) • xZ c 2 (air) 

r 2 + x 2 ^ r 2 + x 2 


(44) 


(45) 

(46) 


where the values of r and x are to be obtained 
from equation (20). 

Equating the real and imaginary parts: 


tZ 2 ( air) 

r 2 + z 2 ’ ' 


(47) 


«" = x ? - 'M r • ( 48 ) 

r 2 + x 2 

Using this method, e was found to be equal to 
10, and <r = 8.8X10- 14 emu for the ground at 
the test site. 

Care must be exercised in packing the earth 
into the line extension to maintain the same 
density in the actual and measuring conditions. 
Repeated measurements indicated practically 
constant a, while e' showed some variation 
depending on the moisture content of the earth. 
The value of 10 may be taken as representing 
average conditions. 


IMPEDANCE OF A CYLINDRICAL 
DIPOLE BEFORE A REFLECTOR 

As indicated above, considerable variations 
were encountered between the measured impe¬ 
dance characteristics of the dipoles used on the 
V-l screen and the theoretical characteristics 
based on prolate spheroidal dipoles as given by 
Stratton and Chu. 5 Certain other treatments 
of the problem were examined in an attempt 
to obtain better agreement between experimen¬ 
tal data and existing theory. 

The values of self impedance obtained from 
Hallen’s formulas as given by King and Blake, 6 


and King and Harrison, 7 and the values we 
calculated from the formulas of Schelkunoff, 8,9 
were compared to the experimental values of 
impedance obtained on the V-l array. The 
latter is the impedance in the presence of the 
reflector; corrections for the mutual impe¬ 
dance between the dipole and its image were 
to be applied on the basis of the theory de¬ 
veloped by Brown. 10 

The values of self resistance based on 
Hallen’s formula were found to be too high; 
results obtained from Schelkunoff’s formula 
showed better agreement, but not good enough 
for engineering purposes; a correction 11 for the 
concentrated capacitance in the vicinity of the 
gap brought this theory into much closer agree¬ 
ment with the measurements. It may be men¬ 
tioned that a modification of Hallen’s solution 
by Gray, 12 yields better results than the origi¬ 
nal, but not as good as Schelkunoff’s. A com¬ 
parison of the latter with our measurements 
is made below. 

The self impedance of a cylindrical dipole is 
given by: 

Z S e\f 

Zofi,smff/+ji[A',-/ 2 (2$)]sin$—[Z„—/i(2$)]cogffl| 

[Z 0 +/i(2|82)]sin/3/.+!X ( +/ 2 (2^)]cosj32— jR, cos02 

(49) 

where 

Z 8 eif = self impedance 
Rt = terminal resistance 

= 60 Cin 2/3/+30((7+ln /3Z—2 Ci 2/3/+Ci 4/3/) 
cos 2/3/ + 30 (Si 4/3/—2 Si 2/3/) sin 2/3/ 
Xt = terminal reactance 

= 60 Si 2/3Z + 30 (Ci 4/3/—In /3 l—C) sin 2/3/ 
—30 Si 4/8/ cos 2/3/ 
/i(2/3Z) = 60(Cin 2(31 — 2 sin 2 (3l) 
f 2 ( 2 (31) = 60(Si 2/3/ — sin 2/3/) 

Z 0 = 120 In - — 120 
a 



X = wavelength 
/ = half length of dipole 
a = radius of dipole 

Ci( ) = cosine integral function, tabulated 13,14 
Si( ) = sine integral function, tabulated 13,14 
Cin( ) = C + In ( ) - Ci ( ) 

C = Euler’s constant ( = 0.5772) 














CYLINDRICAL DIPOLE IMPEDANCE BEFORE A REFLECTOR 


95 


The resistive component of the self impe¬ 
dance is: 

ftseif = §•“ R t [Z„ + /,(2/3?) sin 2/3?—/i(2/3?) cos 2^1 

(50) 

The reactive component of the self impe¬ 
dance is: 

*sei, = | °{hr; ! + x t 2 - z? + 

+ /**(2/3?)] sin 2/3? + [//(2/3?) / 2 (2/3?) 

cos 2 pi + [X</i(2/3Z) 

— Z o / 2 (20?)]} (51) 

where 

D = ( R t cos pi) 2 + {[Z 0 + /i(2jS/)] sin 0/ 

+ [X, + M20l)} cos /3?} 2 (52) 

Figure 41, curve a, is a plot of the resistive 
component of the free space input impedance 
of the dipole used in the V-l array, as obtained 



Figure 41. Resistance characteristics of dipole in 
front of reflector. 


from equation (50). The resistance corrected 
for a gap capacitance of 2.0 ^f is shown in 
curve b, while values of the measured resis¬ 


tance are given in curves c, d, and e for spac- 
ings from the reflector of 33, 28.5, and 24.2 cm 
respectively. Curve b is in good qualitative 
agreement with the measurements. The out¬ 
standing differences are the downward dis¬ 
placement along the frequency scale of the 
experimental curves, and the relatively high 
maximum value of the theoretical curve. Better 
agreement is possible if a decrease in velocity 
of propagation greater than predicted by the 
theory is assumed. 

The corrections for mutual impedance were 
not applied to the theoretical curves, as the dis¬ 
crepancies between the latter and the experi¬ 
mental curves are of the same order of magni¬ 
tude as the corrections involved. To test the 
applicability of the mutual impedance theory 
to dipoles of the proportions used, the reverse 
process was adopted. Starting with the three 
measured resistance curves of Figure 41, the 
three corresponding self-resistance curves were 
deduced by means of the inverse corrections 
for mutual resistance. The three self-resis¬ 
tance curves so obtained are almost identical 
up to a full-wave dipole length, indicating that 
the theory is applicable up to this limit. The 
group of three self-resistance curves is iden¬ 
tified as / on Figure 41. 

The mutual resistance is accounted for in the 
following manner: The resistive component of 
the coefficient of radiation coupling is known 
to be independent of dipole length for two 
identical parallel nonstaggered thin dipoles, 
up to one wavelength long. It may be defined 
as the ratio of the resistive component of 
mutual impedance to the resistive component 
of self impedance. Thus, if the coefficient is 
known, either the self or mutual resistance 
may be obtained provided one or the other is 
known. 

The mutual resistance between two dipoles 
as limited above is : 15 

^mutual = sil ^ / {2(2 + cos2/3Z)Ci2/3s—4 cos 2 0/(Ci pE 

+ Ci (3F) + cos 2 pi (Ci pG + Ci pH) 

+sin 2 pi (Si pH— Si pG— 2 Si pF 4- 2 Si pE)} 

(53) 





96 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


The notation here is the same as above, with 
the additions 

s = half the distance between the two 
dipoles (or the distance from one 
dipole to a reflector) 

E = (V 4s 2 + l 2 - 1) 

F = (V 4s 2 + l 2 + 1) 

G = (2 V s 2 + l 2 “ 1) 

H = (2 \/ s 2 + Z 2 + 1) 

Since equation (53) is the asymptotic ex¬ 
pression for the mutual resistance of two in¬ 
finitely thin dipoles, it may not be compared 
with equation (50) directly to obtain the resis¬ 
tive component of the coefficient of coupling. 
The following expression may be used: 

ft self = 30{(l-cot 2 pi) (Cin 4$)+4 cot 2 pi (Cin2 pi) 
+2 cot pi (Si 4/M—2 Si 2 pi)} . (54) 

The ratio of equation (53) to equation (54) 
is the resistive component of the coefficient of 
radiation coupling. A plot of R mutual /RseU IS 
given in Figure 42 as a function of 2s/A. 



Figure 42. Resistive component of coefficient of 
radiation coupling. 


The deduced values of self resistance given 
by curve / of Figure 41 were obtained using 
Figure 42, since the input resistance may be 
expressed as: 

(55) 


Calculation of the self reactance is based on 
equation (51). The mutual reactance is ac¬ 
counted for as follows: the phase angle of the 
mutual impedance between two identical paral¬ 
lel nonstaggered thin dipoles, up to one wave¬ 
length long, is to a first approximation inde¬ 
pendent of the length, and a linear function of 
the spacing. The linear connection is, for s 
greater than O.lA, 

(f> = — 312 — -j- 42 (56) 

A 

Here <£ is the phase angle in degrees; the other 
symbols are as previously used. 

From the relation 


tan 4 > = 


X 


mutual 


-^mutual 


(57) 


and the previously obtained values of R mutual? 
-^mutual may be determined. The total input re¬ 
actance is then 

*in = *self - ^mutual • (58) 

Since the expression for X muUia i contains tan 
</> as a factor, the correction for X mutU ai as ex¬ 
pressed in equation (58) may have much larger 
values than the corresponding correction 
R m utuai in equation (55). The former correc¬ 
tion was therefore applied directly to the values 
of X seU as obtained from equation (51), for 



Figure 43. Reactance characteristics of dipole in 
front of reflector. 


the dipoles used on the V-l array, for three 
values of the spacing s from the reflector. Fig¬ 
ure 43 is a plot of these: curve A is for a spac- 


Rin -^self -^mutual • 














CATHODE-RAY INDICATION AND AUTOMATIC CONTROL 


97 


ing of 38 cm, or a/ 4 at 227 me; curve B , 28.5 
cm, 263 me; curve C , 24.2 cm, 310 me. The 
corresponding experimental curves are shown 
at a, b, and c of the same figure. An examina¬ 
tion of these curves indicates that the correc¬ 
tions for mutual reactance are of the correct 
order of magnitude. As in the case of the 
resistance curves of Figure 41, the experi¬ 
mental reactance curves correspond to a veloci¬ 
ty of propagation lower than that predicted by 
the theory, and the values of computed reac¬ 
tance are high. 

The accuracy of wavelength determinations 
made in the course of impedance measure¬ 
ments is sufficiently high to preclude the pos¬ 
sibility of experimental error accounting for 
the difference in velocity of propagation in¬ 
dicated by these two sets of curves. 

4--la CATHODE-RAY INDICATION AND 

AUTOMATIC CONTROL 

Subsequent to the expiration of the contract, 
several methods of cathode-ray indication equiv¬ 
alent to plan position indicators [PPI] were 
developed and an automatic control was added 
to the flat array to indicate the practicability 
of the arrays when used for direction finding. 

In searching it is desirable to rotate the an¬ 
tenna array and provide a visual means of 
locating the azimuth. To accomplish the rota¬ 
tion and also to provide means of automatically 
obtaining a bearing once the signal quadrant 
is known, an amplidyne servo system was in¬ 
stalled. This was used to drive the antenna 
shaft either (1) through means of a manually 
operated selsyn control, or (2) automatically 
through suitable output amplifiers connected to 
the differential voltage developed across the in¬ 
dicator meter circuit. These two arrangements 
provided means for rotating the array to any 
desired azimuth when the selsyn was used, or 
to automatically orient the array to the signal 
bearing when the receiver output differential 
voltage was used as the control. The maximum 
speed of antenna rotation from either arrange¬ 
ment was 6 rpm. 

In addition to the L-R indicator meter, 
which indicates when the array is on bearing, 
a long persistent CR tube was used in the com¬ 
binations which follow. The means of placing 


the CR spot or trace, depending upon the 
presentation employed, was to gear a resistor 
control to the antenna shaft and provide elec¬ 
trical connections from this to the deflecting 
plates of the CR tube. The resistor consists of 
a circular strip with two brushes at 90° from 
each other. If direct current is applied to the 
proper terminals of the resistor strip, the CR 
spot is moved from the center of the tube to an 
angular position corresponding to the location 
of the resistor brushes. 

• Under the above condition, rotation of the 
brushes produces a circular trace. The resistor 
control being geared to the antenna shaft, 
therefore, produces a trace which is synchro¬ 
nized with the antenna array. This is shown 
schematically in Figure 44. Several forms of 


-ANTENNA SHAFT 



Figure 44. Schematic representation of circular- 
trace generator. 

presentation were tested, which, in each case, 
indicated the array position and the relative 
amplitude of the signal. 

Indication Presentation 

The first method employed was to superim¬ 
pose on the circular trace the differential volt¬ 
age developed across the L-R indicator meter. 
The pattern, Figure 45A, is such that signals 
to the left of the bearing appear as an increase 
in the circle and are, therefore, outward, while 
at the right of the bearing the patterns are in¬ 
ward. At the bearing position, the trace is 
evenly divided in amplitude about the circle. 
This arrangement is unmistakable but also un- 
symmetrical and, therefore, requires a slight 
amount of interpretation. 

















98 


ULTRA-HIGH-FREQUENCY DIRECTION-FINDING STUDY 


A second method is to connect the d-c voltage 
across the rotatable resistor and in series with 
the output rectifier from the receiver without 
going through the L-R meter switch. In this 


o° 



. A 


line at the bearing is obtained as shown in 
Figure 45B. 

Another arrangement is to drive the circular 
trace inward rather than outward. This forms 


o° 



180° 

B 


Figure 45. A shows pattern secured by superimposing on circular trace differential voltage developed across 
the L-R indicator meter. B shows d-c voltage connected across rotatable resistor and in series with rectifier 
output from receiver without going through L-R meter switch. 


case the circular trace is maintained and a pat¬ 
tern which increases the circular trace on or to 
either side of the null and drops to a balanced 


o° 



180 ° 

A 


a more suitable pattern, since the bearing is 
indicated by an arrow formed by the parts of 
the face of the tube which were not illuminated 

o° 



B 


Figure 46. A shows circular trace driven inward rather than outward. B shows lobe-switched output con¬ 
nected to produce trace of two lobes. 



CATHODE-RAY INDICATION AND AUTOMATIC CONTROL 


99 


by the trace. Figure 46A shows this pattern. 

A fourth arrangement is to connect the lobe- 
switched output in such a manner as to produce 
the trace of both antenna lobes. In this case 
the intersection of the lobes indicates the bear¬ 
ing as illustrated in Figure 46B. 

Other arrangements were employed connect¬ 
ing the antenna array as a differential array 
forming, in effect, an Adcock antenna and 
tracing the pattern and null directly on the 
tube. (See Figure 47A.) A reversed connection 
of this arrangement, shown in Figure 47B, 


It appears that the maximum utility of the 
CR tube indicator is to locate roughly the 
source of the signal with an accuracy of ±2°. 
A bearing may be read more accurately if ob¬ 
tained by the automatic control once the 
quadrant has been located. The bearing scale 
for the automatic control was read directly 
from the azimuth scale mounted on the antenna 
array, although provisions are made in the 
amplidyne system to read the indication from 
a separate selsyn which is geared to the anten¬ 
na shaft. 


o° 



t 


180 ° 

A B 

Figure 47. A shows effect of antenna connected as differential array forming Adcock antenna. B shows effect 
of reversing connections from those producing A. 



produces a trace which draws a line outward 
to the edge of the CR tube at the bearing indi¬ 
cation point. The antenna arrangements for 
the two latter patterns do not require the lobe¬ 
switching mechanism and are, therefore, some¬ 
what simpler. However, this arrangement can¬ 
not readily be employed as an automatic direc¬ 
tion finder or be electrically connected to the 
servo system so that the bearing is obtained 
automatically. 


Many other presentation arrangements are 
possible using the CR tube. The methods of in¬ 
dication presentation suggested above, with the 
exception of the system which presents the 
direct or reversed patterns of the Adcock ar¬ 
rangement, are based on the lobe switching of 
the antennas. It should be noted that all of the 
above illustrations show the bearing at 0° azi¬ 
muth. The patterns in each case rotate with 
bearing. 





Chapter 5 


ERRORS IN DIRECTION FINDERS 


N umerous projects under Division 13 were 
concerned with the fact that direction 
finders of various types do not give consistent 
or accurate bearings in spite of the fact that 
they can be erected with great care and made 
up of precision apparatus. Some of these 
errors were found to be due to the fact that 
elevated structures do not have all parts equi¬ 
distant from the reflecting or semi-conducting 
ground; that waves arriving from the iono¬ 
sphere are polarized in heterogeneous ways; 
that waves traversing regions near the mag¬ 
netic poles do not always follow the great- 
circle route; and there are still other reasons 
why d-f results do not have the accuracy de¬ 
sired. The background for these troubles will 
be found discussed in Chapters 1 and 2, and, in 
fact, throughout the summaries of d-f projects 
reported in this volume. 

si PROJECT C-17 a 

This was the first of a continuing series of 
projects for study of certain errors of shielded- 
U Adcock direction finders. 

Under Project C-17 1 will be found a general 
review of the directive properties of radio 
waves and wave collectors, giving reasons why 
simple loop and dipole antenna d-f systems are 
not accurate under normal conditions of h-f 
wave propagation. The fact is that spaced-an- 
tenna systems to eliminate the faults of the 
simple loop or dipole are in theory highly ac¬ 
curate but in practice are not so. The impor¬ 
tance of taking rapid bearings, of making all 
antennas of a given spaced-antenna d-f system 
identical, and of limiting unwanted pickup 
from extraneous conductors is discussed in this 
review, which also evaluates various known 
wave-collecting systems. 

This review found the shielded-U Adcock 
especially promising. Since the nature and 


a Contract No. NDCrc-149, Radio Corporation of 
America. 


extent of the shield required to produce suf¬ 
ficiently accurate bearings on sky waves had 
never been fully studied, Project C-17 was set 
up to study the design and properties of this 
particular type of antenna system. Attention 
was directed particularly toward portable 
equipment. 

A precise, demountable, shielded-U Adcock 
antenna was built on top of a station wagon, 
for portability, and a receiver with calibrated 
attenuator was installed in the station wagon 
to measure antenna responses under various 
conditions of wave incidence. Conductors were 
provided for building up elevated artificial 
ground planes of varying extent and complete¬ 
ness. A local source of test signals of definite 
polarization was provided, together with a bal¬ 
loon and rigging to elevate this source for pro¬ 
duction of sky-wave signals. A transit was 
used for observing polarization and direction 
of arrival of the test signals. 

Measurements were made with this system 
over the range 7 to 18 me with antennas con¬ 
nected to the input transformer of the receiver 
directly or through cathode followers, the two 
methods giving about the same errors. The 
quantities measured were mostly maximum/ 
minimum ratios for directive patterns and 
minima positions for ground waves, and the 
ratio of maximum responses to vertically and 
horizontally polarized ground and sky waves. 

Trouble was experienced from the beginning 
with the inadequacy of the artificial ground 
systems tried as a part of the shielding of the 
Adcock U—trouble which has been observed in 
all other d-f projects summarized in this vol¬ 
ume. Radial-wire counterpoises were found to 
be wholly inadequate, radial plus ring-wire 
counterpoises gave good results on ground 
waves but had excessive errors on unfavorably 
polarized sky waves. Netting with radial wire 
extensions, shown undergoing tests in Figure 1, 
worked fairly well with ground waves and 
showed only moderately excessive errors with 


100 



PROJECT C-38 


101 


sky waves. The netting, however, was not con¬ 
veniently portable. 

The general conclusion was that a carefully 
made portable shielded-U Adcock using a de¬ 
mountable elevated counterpoise can be highly 
accurate for horizontally arriving signals, but 
cannot be made outstandingly free of polariza- 


ter of a 65-ft diameter spider web arrange¬ 
ment with 24 radial and 8 ring wires, with the 
8-ft copper disk in the center. 

Considerable work was carried out with 
balloons, with attendant difficulties which 
limited the amount of downcoming-wave data 
obtained. 



Figure 1. Artificial ground system composed of netting with radial wire extensions, shown undergoing tests. 


tion errors on downcoming signals without un¬ 
due sacrifice of portability. Standard-wave 
errors of the order of 10° at best were attained. 

511 Apparatus Employed 

A continuous copper disk 8 ft in diameter 
was mounted on top of the station wagon and 
determined the size of the antenna system. 
Thus the antenna spacing was arbitrarily set 
at two-thirds of the disk diameter or about 51/2 
ft. This spacing was A/6 at 30 me and was 
A/36 at 5 me. This small spacing made the 
system rather insensitive, a 1° change in azi¬ 
muth of an arriving signal producing a phase 
change of antenna voltage of only 10 minutes 
of arc at 5 me. The height of the antennas was 
121/2 ft, which was %A at 30 me. 

Many types of counterpoise systems were in¬ 
vestigated and the one with the greatest density 
of conductors was best, but a larger one with 
fewer conductors was more practical and fair¬ 
ly good. This was made up of 48 radial wires 
each 100 ft long attached to the outer perime- 


In constructing the test oscillator to be used 
in the work with the shielded-U Adcock, care 
was taken to see that the purity of polarization 
was high. This was secured by making the 
test oscillator long and narrow to minimize the 
possibility of r-f current flow in any direction 
other than that of the antenna rods attached to 
its ends. Electrical symmetry was provided by 
connecting the case of the miniature battery- 
powered transmitter to the center of the coil 
feeding the two rods of the symmetrical dipole 
antenna. 

5.2 PROJECT C-38 2 

In earlier work, tests had been made of a 
counterpoise made up of radial wires and ring 
wires connected at the points where rings 
crossed radials. Further tests were made under 
C-38 b with a counterpoise of ring wires only. 
Results were, as expected, decidedly worse than 
with counterpoise arrangements tried earlier. 

b Contract No. OEMsr-338, Radio Corporation of 
America. 






102 


ERRORS IN DIRECTION FINDERS 


The purity of polarization of the test trans¬ 
mitter developed under Project C-17 was ex¬ 
amined and it was found that the ratio of 
vertical receiving antenna output with trans¬ 
mitter antenna horizontal and then vertical 
was over 500. 

Some unsuccessful trials were made of a 
large kite to supplement the balloon as support 
for a source of high-angle downcoming waves. 
The balloon rigging was revised to give im¬ 
proved operation over a wider range of 
conditions. 

5,2,1 Tests at Holmdel 

Arrangements were made to take the balloon 
rigging, test transmitter and other auxiliary 
apparatus to Holmdel, New Jersey, where the 
Bell Laboratories were developing under 
Project C-16 (summarized in Chapter 1 of this 
volume) a shielded-U Adcock for fixed-station 
service. Here polarization error measurements 
were made with steeply downcoming waves. 
Some description of the Holmdel equipment will 
be found in Chapter 1. The test transmitter 
was hoisted to the top of a 50-ft tower at 
Holmdel and hung approximately in line with 
the east-west Adcock pair described in the C-16 
summary. Measurements were taken at six 
frequencies from 3.46 to 17.30 me, with the 
test transmitter hung from the tower at 1.5° 
intervals from an elevation of 1.5° to 13.75° 
and when suspended by the balloon to eleva¬ 
tions corresponding to 50° or 60°. 

At each frequency and elevation, output of 
both Adcock antenna pairs was recorded both 
with the transmitter dipole vertical and with 
it horizontal. Unexplained minima of un¬ 
wanted pickup for a transmitter elevation of 
about 5° were observed at all frequencies and 
were very pronounced at the higher ones; no 
corresponding horizontal field minima were 
observable. 

Vertical to horizontal field-strength ratios at 
the center of the Adcock system, both for the 
Project C-17 tests and those at Holmdel, were 
computed using a number of terms of the 
series-expansion solution of Maxwell's equa¬ 
tions given by Burrows. 4 The results indicated 
a tremendous enhancement of vertical field 


under the short-range transmission conditions 
used in the tests. Therefore, standard-wave 
errors for distant signals as determined from 
the above computed test-signal fields were 
much greater than such errors as commonly 
determined directly from measured ratios of 
wanted output for vertically polarized signal 
to unwanted output for horizontally polarized 
signal. 

The directly measured results indicated that 
the Holmdel (C-16) Adcock was markedly less 
subject to polarization errors than the elevated- 
counterpoise Adcock of Project C-17, and was 
of the general quality (2° to 10° apparent 
standard-wave error in the range 17.5 to 
3.5 me) which other recent work had shown to 
be typical of good direction finders. Similar 
results for the C-17 Adcock with the better 
counterpoises ran from 7° to 15° in the fre¬ 
quency range 7.5 to 17.5 me. 

Extreme enhancement of local vertical fields 
is a matter of such tremendous importance to 
d-f testing, since it would completely invalidate 
almost all previous work, that it was studied 
further as reported under Project C-57. Ap¬ 
proximate but seemingly sound application of 
general field theory to results of the Holmdel 
tests indicated improbably large standard-wave 
errors on distant signals. 

52,2 Conclusions 

The final report 2 on Project C-38 includes 
further general discussion of d-f design princi¬ 
ples and testing methods, which leads to a 
number of conclusions. 

Optimistic beliefs resulting from earlier 
work on the freedom of Adcock systems from 
polarization errors were not borne out by this 
or other recent work. In agreement with re¬ 
cent results of others on H Adcocks, it was 
' concluded that shielded-U Adcocks are subject 
to a first-order error source of nature still un¬ 
known. In particular, the elevated counterpoise 
shielded-U system of Project C-17 did not com¬ 
pare as unfavorably with other systems as 
was at first supposed, so conclusions from its 
study are given in the form of concrete pro¬ 
posals for counterpoise design. 

Since no direction finder can work well with 
all types of waves received, a “directive di- 


cunudkn i i 





PROJECT 057 


103 


versity” system was proposed in which some 
one of a group of two or three spaced-antenna 
direction finders, at the same location but each 
using a different type of antenna, will respond 
accurately to any coherent signal received. Use 
of devices to warn against vertically down¬ 
coming signals was suggested. 

Knowledge of the means whereby polariza¬ 
tion errors arise was not sufficient at the time 
the work was done to permit either sound de¬ 
sign of direction finders or safe extrapolation 
from errors measured under usual test condi¬ 
tions to determine performance under widely 
different operating conditions. Inevitable pres¬ 
ence of the ground improves performance of 
wave collectors at certain heights and injures 
their performance at other heights; whether 
good or bad, the effect is stronger the more 
conducting the ground. In general it was con¬ 
cluded that improvement of the direction finder 
itself was more to be desired than an equal 
improvement by choosing a site on better 
ground. 

53 PROJECT C-57 3 

The great importance of having a truly re¬ 
liable method of determining polarization er¬ 
rors, because of their probably larger magni¬ 
tude in practical equipment, made it desirable 
to continue the work undertaken in the previous 
projects and to examine the experimental 
methods and the theoretical calculations of 
wave-field components used in testing under 
those projects. 

The startling nature of the theoretical results 
obtained under Project C-38, c which appeared 
to invalidate practically all prior d-f measure¬ 
ments, indicated the desirability of a more 
thorough study. Thorough examination of the 
exact series-expansion solution of Maxwell’s 
equations given by Burrows, 4 from which the 
approximations used in Project C-38 were ob¬ 
tained, showed it to be unsuitable for computa¬ 
tion under just the conditions for which ex¬ 
treme enhancement of local vertical fields had 
been computed and reported under that project. 5 

A new approximate solution of Maxwell’s 

c Contract No. OEMsr-838, Radio Corporation of 
America. 


equations, suitable for computing under the 
conditions of direction-finder testing, was de¬ 
veloped from the exact solution in integral 
form given by van der Pol. 6 Comparisons with 
unpublished work of K. A. Norton showed this 
solution to be fundamentally the same as the 
one recently reported by him. Both solutions 
are valid under the short-range, high-angle 
conditions of d-f testing and both assume high 
ground conductivity. The new solution, in the 
relatively simple form given it by Norton, was 
used to re-evaluate the Holmdel results of 
Project C-38 and to analyze new experimental 
results obtained under Project C-57. In each 
case, the vertical electric field component pro¬ 
duced near a horizontal rod antenna by curva¬ 
ture of the wave fronts was computed, as well 
as the horizontal electric field of the horizontal 
rod antenna and the vertical electric field of a 
vertical rod antenna. 

Application of these results to the Holmdel 
data of Project C-38 showed clearly that no re¬ 
liable measurement of polarization error of the 
Holmdel Adcock had been obtained. Spurious 
vertically polarized signal due to wave-front 
curvature near the horizontal test source had 
obscured the unwanted horizontal field pickup 
of the Adcock. This field curvature was evi¬ 
dently also the cause of the apparent minimum 
of error found at Holmdel for waves arriving 
at 5° elevation. A few of the balloon observa¬ 
tions appeared to exhibit real polarization 
errors and permitted a rough estimate of 
standard-wave error as varying from 9 to 61/2° 
between 5 and 9 me. Pickup ratio, where de¬ 
termined, is apparently very low; good operat¬ 
ing accuracy results from placing the system 
right on the surface of good ground. Some 
data taken by Bell Telephone Laboratories at 
Holmdel with both rod- and loop-antenna 
sources of test signal showed the same curva¬ 
ture effects. Up to the time this work was con¬ 
cluded, no measurements made on the Holmdel 
Adcock had been good enough to give an ac¬ 
curate determination of its polarization errors. 

Because a horizontal loop transmitter does 
not produce spurious vertical electric fields due 
to wave-front curvature, further tests were 
made on the C-17 elevated-counterpoise Adcock 
to compare such a source with the horizontal 




104 


ERRORS IN DIRECTION FINDERS 


rod or electric dipole radiator previously used. 
The rod-shaped test oscillator built for Project 
C-17 was modified to work with either rod or 
loop antenna and comparable tests were made 
with both source types. 

No difference was found between measure¬ 
ments made on the elevated-counterpoise Ad¬ 
cock with rod and loop transmitters. The field 
computations indicated that real polarization 
errors were measured and were so great as to 
obscure the considerable field curvature effects. 
Pickup ratios were very low, especially for 
signals arriving at high elevation angles. They 
were of the same order as those estimated for 
the essentially similar Holmdel system, but the 
aid given to overall accuracy by good ground 
at Holmdel was lacking for the elevated- 
counterpoise system as measured at Princeton. 
Standard-wave error at 7 me and over rather 
poor ground was found to be 32°. 

Special tests of the loop transmitter showed 
that field-curvature effects were not eliminated 
but were reduced at least five-fold by its use. 
The rod transmitter is inadequate for measur¬ 
ing standard wave errors below 20°, except for 
high elevation angles, while the special loop 
transmitter used in this project should measure 
reliably errors as small as 4°. 

Introduction of damping resistors into the 
elevated-counterpoise structure failed to re¬ 
duce errors but did show how size relations 
between conductors acted to equalize errors 
over a wide frequency band. No evidence could 
be found that voltages induced in the counter¬ 
poise by horizontally polarized signals were fed 
into the antennas by capacitance, but it 
was noted that a single vertical antenna 
mounted eccentrically above the counterpoise 
was markedly more responsive to horizontally 
polarized signals than was a centrally located 
vertical antenna. 

Ordinary testing equipment and methods are 
clearly inadequate for the study of polarization 
errors of really good direction finders. At the 
close of this work, it appeared that no fully 
adequate test had yet been made of the down¬ 
coming wave performance of any very good 
direction finder. Vertical field enhancement 
near a local transmitter is not as extreme as 
the result of Project C-38 had indicated and 


is unimportant at high angles. It is quite im¬ 
portant at the low elevations and short ranges 
used in much d-f testing. 

54 PROJECT C-78 d 

Project C-78 8 was concerned with the 
measurement of errors of radio direction 
finders and served to correlate and evaluate 
knowledge of measuring techniques gained in 
the work of Projects C-17, C-38, and C-57. The 
whole problem of such measurements was sur¬ 
veyed including the question of what to 
measure, how to measure it, of the range of 
measurement necessary or desirable, and of 
the characteristics required in the measuring 
equipment. Because of the great importance 
of the technical capabilities of radio direction 
finders to their user, methods of performance 
testing require careful specification. Overall 
tests designed to simulate actual operating con¬ 
ditions and to yield direct information as to 
accuracy of bearings are highly desirable. 

Noise level and accuracy of equipment 
auxiliary to the d-f antenna system, like ac¬ 
curacy of reading at various steady signal 
levels, can and should be separately determined 
by normal laboratory methods. Conditions for 
determining reading errors on actual fluctuat¬ 
ing signals cannot readily be specified. Errors 
due to good signals arriving by laterally dis¬ 
torted paths are not errors of the direction 
finder itself, while signals arriving from eleva¬ 
tions above about 60° should not be used for 
direction finding. Testing methods must avoid 
conditions which cannot be specified clearly or 
should properly be excluded from measurement. 

Errors due to interference among signal 
components arriving over multiple paths are of 
great importance, as are errors caused by elec¬ 
trical inhomogeneities in the immediate sur¬ 
roundings of a direction finder. Conditions 
for measurement of these errors could not yet 
be specified at the conclusion of this project. 
Test methods should avoid producing such 
errors, yet be adaptable to their controlled 
production when the art permits specification 
of appropriate tests conditions. 

Failure of actual d-f wave collectors to re- 

d Contract No. OEMsr-838, Radio Corporation of 
America. 




PROJECT C-78 


105 


semble exactly their ideal prototypes, even 
when well located and receiving a steady single¬ 
component signal, was an important source of 
error at the time this work was done. Condi¬ 
tions for measurement of such errors could 
already be specified and they could and should 
have been measured reliably, but this had 
practically never been done. These errors are 
of two types often called “calibration” and 
“polarization” errors (night effects), and it 
was to their measurement that Project C-78 
was directed. 

Actual distant signals have very complex 
properties which are usually incompletely 
known and are therefore poorly suited for per¬ 
formance testing, though limited data can be 
obtained statistically from large numbers of 
distant-signal observations. Reliable and com¬ 
plete testing requires a fully controllable local 
source of test signals, arranged to simulate the 
properties of certain typical distant signals. A 
few actual distant-signal observations are de¬ 
sirable to check the validity of the local-source 
test method. 

Measurements made with local sources under 
simplified limiting conditions may not always 
be reliable guides to performance under all 
operating conditions. Even when special 
detailed knowledge of a particular direction 
finder permits general conclusions to be drawn 
from simplified measurements, the rigorous 
theoretical work required may be less con¬ 
venient than more complete direct measure¬ 
ments. The pronounced effect of the inevitable 
presence of the earth near every direction 
finder cannot always be treated as separate 
from its intrinsic performance; in some cases 
only overall performance of ground and direc¬ 
tion finder together is significant. 

Carefully interpreted measurements from a 
lower limiting frequency between 6 and 10 me 
to an upper limit between 18 and 30 me can 
indicate performance of similar direction 
finders over the entire h-f band from 1.5 to 
30 me. Model measurements at very high fre¬ 
quencies avoid some difficulties of full-scale 
testing but require development of suitable 
models and may be misleading because the 
model does not accurately simulate obscure 
imperfections important in the original. 

Variation of error with signal-arrival azi¬ 


muth on favorably polarized signals and varia¬ 
tion of error with signal-arrival elevation on 
unfavorably polarized signals must both be 
determined at several frequencies. The results 
can only be fully shown as graphs, but effort 
should continue to express a maximum of in¬ 
formation by a few figures of merit. Such mea¬ 
surements should be carried out over very uni¬ 
form highly conducting ground to determine 
ultimate performance capability and over uni¬ 
form poorly conducting ground as well to 
determine possible impairment of performance. 

The primary instrument used in d-f testing 
is a signal field and test methods must be 
planned on the basis of accurate knowledge of 
its characteristics. Approximate expressions 
defining this field, as developed by Norton, 
show its properties to be quite complex, es¬ 
pecially near the signal source. This complexity 
is caused mainly by the presence of the earth’s 
surface. 

Signals from a local source differ from those 
from a distant source in two ways. The local 
signal shows different rates of attenuation with 
distance for components plane-polarized re¬ 
spectively parallel and perpendicular to the 
vertical plane of wave travel, while the distant 
signal shows no such difference. The local 
signal, spreading from a small source, has 
curved wave fronts which cause somewhat 
different fields to appear at laterally separated 
parts of the direction finder, while the distant 
signal has plane-wave fronts. Each of these 
differences can seriously confuse d-f error 
measurements. Both can be avoided only if all 
measurements are made at transmitter-receiver 
distances of at least several tens of wavelengths. 

The main method of measurement used in 
recent work involves two observations of the 
output of the d-f wave-collector system under 
test, one in a field of the polarization to which 
the collector elements were designed to respond 
and the other in a field to which no response was 
intended. The ratio of these outputs, for equal¬ 
ly strong incoming signals, gives the maximum 
polarization error to be expected. Separate 
observation with signals of limiting polariza¬ 
tion avoids phasing difficulties of earlier work 
where both signals were present at once, but 
requires test signal sources of extreme polari¬ 
zation purity. 



106 


ERRORS IN DIRECTION FINDERS 


Projects C-17, C-38, C-57, and C-78 used a 
signal source designed to achieve pure polari¬ 
zation by being entirely self-contained and con¬ 
forming in outline to the intended antenna, a 
short rod (electric dipole) or small loop (mag¬ 
netic dipole). The frequency was stabilized by 
crystal control of a master oscillator driving a 
balanced power amplifier coupled to the highly 
reactive antenna by an autotransformer. This 
source was light in weight for convenience in 
elevated operation and its power output was 
maximized by use of efficient miniature power 
tubes and miniature batteries under very heavy 
load. Figure 2 shows a close-up of the interior 


formance. Unwanted emission, while not defi¬ 
nitely determinable by this method, seemed to 
be at least one per cent of wanted emission in 
field strength. This purity is adequate for tests 
in which a direction-finder null may be used to 
discriminate against unwanted emission but 
quite inadequate for more exacting tests. An 
appreciable electric-dipole moment was exhibi¬ 
ted by the magnetic-dipole source, probably be¬ 
cause of the breaks in the loop required to con¬ 
nect the generator. Elimination of unwanted 
electric field components due to wave-front 
curvature, attractive in principle, was thus 
found very difficult in practice. 



Figure 2. Internal arrangement of test source employed in Projects C-17, C-38, C-57, and C-78. 


of the 4-in. by 2-ft cylindrical test source, with 
inserts showing its incorporation into rod and 
loop radiators. 

The usual method of determining purity of 
transmitter polarization, by observing the out¬ 
put of a receiving antenna of supposedly pure 
polarization with the transmitter in various 
orientations, was shown by a complete analysis 
to be generally incapable of giving the desired 
result. Tests of this type, using an accurately 
vertical rod receiving antenna centered over an 
accurately horizontal circular elevated counter¬ 
poise, were made on signals from the above 
source and indicated rather disappointing per- 


The output-ratio method of d-f testing is in¬ 
direct and slow, beside requiring inconvenient 
manipulation and sometimes needing an un- 
attainably pure source. Some other means of 
avoiding error reduction by chance favorable 
phasing of various field components would 
avoid these difficulties. An improved method of 
d-f error measurement based on a novel test 
signal source was proposed in the final report 
on Project C-78. 8 The proposed signal source 
would use an antenna unit consisting of two 
distinct radiators of different polarization, 
preferably a vertical rod and horizontal loop. 
These would both be fed from a common r-f 







PROJECT C-58 


107 


generator, with constant relative amplitude 
and continuously varying relative phase. Po¬ 
larization error would be observed as a swing¬ 
ing bearing and measured by amplitude of 
swing. Error measurement would thus be 
direct, rapid, and experimentally convenient, 
and no critical control of orientation of highly 
elevated equipment would be necessary. Slight 
polarization impurity would cause only small 
inaccuracy of error determination, instead of 
seriously obscuring the significance of the re¬ 
sults; careful design of the test source would 
nevertheless be required to maintain fairly 
good purity. 

At the large distances so necessary to insure 
freedom from confusing local-field effects, full 
freedom of control of position of the test 
source is only possible by supporting the 
source from an aircraft. Airplanes are not 
convenient for such work but a nonmetallic 
dirigible airship would be very valuable. Cap¬ 
tive balloons are inferior to dirigibles but per¬ 
haps more practical; they should be of good 
aerodynamic form, be lightly loaded and carry a 
source which does not require adjustment of 
orientation in flight. A captive balloon has 
been found quite useful even though all three 
of these conditions for satisfactory operation 
were violated. Tall towers or poles provide 
very convenient support but their range of 
usefulness is necessarily limited. 

Sites for testing d-f performance must be 
much more critically chosen than even d-f 
operating sites. They must be clear, flat, and 
electrically homogeneous over a radius of sev¬ 
eral tens of wavelengths at the lowest fre¬ 
quency to be used and to sufficient depth to 
attenuate the transmitted wave by ten times. 
Wastelands are fortunately very suitable, salt 
marshes as sites of high conductivity and 
deserts as sites of low conductivity. 

By use of a source of the type proposed sup¬ 
ported from an aircraft over well chosen sites 
and working at adequate distances, d-f per¬ 
formance should be assessable with an ease, 
completeness, and reliability not approached in 
any tests hitherto made. Tests by these meth¬ 
ods can be extended to include effects of multi- 
path wave interference and of inhomogeneous 
sites if the art advances sufficiently to permit 
appropriate test conditions to be specified. 


55 PROJECT C-58 9 

5,51 Causes of “Swinging” Bearings 

The original development of the Adcock an¬ 
tenna system was for the purpose of rendering 
an associated direction finder insensitive to 
that component of a radio wave whose electric 
field is polarized perpendicular to the plane of 
incidence (horizontally polarized). Theoretical 
computations, as well as tests with a controlled 
local target transmitter, indicated that the 
Adcock antennas developed under Project 
C-34 10 were capable of discriminating to a very 
high degree between the desired vertically po¬ 
larized and undesired horizontally polarized 
waves of a radio signal. Nevertheless, tests on 
sky-wave transmissions revealed swinging 
bearings on the cathode-ray indicator of the 
direction finder, typical of so-called polarization 
error. Both the magnitude of the bearing 
oscillation and the percentage of time that the 
cathode-ray indication departed from the cor¬ 
rect azimuth made it appear likely either that 
theoretical computations of wave discrimina¬ 
tion for these antennas were grossly in error, 
or that the downcoming sky waves were not 
polarized at random according to the generally 
accepted hypothesis of ionosphere reflections, 
if it were assumed that the swinging of the 
bearing was due to polarization error. 

Because the values of polarization discrimi¬ 
nation by Adcock antennas were more or less 
substantiated by tests with local target trans¬ 
mitters, while the distribution of polarization 
of downcoming sky waves remained unproven 
by any physical tests whose results could be 
directly associated with the apparent polariza¬ 
tion of Adcock direction finders, the desirabil¬ 
ity of tests on the polarization of downcoming 
sky waves was clearly indicated. A part of 
Project C-58 e concerns the study of polarization 
of radio waves between 5 and 20 me. 

In agreement with this contract, there was 
built and installed at Great River, New York, 
an equipment since called the polariscope, a 
description of which follows. This equipment 
permits the ratio between the vertical electric 


e Contract No. OEMsr-745, Federal Telephone and 
Radio Corporation. 




108 


ERRORS IN DIRECTION FINDERS 


and horizontal electric components of a radio 
wave to be seen at a certain point while radio 
bearings are observed on a cathode-ray indi¬ 
cator also to be described below. 

The Bureau of Standards, aware of these 
facts, asked the contractor to observe bearings 
and polarizations of the Bureau’s station 
WWV, Beltsville, Maryland, transmitting suc¬ 
cessively with two different types of antennas, 


equipment is given in Figure 3. The Type A 
indicator is identical with the d-f indicator 
used in SCR-502. Figure 4 is a photograph 
of the antennas used with the polariscope, and 
these antennas consist of two crossed dipoles 
12 ft in length mounted on a revolving boom 
20 ft long. This entire boom with its central 
column can be revolved from within the con¬ 
trol room so that it may face the direction from 



Figure 3. Block diagram of polariscope. 


radiating at certain times vertical and at other 
times horizontal polarization. This transmit¬ 
ter is sufficiently distant from Great River that 
no ground wave is present. 

5,5,2 Description of Polariscope 

The circuits and antenna design of the direc¬ 
tion finder are approximately the same as 
described in Chapter 9, dealing with the SCR- 
502 (Project C-34). 10 

A block diagram of the polariscope and d-f 


which the signal arrives. This assembly is 
mounted on a small wooden tower with its 
central axis about 20 ft above the surface of 
the earth. 

Each antenna, both horizontal and vertical, 
has its own balanced cathode-follower coupling 
unit, which in turn feeds into a balanced dual 
coaxial cable leading into the central control 
room about 50 ft away. The vertical antenna 
is connected to the vertical stator of the goni¬ 
ometer, and the horizontal antenna to the hori¬ 
zontal stator of the goniometer. In this manner, 


















































































PROJECT C-58 


109 



Figure 4. Antenna used with polariscope studies. 












110 


ERRORS IN DIRECTION FINDERS 


the pattern developed on the face of the indi¬ 
cator unit indicates the amplitude and phase 
relationship between the vertical and hori¬ 
zontal components of the electric field of the 
received wave. 

In addition to the visual indication on the 
cathode-ray oscilloscope indicator, automatic 
recordings of the amplitudes of the two com¬ 
ponents of the wave are made by two Ester- 
line-Angus recorders, with a paper speed of 3 
in. per minute, set up in conjunction with two 
separate receiving channels fed by the two 
dipoles of the polariscope. 

The program for the reception of WWV was 
to observe the bearings and the polarization 
for 5 minutes; then to align the two receiv¬ 
ers alike and record the intensities of the two 
components for 5 minutes. After this the 
bearings and polarization were visually ob¬ 
served for 5 minutes again. The whole 
procedure was repeated every 15 minutes for 
each type of transmission. The last half of 
each hour was used for a standby period, 
during which time the two receivers were re¬ 
calibrated for identical sensitivities with the 
next frequency to be tested. 

Analysis of WWV Observations 

Three days’ operation of this equipment re¬ 
sulted in indications and records of bearing 
errors with wave polarization, the analysis of 
which is as follows: 

1. The horizontal vector of downcoming 
waves from both the vertically polarized trans¬ 
mitter and the horizontally polarized transmit¬ 
ter was found to have random polarization. 
This was in accord with the generally accepted 
theories on polarization of sky waves at these 
frequencies. 

2. During periods when the sky-wave po¬ 
larization was horizontal or a very few degrees 
from horizontal, bearing indication was in 
error or indeterminate. This was the expected 
polarization error. 

3. Frequently, even when the sky wave was 
vertically or nearly vertically polarized, there 
were wild oscillations of the bearing indicator 
and deterioration of the null. 

The above results indicated that the oscilla¬ 
tions of the d-f bearing were not due solely to 
polarization error as had been assumed. The 


oscillation of the bearing during periods when 
the wave was approximately vertically polari¬ 
zed must be due to some other phenomenon. 
The following hypothesis and tests were an 
outgrowth of the analysis of the above polar- 
imeter investigations. 

Wave Interference Errors. The hypothesis 
which assumed swinging bearings to be due 
to strong horizontally polarized components in 
the downcoming sky wave, in general also 
assumed that the wave was reflected from a 
single point in the ionosphere. Were this the 
case, then swinging bearings would feasibly 
be due only to horizontal polarized sky waves. 

But consider the result of combined waves 
from two or more points in the ionosphere. If 
these reflection points differ even by a degree or 
less, the combined electric vector at the d-f an¬ 
tenna will be a function of the instantaneous 
phase difference between the several com¬ 
bined waves. 

An analysis of more than two rays becomes 
extremely involved, therefore the combination 
of only two waves will be discussed here. The 
result of the combination of several rays 
which, in practice, frequently arrive at the 
direction finder from a single transmitter, will 
be a still greater variation in the bearing 
indications. 

Figure 5 is the special example of two rays 
whose azimuths differ (for the sake of clarity) 
by a much larger angle than that usually ex¬ 
perienced in practice. In practice, relative 
magnitudes of the separate waves vary as do 
their instantaneous phases. This is due to the 
fact that the separate rays apparently arrive 
from regions in the ionosphere which differ in 
height and density of ionization and where 
ionization conditions are not necessarily stable. 

A simple example of the mechanism whereby 
two vertically polarized waves from slightly 
different azimuths can result in a large d-f 
error is as follows: 

First consider ray C in Figure 5 to arrive at 
point O with its instantaneous error vector 
directed upwards and ray D to arrive at the 
same point with its vector similarly directed 
upwards. As long as these vectors remain in 
such phase, a direction finder located at point 
0 will provide an indication between C and D. 
(Since in practice C and D usually differ by a 



PROJECT C-58 


111 


very small angle, this may be called the correct 
bearing indication.) 

Now consider a later time when the ray 
from C, while still vertically polarized, has 
altered its phase with respect to the ray from 


the amplitude of the waves from C and D were 
identical, the signal would be very weak and 
the d-f bearing indication would be 90° in 
error. With unequal amplitudes, the error is 
less than 90°. 



Figure 5. Equiphase curves for resultant of two vertically polarized waves of different magnitudes moving 
in different directions. 


D by 180°. Then the desired vertical compo¬ 
nents tend to cancel while the component of 
the vector which bisects the angle between C 
and D tends to be additive. In such a case, if 


In a typical situation, the combination of 
two rays will occur with varying phase and 
relative amplitudes, thus providing an oscil¬ 
lating indication of bearing. The combination 















































112 


ERRORS IN DIRECTION FINDERS 


of more than two rays will, of course, increase 
the complexity of the resultant vector to which 
the direction finder responds. 

Because in the crossed Adcock direction 
finder there is in general a small azimuth error 
which varies with the vertical angle of in¬ 
cidence (the octantal error), two rays arriving 
from the same azimuth direction, but from 
different layers of the ionosphere will also re¬ 
sult in an interference error whose behavior 
is similar to that due to two rays arriving 
from slightly different azimuths. 

Test of Wave Interference Errors. To exam¬ 
ine the error resulting from the interference 
of two waves whose azimuth and relative phase 
may be controlled, the direction finder em¬ 
ployed in the previously described polarization 
test was used in the reception of signals from 
two local target transmitters. 

These two transmitters were located about 
1,000 ft away from the Adcock antennas of the 
direction finder, and were spaced so that the 
azimuths of arrival differed by about 2°. One 
of the transmitters was set on a given fre¬ 
quency (about 5 me), while the other transmit¬ 
ter frequency was varied by hand to be as 
close as possible to the frequency of the fixed 
transmitter. 

As the frequencies of the two transmitters 
fell within the band width of the direction 
finder, a confused and rapidly changing fluc¬ 
tuation was observed which in general pointed 
toward the two transmitters. 

When the two transmitter frequencies were 
brought as close together as was possible (for 
brief periods two frequencies were apparently 
within 1 or 2 cycles), the d-f indication was 
that of a slowly oscillating bearing whose 
quality was highest when it pointed in the 
direction of the two transmitters and which 
deteriorated to almost no indication when it 
approached a bearing 90° from the line bi¬ 
secting the angle between the two transmitters. 

It was suggested that a further experiment 
to investigate precisely the errors due to any 
given phase difference and amplitude between 
the two incoming rays could be performed by 
feeding two displaced transmitting antennas 
from a single transmitter with a phase and 
amplitude adjustment in the line to one of the 


antennas. But, because the test with the two 
separate transmitters satisfactorily proved 
that two vertically polarized waves arriving 
from slightly different azimuths can cause 
bearing oscillations resembling polarization 
errors, it was decided that, in view of the fact 
that a combination of only two rays was artifi¬ 
cial, further tests of this type should not be 
pursued at this time. 

Conclusions 

The experience with the polariscope and the 
later tests which simulated two sky waves have 
indicated that the d-f bearing oscillations and 
deterioration nulls are not due entirely to po¬ 
larization error. 

In America it has been assumed that efforts 
to disseminate horizontally polarized waves 
would ultimately result in a direction finder 
whose bearings would be steady and precise 
beyond those which had been designed in the 
past. 

The facts that the prevalent multiple-ray 
transmission of radio signals gives rise to in¬ 
terference errors in Adcock direction finders 
would indicate that too great effort in reduc¬ 
tion of polarization error are not warranted. 

At present, it appears that the interference 
error cannot be reduced by any d-f system in 
which the antennas cover such a limited area 
as do the Adcock antennas. The Musa (mul¬ 
tiple unit steerable antenna developed by Bell 
Telephone Laboratories), does reduce inter¬ 
ference errors. The Musa system, however, is 
necessarily a very large installation which can 
be used for direction finding over a very small 
azimuth only. 

56 PROJECT 13.1-84 

Under Project 13.1-84 f a very great deal of 
work was done to determine the essential char¬ 
acteristics of the ground under d-f installations 
to make the apparatus as useful and as free 
from errors as possible. Part of the project 
was to develop, if possible, simple equipment 
which a relatively untrained person could take 
to a site selected for a d-f installation, perform 
a simple and not too critical experiment, and 
by means as simple as reading a meter, perhaps 

£ Contract No. OEMsr-1026, Federal Telephone and 
Radio Corporation. 




PROJECT 13.1-84 


113 


like a tube-testing meter, determine whether 
the site was suitable or not. All previous work 
under Division 13 projects had indicated the 
extreme importance of locating d-f apparatus 
on sites with good ground characteristics. 

The final report 11 on Project 13.1-84 shows 
that the phenomena involved are too complex 
for a single instrument of simple type to be 
constructed for the job to be done. 

An extensive bibliography is contained in 
the final report 11 and this report should be 
consulted by anyone seriously engaged in site 
investigations. The bibliography has references 
to characteristics of the soil as of interest to 
a chemist, or from the standpoint of electro¬ 
chemistry or physical chemistry. The report 
itself contains much historical material deal¬ 
ing with our knowledge of the conductivity of 
the soil, its dielectric behavior, its d-c and h-f 
resistance, and of methods explored for deter¬ 
mining these factors. 

5,6,1 Methods of Measurement Studied 

A comparison of resistivities as measured by 
direct currents and by alternating currents 
indicates that electrode effects cause the d-c 
resistivities measured at high radio frequencies 
before a dispersion occurs. 

Nevertheless the final report indicates that 
d-c measurements may be a very practical 
method for determining resistivity of soil 
samples. 

R-F Measurements 

Considering the soil as an imperfect dielec¬ 
tric, a whole series of experiments was per¬ 
formed to determine the relative qualities of 
soils as dielectrics. By the use of a Q-meter, in 
which the soil is inserted between the plates of 
a capacitor and the overall loss factor of soil 
plus capacitor determined, the conclusion was 
reached that Q-meter measurements were not 
practicable over a wide frequency range. The 
method is not suitable for measuring resis¬ 
tance of sample soils greater than 50,000 ohms, 
the reliability of the method increasing as the 
resistance decreases. 

The conclusion was reached that the Q-meter 
method could be used at spot frequencies for 
soil samples in Lucite containers. 


Since it is well known that the inductance 
of a coil at radio frequencies depends upon the 
core material, the Q-meter can be used to mea¬ 
sure soil characteristics by placing the sample 
inside a coil whose inductance, without the 
soil, is known. 

It was found that measurements by this 
method did not give quantitative results but 
with care could be made to show relative 
quality. The inductance method is more sensi¬ 
tive and more easily applied than the capaci¬ 
tance method. 

Bridge Methods 

The most accurate way of measuring the im¬ 
pedance of a soil sample at radio frequencies 
is to use a suitable bridge circuit. This method 
requires more skill, is more tedious, and the 
range of values measurable is less than in some 
other methods. In addition an auxiliary 
generator and detector are required. 

Because of variations of the weather and 
the averaging of soil constants in wave propa¬ 
gation, it seems unnecessary to measure the 
conductivity with an accuracy greater than 50 
per cent of the mean value of several measure¬ 
ments. Furthermore, measurement of conduc¬ 
tivity alone seems to be all that is necessary 
to determine the characteristics of ground 
material of a possible site for a d-f installa¬ 
tion. Thus the bridge method, while yielding 
a high degree of precision, is too complex for 
the job to be done. 

Methods Using Antennas and Transmission 
Lines 

One of the most effective means of determin¬ 
ing the effect of a site upon a d-f installation 
is to set up a portable direction finder of known 
properties and to observe how its operation is 
affected by the site. 

Thus an antenna and its characteristics are 
a function of the ground upon which it is 
erected, its input impedance, the ground losses, 
and directional patterns being functions of the 
ground. Properties of transmission lines which 
are most susceptible to investigation are at¬ 
tenuation constant and velocity of propagation. 
Preliminary measurements indicated that such 
studies could be quite effective but the decision 
was reached that more work would be neces- 



114 


ERRORS IN DIRECTION FINDERS 


sary to find out if the methods would be prac¬ 
ticable. Furthermore, such experiments could 
be performed only with engineering super¬ 
vision, such was the state of the art of mea¬ 
suring equipment of the kind required for an¬ 
tenna or transmission-line measurements. 

Similarly, measurements of field strength at 
different frequencies were abandoned when it 
was found that results were inconclusive. Re¬ 
flection coefficients and wave tilt were studied 
as a function of ground constants. Limited ex¬ 
perience indicates that the method is applicable 
to measurements of soil constants under con¬ 
ditions of (1) no obstructions between trans¬ 
mitter and receiver, (2) no reradiators near 
enough to cause trouble, (3) elevation of the 
target transmitter above ground or else use of 
rather large power input. These limitations 
were discouraging from the standpoint of port¬ 
ability and the necessity of determining ground 
characteristics under varied conditions. 

Audio-Frequency Methods 

Mapping a site by plotting equipotential 
lines between ground electrodes at audio fre¬ 
quency required less time than mapping by 
plotting equifield lines about a transmitting 
antenna at radio frequencies. The use of audio 
frequencies and ground rods eliminates pro¬ 
nounced disturbances caused by above-ground 
reradiators observed in plotting equifield r-f 
field lines. The method employed in Project 
13.1-84 is described in the final report 11 and 
is applicable to the picking of a site for a d-f 
station. The method can be used, also, for 
locating large bodies of metal under the sur¬ 
face of the ground and this is discussed in the 
final report, together with the use of r-f 
devices such as mine detectors for locating 
small metallic bodies. Circuits for such devices 
as constructed under the project will be found 
in the final report. 

Wenner-Gish-Rooney Method 

In this manner the following process is car¬ 
ried out: four copper-coated rods % in. in 
diameter and 1 ft long are used as electrodes. 
They are equally spaced along a straight line 
and voltmeter readings are taken for several 


values of spacing between 1 and 35 ft. A bat¬ 
tery-operated vibrator delivering approxi¬ 
mately a square-wave alternating current of 
110 volts is connected through a milliammeter 
to the outer electrodes and a battery-operated 
vacuum-tube voltmeter is connected between 
the inner electrodes. The current flowing 
through the outer electrodes and the voltage 
between the inner electrodes are read for each 
of the chosen spacings. At close spacings the 
electrodes are driven into the ground only an 
inch or so, at greater spacings the electrodes go 
into the ground to depths of up to 1 ft. 

In this manner a plot of an area showing 
effective resistance as a function of depth can 
be obtained. The method is easy to apply, is 
sufficiently accurate in indicating the resistiv¬ 
ity of the top surface of the ground and is the 
best method of obtaining in a qualitative man¬ 
ner the resistance as a function of depth. 

The final report ends with some data on the 
ground requirements for direction finding in 
various frequency bands, indicating in a par¬ 
ticular case that 50 tons of coal dust screen¬ 
ings, either soft or hard coal, should be put 
down to a depth of 1 in. under installed ground 
mats and to a radius of 10 ft beyond the guy 
wire anchors. The ground mats and the coal 
dust layer are covered to a thickness of 3 in. 
and this is tamped down tightly. 

Treatment of this sort produced a ground 
which contributed little trouble to the d-f sta¬ 
tion involved. 

5.7 PROJECT C-19 12 

The loop direction finder has been found 
lacking as a dependable instrument for naviga¬ 
tional and other purposes because of inac¬ 
curacies under certain operating conditions. It 
is often impossible to get a bearing at all or the 
azimuth of the observed bearing may be greatly 
in error or may vary from moment to moment. 
These errors have been under continual study 
since the loop direction finder came under prac¬ 
tical use during World War I and the basic 
causes for the different types of errors are now 
well understood. Most of the errors have 
proven capable of elimination, but a notable 
exception has been polarization error which in¬ 
cludes the so-called night effect and airplane 






PROJECT C-19 


115 


effect. Project C-19 g was to study a particular 
and new means for attacking this type of error. 
Project 13-122 13 studied this compensation 
means critically and reported on difficulties 
with it. 

5,71 Normal Loop Operation 

The ideal case for loop operation is a verti¬ 
cally polarized wave (in which the electric 
vector is always in the vertical plane through 
the direction of travel) proceeding along the 
surface of the earth and following a great-circle 
path between the transmitter and the receiver. 
In this situation the loop has the well-known 
“figure eight” directional response, a mini¬ 
mum or null being obtained when the plane of 
the loop is at right angles to the direction of 
arrival of the signal. 

Under these conditions and if the loop sys¬ 
tem itself has no “instrumental” errors, the 
bearing of the distant station can be ascer¬ 
tained with considerable accuracy. 

5 72 Wave Errors 

If, however, there are abnormalities in the 
wave itself, a loop which will operate perfectly 
on normal waves will show errors in bearing, 
hazy bearings or no bearing at all. 

The several wave errors are as follows: 

1 . Coastal refraction is the phenomenon re¬ 
sulting when the received signal travels 
obliquely across a boundary between two soil 
types, notably ocean-to-shore transition at a d-f 
site located some distance from the coast line. 
The wave is actually refracted and appears to 
come from an incorrect direction. 

2 . Lateral deviation is a phenomenon in 
which the wave does not travel a great-circle 
course but deviates as much as 10° from this 
course. 

3. Scattered signals is another form of wave 
error in which the signals seem to arrive from 
several directions, apparently from scatter 
sources in the ionosphere or on the earth’s 
surface which appear to reradiate some of the 
original energy. 


g Contract No. NDCrc-159, Stanford University. 


5 7 3 Polarization Errors 

Far more common than the anomalies of 
scattered signals and lateral deviation are the 
errors due to irregular polarization of the re¬ 
ceived wave. The symptoms are of several types 
as follows: (1) Bearing sharply defined and 
stable but apparent azimuth incorrect; (2) 
bearing sharply defined but shifting in direc¬ 
tion over a period of time, often quite rapidly; 
(3) blurred, indefinite null point, although a 
minimum of correct bearing may be detected. 

Polarization errors occur when the received 
wave is not a simple, vertically polarized wave 
but contains a horizontal component as well. 
This horizontal component arises from the 
rotation of the plane of polarization of the sky 
wave in its reflection from the ionosphere. 
When such a wave arrives at £ d-f station, the 
operator turns the loop to get a null indication 
but is able to do so only when he has oriented 
the loop in such a manner that loop voltages 
due to the vertical component (vertical loop 
conductors) and due to the horizontal compo¬ 
nent (horizontal loop conductors) are equal and 
opposite. This is not the loop position which 
gives a null on the vertical component only be¬ 
cause the angle of arrival is such that there is a 
phase difference between the two components. 
Therefore, the operator gets a wrong bearing. 

574 Attacking the Problem 

Two possible modes of attacking this prob¬ 
lem present themselves. One possibility is to 
design a collector with only vertical elements. 
This leads to the Adcock antenna which is quite 
useful for many applications. Its great disad¬ 
vantage is the fact that its pickup, unless the 
structure is quite large, is small whereas the 
loop can have many turns with correspondingly 
greater sensitivity. 

The mode of attack pursued under Project 
C-19 was to accept the situation of having trou¬ 
blesome errors due to the horizontal pickup but 
to compensate the unwanted voltage by another 
horizontal voltage secured from an additional 
antenna mounted with the loop and rotating 
with it. This forms the so-called compensated 
loop which has been discussed in the literature 
and on which patents have been granted. 14 




116 


ERRORS IN DIRECTION FINDERS 


In the system proposed, the voltage induced 
in the auxiliary antenna would be expected to 
behave in the same manner as the voltage 
induced in the loop by the horizontal polariza¬ 
tion. Then this voltage would be coupled into 
the loop in such a manner as to provide neu¬ 
tralization for the unwanted voltage. 

Two basic problems are to be solved. First, 
what must be the network characteristics used 
for coupling the neutralizing voltage into the 
loop and, second, to what extent does the neu¬ 
tralization become incomplete if one of the 
operating variables change. 

The bulk of the final report is devoted to a 
study of these basic problems including the 
effect of a wave reflected from the earth’s sur¬ 
face, the effects of vertical and horizontal 
polarization or a combination of the two, errors 
in the uncompensated loop and in the compen¬ 
sated arrangement, calculations on typical 
situations, the effect of wavelength on compen¬ 
sation, variation of height of antenna above 
ground, and height of auxiliary antenna with 
respect to the loop. There is considerable ma¬ 
terial relating to the measurement of ground 
reflection coefficients, voltage ratios and phase 
angles, field strength, etc. 

Results Obtained 

As a result of the theoretical analysis and 
extensive field tests, it is concluded that the 
system would work for short as well as for long 
waves, calculations being given for a range of 
from 1- to 1,000-meters wavelength, that it will 
function for any type of soil condition, that it 
will not work on airplanes where extreme 
changes in soil type would occur over which the 
plane flies or where large variations in height 
above ground would occur. The system works 
best at fixed heights which are small (A/10 or 
less) compared to the wavelength. 

On actual demonstration of experimental 
equipment and a Sperry Mk-1 automatic direc¬ 
tion finder good compensation was secured. 

An extensive bibliography is included in the 
final report. 12 

5,7-6 Compensated-Loop Direction Finder 

The Signal Corps of the U. S. Army in Janu¬ 
ary 1944, requested NDRC to perform research 


on a loop antenna satisfactory for direction 
finding on transmitters up to 30 miles away in 
the h-f band which would be as satisfactory at 
nighttime as during the day. 

Under the continuing Project C-58, 9 some 
investigations were made on compensated-loop 
direction finders. 

In the past considerable work has been per¬ 
formed in attempts to compensate loop anten¬ 
nas against response to horizontally polarized 
waves. In almost all such investigations the 
general problem of downcoming sky waves at 
any angle has been attacked. The failure to 
design a satisfactory compensated-loop direc¬ 
tion finder may have been due to the too gen¬ 
eral nature of the problem. 

A loop direction finder compensated against 
night errors for transmissions of not more 
than 30 miles would require that compensation 
be against vertically or nearly vertically down¬ 
coming waves only. It might be expected that 
this special problem could be solved more easily 
than the general loop compensation which had 
as yet no satisfactory practical solution. 

Although the final form of compensated loop 
might for reasons of portability be a single 
rotatable loop antenna with the necessary at¬ 
tachments for response to horizontally polar¬ 
ized downcoming waves, it was decided that for 
reasons of convenience during the experiments 
a fixed crossed loop be employed. 

Directly below the crossed loops were in¬ 
stalled crossed horizontal dipoles. An injection- 
loop transmitter was located 30 ft directly 
above this collector, to generate the vertically 
downcoming wave. 

Both the loop antennas and the horizontal 
dipole antennas fed cathode-follower coupling 
units. In the dipole coupling units both ampli¬ 
tude and phase were adjustable. 

Experiments were made at night on a trans¬ 
mitter located 25 miles away. The loop trans¬ 
mitter located above the collector assembly was 
tuned to the frequency of the distant transmit¬ 
ter and then the compensating dipole antenna 
coupling units were adjusted to minimize the 
signal. A reduction of about 10 db was easily 
accomplished. The distant transmitter was then 
turned on, and it was found that on the 
cathode-ray indicator the swinging of the bear- 




PROJECT 13-122 


117 


ing was reduced from four to eight times as 
compared with the uncompensated loop. 

It was found that the improvement was best 
when the injection loop transmitted at precisely 
the same frequency as the distant transmitter. 
Also, an adjustment made when the ground 
under the loops was dry became worthless when 
a brief rain altered the conductivity of the 
ground. On the whole, the compensation was 
not complete, and the apparent requirement for 
adjusting the coupling units by means of an in¬ 
jection signal exactly the same frequency as the 
distant signal was a serious limitation. During 
the experiment it was also found that a slight 
frequency shift by the injection transmitter re¬ 
quired a large adjustment in the coupling unit 
controls. 

In the May 1945 issue of Proceedings of the 
I.R.E. 15 an article by J. N. Pettit and A. W. 
Terman on compensated-loop direction finders 
concluded with some encouraging remarks on 
the possibility of compensating a loop antenna 
by means of a horizontal dipole. Because this 
conclusion apparently differs from that reached 
in the report of Project C-58 on compensated 
loops, the NDRC requested a comparison and 
discussion of the two reports to resolve the 
apparent contradictions. 

5-8 PROJECT 13-122 13 

581 Discussion of Project C-19 

Under Project 13-122, h a final report was 
prepared which discusses the work accom¬ 
plished under Project C-19. 12 The gist of this 
discussion follows. 

The compensated loop was studied under 
Project C-58 and the report on that project 
states that results were rather discouraging. 
The important item to be determined is whether 
the findings of Project C-19 were corroborated 
by the work in Project C-58 or whether there 
is some basic difference between the results. 
It is concluded that, basically, there is no theo¬ 
retical disagreement. However, it is shown that 
the coupling networks should, if possible, in¬ 
clude means for resolving the differences in the 

h Contract No. OEMsr-1490, Federal Telephone and 
Radio Corporation. 


internal impedances of the loop and dipole 
antennas. 

The report of Project C-19 on the investiga¬ 
tion of compensation in direction finders is a 
mathematical investigation to determine the 
phase angle and the amplitude ratio between 
the voltage induced in a loop antenna by a 
downcoming horizontally polarized wave, and 
the voltage induced by the,same wave in a hori¬ 
zontal dipole mounted at the center of the loop. 
It was shown mathematically that in the pres¬ 
ence of grounds of medium conductivity, or 
better, this amplitude ratio and phase differ¬ 
ence remain nearly constant for varying angles 
of incidence, and for varying frequency. For 
instance, with wet soil, between the wave¬ 
lengths of 1 and 20 meters, the amplitude ratio 
varies from 6.4 to 6.8 and the phase shift varies 
from 9.9° to 8.5°. These are calculations of the 
voltages induced into the antennas and do not 
take into account the internal impedances of 
the antennas. Assuming that the voltages, once 
they were introduced into the antennas are 
available, the report shows that the compen¬ 
sation requirements vary slowly with fre¬ 
quency; and that for various types of soil, 
except very dry soil, the ratio of the two volt¬ 
ages and the phase shift between them remain 
constant, provided that the antennas are 
mounted less than A/10 above the ground. 

Coupling Network 

It was concluded that it was necessary to 
design a circuit which would give constant 
phase shift, constant amplitude ratio, and good 
stability with varying frequency. Such a cir¬ 
cuit was designed under C-58. For convenience 
of indication a crossed-loop system with two 
horizontal dipoles was used. An instantaneous 
cathode-ray indicator for bearing indication 
was employed. There were direct low-impedance 
connections between the loops and the goniome¬ 
ter. Each dipole antenna was then coupled to 
the corresponding low-impedance connection 
through a set of two balanced cathode-follower 
coupling units. One cathode follower operated 
without phase shift and the other cathode fol¬ 
lower in the set had its phase shifted by 90°, so 
that by combining the two and changing their 




118 


ERRORS IN DIRECTION FINDERS 


relative gains, the output phase could be shifted 
from 0° to 90°. 

Although this coupling unit is of the type 
that was indicated by the conclusion of the C-19 
report, it does not take into account the vary¬ 
ing impedance of the dipole antenna with fre¬ 
quency and the varying impedance of the loop 
antenna with frequency. To employ the ratio 
of the two induced voltages, it would be neces¬ 
sary, if it is at all possible, to obtain these volt¬ 
ages, for combination, without any phase shift, 
or amplitude ratio shift, introduced by either 
internal impedances or external added impe¬ 
dances in the antenna units. 

The final proof submitted in Project C-19 
was a d-f test at one frequency and at one 
downcoming angle. The direction finder was 
adjusted for good results with the target trans¬ 
mitter and it is shown that the type of polar¬ 
ization transmitted by the target transmitter 
does not thereafter introduce any error. This 
test was repeated in Project C-58 as stated in 
their report of July 1943. 16 For the purpose a 
polarized transmitter was installed atop a 90-ft 
tower. However, in the report of September 
28, 1943, 17 on the problem of making the ad¬ 
justments with the target transmitter, it is 
noted that without the help of a target trans¬ 
mitter producing a downcoming wave at the 
frequency of the transmitter to be observed, 
the various adjustments of amplitude and phase 
cannot be carried out with certainty, and that 
the practical development of such a system for 
the Armed Forces did not look promising. 

Comparison of Reports 

The final report on C-58 contains no find¬ 
ings in conflict with the results of C-19. The 
report of May 28, 1943 (C-58) 18 states that 
the phase difference seems to remain constant, 
but a great deal of difficulty is encountered in 
checking the amplitude relationships, since 
they seem to vary. It is also stated in the 
report of July 1943, 16 that “the phase and am¬ 
plitude relationships remain constant over 


long periods of time and the various states of 
polarization.” This finding seems to agree very 
closely with the theoretical calculations made 
under Project C-19. 

Since it was necessary to work for a practi¬ 
cal solution from the mathematical conclusions, 
it was necessary to investigate the amplitude 
and phase relationships between the two vol¬ 
tages to be opposed as a function of: (1) po¬ 
larization, (2) ground angle of the sky wave, 
(3) frequency, and (4) ground constants. 

Once these relationships were proven to be 
constant, or very nearly so, it was necessary 
to devise some circuits which could be adjusted 
easily and with certainty. In the report of 
September 28, 1943 17 it is stated that a target 
transmitter is needed for making these adjust¬ 
ments. This seems to be a very reasonable 
assumption unless the ground conditions can 
be measured (which would be rather un¬ 
reliable, since the ground might vary over very 
large areas), and the adjustments to be made 
then calculated from those measurements. This 
solution did not seem practical for a useful 
military direction finder. 

Conclusions 

The investigations under Project C-58 on 
compensated loops revealed a problem not men¬ 
tioned in the Pettit and Terman report. That 
is, the varying impedances of the antennas 
with varying frequency and ground conditions 
effectively prevent the use of the voltages 
induced in infinite-impedance antennas to com¬ 
pensate against horizontally polarized waves. 
The voltages discussed in the former report 
must be assumed to exist in infinite-impedance 
antennas, but such antennas are not available 
in practice. Since the loop antenna’s principal 
value is that it may be tuned, and when tuned 
its impedance is a critical function of frequen¬ 
cy, the conditions of infinite impedance for an¬ 
tennas in a compensated-loop system are not 
practicable. 






Chapter 6 


CORRELATION OF D-F ERRORS WITH IONOSPHERE MEASUREMENTS 


P RIOR to the war no coordinated study of 
ionosphere transmission and d-f errors at 
high radio frequencies had ever been attempted. 
Such a study was desirable from the stand¬ 
point of determining the causes of deviations 
from great-circle transmission paths and to 
establish criteria for the presence and extent 
of d-f errors caused by the ionosphere. 

At a series of conferences called by Division 
13, NDRC, and beginning in late January 1941, 
plans were made for systematic observations of 
ionosphere characteristics and d-f errors in 
the range 2 to 30 me in which waves are re¬ 
flected from the ionosphere. The general plan 
was to have simultaneous ionosphere and d-f 
observations at a number of points on this 
continent. As a result projects were set up in 
Division 13 to implement this coordinated study. 
Numerous ionosphere laboratories furnished 
data and numerous d-f stations furnished bear¬ 
ing information over considerable periods. The 
work was coordinated and cleared through the 
National Bureau of Standards [NBS] to whom 
the observations were sent. 

The ionosphere reports submitted under 
these several projects were used by the various 
branches of the Armed Forces. The establish¬ 
ment of channels for reception of incoming 
data and techniques for processing it led to the 
development of a service known as the Inter¬ 
service Radio Propagation Laboratory [IRPL] 
devoted to prediction and forecasting of h-f 
radio propagation conditions on a worldwide 
basis. The advantages of this work to the com¬ 
munications of the Armed Forces during the 
war are obvious. 

61 PROJECT C-13 

The several projects in Division 13 dealing 
with the coordination of ionosphere measure¬ 
ments and d-f errors are C-13, 13.2-88, 13.2-90, 
13.2-91, 13.2-92, and 13.2-99. 

Section IV of the final report on Project 
C-13 1 will serve to show future investigators in 


correlating d-f errors and ionosphere condi¬ 
tions what was attempted and will offer valu¬ 
able suggestions as to the layout of the job to 
do this kind of work. Studied in connection 
with the final report on Project 13.2-92 2 and 
the bimonthly reports of the IRPL-G series 
beginning with IRPL-G1, July-August 1944, 
the early groundwork for the present improved 
services performed may be ascertained. 

Section III of the C-13 report 1 describes the 
apparatus used. Better and simpler equipment 
was subsequently developed. Section V indi¬ 
cates the progress of the project with applica¬ 
tion to radio transmission up to the date of the 
end of the project. Section II summarizes 
types of normal and abnormal ionosphere and 
field-intensity characteristics observed and 
shows some of these in the form of graphs and 
of continuous records of relative field intensi¬ 
ties over certain propagation paths. The origi¬ 
nal tabulations and records are on file at NBS 
and the cooperating laboratories. 

62 PROJECT 13.2-92 

At the termination of Project C-13 a new 
project, 13.2-92, was instituted. The work 
accomplished under this project is as follows. 

The correlation of d-f errors and causative 
ionosphere conditions was carried out by five 
cooperating laboratories located in Washing¬ 
ton, D. C., Alaska, California, Puerto Rico, and 
Massachusetts. 

The d-f measurements at all the laboratories 
were made with the Navy type DAB spaced- 
loop direction finder. This and the other equip¬ 
ment employed are described in the final 
report on Project 13.2-92. 2 Measurements were 
made on a large number of stations distributed 
in azimuth, distance, and frequency. The re¬ 
sults obtained on approximately thirty repre¬ 
sentative stations dealt with in the report show 
relationships of bearing errors, field intensi¬ 
ties, maximum usable frequencies, and skip 
distances, geomagnetic disturbances, absorp- 


119 


120 


CORRELATION OF D-F ERRORS WITH IONOSPHERE MEASUREMENTS 


tion, and transmitter antenna directivity. 
Mention is made of effects of sporadic-E, 
scattering, and ionosphere disturbances. 

The results demonstrated that deviations, 
often in excess of 50°, occur in transmissions 
received at the NBS d-f site from stations 
located in England and Germany. The influence 
of auroral absorption zone on bearing accuracy 
over these paths is analyzed and indicates that 
the steep gradient of absorption between paths 
passing near and through the zone reasonably 
accounts for the effects on low operating fre¬ 
quencies. On high operating frequencies, the 
dropping of the calculated maximum usable 
frequency for the path below the value of the 
operating frequency seemed to account fairly 
well for the large deviations. 

Correlation was found between bearing er¬ 
rors and field intensities, the large errors oc¬ 
curring when field intensities are relatively 
weak. Considerable evidence that large errors 
might be predicted at times when the maximum 
usable frequencies fell below the operating fre¬ 
quency was also obtained. Simultaneous oc¬ 
currences of large errors and severe geomag¬ 
netic disturbances were observed on the Berlin- 
Sterling (D.C.) path and on the Daventry- 
Sterling path. Only slight evidence of geomag¬ 
netic disturbances on bearing accuracy for paths 
other than those passing near or through the 
auroral zone was discovered. 

The program was considered sufficiently well 
under way at the end of the project to enable 
its being taken over by IRPL. Thus the spon¬ 
sorship of the project by NDRC ended June 
30, 1944. 

«-3 PROJECT 13.2-88 

The final report 3 on this contract with 
Stanford University deals briefly with choice 
of site and construction, goes into detail on the 
calibration and adjustment of the DAB direc¬ 
tion finder and mentions preliminary conclu¬ 
sions deduced from the results of data observa¬ 
tions. 

Calibration was accomplished by means of a 
target transmitter consisting of a small crystal- 


a Contract No. OEMsr-1122, Stanford University. 


controlled oscillator in a metal case with a 4-ft 
vertical antenna. Measurements were made at 
30° intervals and at 300, 400, 500, and 600 ft 
at fixed frequencies ranging between 2.00 and 
17.32 me. Errors in bands I and II were bad. 
It was found possible to minimize these errors 
by redistributing the loop inductances. The er¬ 
ror in every case was taken as the difference 
between the true bearing and the mean of the 
direct and reciprocal bearings measured by 
the DAB. 

Beginning March 1, 1944, after a prelimi¬ 
nary period of training, regular observations 
were begun on stations in areas suggested by 
NBS. These stations were in Alaska, Russia, 
Mexico, Hawaii, Japan, China, and Australia. 
Data were recorded on weekly summary forms 
and copies sent to NBS. 

For the most part, large deviations were 
observed to occur during periods when the 
maximum usable frequency for the path was 
below the operating frequency of the transmis¬ 
sion being observed. However, exceptions to 
this were noted, especially over multi-hop paths 
in the Pacific area. 

The correlation of bearing deviations with 
field intensity was good, in that nearly all large 
deviations were accompanied by corresponding¬ 
ly low field intensities, although the converse 
was not always true. 

6-4 PROJECT 13.2-90 

The primary object of this project 15 was to set 
up a Navy DAB unit at a site appropriate for 
proximal d-f observations, as a means of study¬ 
ing ionospheric and radio transmission factors 
of importance in the deviation of long distance 
d-f bearings. Actual operation of the equip¬ 
ment was carried out under contract between 
the University of Puerto Rico and IRPL. The 
final report 4 gives the methods of calibration 
employed, the means by which the lower-fre¬ 
quency bands were made to have smaller posi¬ 
tive errors than they originally had, namely, 
by readjusting the loop inductances as was 
done under Project 13.2-88. 


b Contract No. OEMsr-1101, University of Puerto 
Rico. 





PROJECT 13.2-99 


121 


6 - s PROJECT 13.2-91 

Work on this project 0 was carried out in 
Alaska, where the Department of Terrestrial 
Magnetism, Carnegie Institution of Washing¬ 
ton, set up a Navy DAB-3 direction finder near 
the University of Alaska. Direction finder 
measurements were made on a 24-hour basis 5 
in April 1944 and continued until the project 
was taken over in July by IRPL. Among the 
accomplishments were the plotting of mean 
hourly deviations from true bearings of the 
stations observed, and production of scatter 
diagrams of (1) mean diurnal bearing devia¬ 
tions versus mean diurnal geomagnetic K-fig- 
ure, and (2) mean diurnal bearings of observed 
stations for a mean diurnal geomagnetic K- 
figure of 30 (considered normal) on polar 
coordinates, showing the direction of their 
deviation from the true bearing. 

Primary conclusions from these analyses, 
which were continued after the project came 
under IRPL, was that bearing deviations seemed 
roughly to go in the direction of a north- 


south line with increasing geomagnetic K-fig- 
ure. In general it seemed that preliminary re¬ 
sults strongly confirmed the desirability of 
evaluating directional bearings in the light of 
radio wave propagation characteristics, but did 
not show much promise of isolating systematic 
trends to permit application of predetermined 
correction factors for general use. 


«- 6 PROJECT 13.2-99 

As in the other projects of this series, a 
DAB direction finder was set up, this one near 
Cambridge, Massachusetts, by Harvard Uni¬ 
versity, and bearings were taken on the re¬ 
quired stations, the data 6 being submitted to 
the Bureau of Standards. In addition plans 
were prepared for conducting sweep-frequency 
ionospheric observations by automatic equip¬ 
ment. This involved the construction of the 
necessary equipment. This project d was taken 
over by IRPL at the termination of the NDRC 
contract. 


c OEMsr-1151, Carnegie Institution of Washington. 


d Contract No. OEMsr-1252, Harvard University. 





Chapter 7 

MISCELLANEOUS DIRECTION-FINDER RESEARCH 


S EVERAL projects under the supervision of 
Division 13 carried out certain technical 
work on direction finding which is not con¬ 
veniently or logically placed in one of the other 
chapters of this volume. This work is sum¬ 
marized here, to complete the record. 

7.1 TESTS ON DIRECTION-FINDER 
SYSTEMS—PROJECT 13-110 

In July 1945 Section 13.1 of Division 13 set 
forth the general thesis that standards for d-f 
systems should be worked out so that the 
adoption of suitable standardized procedure 
(of testing d-f systems) will result in simplifi¬ 
cation of correlation of data when two or more 
direction finders are to be compared. At the 
time, the Services were employing a wide 
variety of direction finders, differing in respect 
to their collector systems, bearing indicators, 
and methods of resolving bearing information. 
The frequency band coverage included all com¬ 
munication frequencies from very low through 
ultra-high frequencies. 

A conference of representatives of the vari¬ 
ous Service laboratories with members of Divi¬ 
sion 13, NDRC, set up proposed test procedures. 
Central Communications Research, Cruft Labo¬ 
ratory, Harvard University, was assigned the 
task a of investigating the practicability of the 
proposed procedures by applying them to exist¬ 
ing d-f systems and, at the close of the war 
with Japan, four Army units, SCR-502, SCR- 
503, SCR-551 and a developmental model of 
CRD-2, and a Navy DAB installation had been 
set up by the laboratory staff. 1 Some measure¬ 
ments had been made and arrangements were 
completed for continuing the work actively 
under a Navy contract. 

7 2 SURVEY OF AIRBORNE DIRECTION 
FINDERS—PROJECT 13-109 

A survey of existing airborne d-f systems 
for high frequency, very high frequency, and 

a Contract No. OEMsr-1441. 


ultra-high frequency was conducted under 
Project 13-109 and the revised final report 2 
gives the result of this investigation together 
with recommendations for future work. The 
report gives frequency range, present status, 
type of indication, type of antenna, weight, 
wind drag, power consumption, and special 
features for the following direction finders: 
homing equipment; RC-138-T1, AN/ARA-8, 
AN/APD-1, M-3100, and C-1900; manually 
rotatable AN/APA-24; automatic direction 
finders DBH, DBA, CXGJ-2, CXGJ-5, CXHT, 
CXHM, CXGG-2, CXGL-2, M-2300, M-3000, 
and M-4500. These instruments cover the fre¬ 
quency range from 0.25 to 5,000 me. 

Recommendations for Future Work 

At the time the report was written no 
existing equipments covered the lower portion 
of the radio spectrum. Two pending develop¬ 
ments included frequencies just above 2 me, 
the DBA and DBH, but these were designed 
primarily for shipboard installation, and it 
was only the expressed need for airborne auto¬ 
matic direction finders for high frequencies 
which prompted the Naval Research Labora¬ 
tory to suggest that these equipments might 
be satisfactory for aircraft installation. 

Previous experiments in airborne h-f direc¬ 
tion finding, as well as knowledge of the re¬ 
radiation characteristics of aircraft discour¬ 
aged the installation of h-f direction finders 
with 360° bearing indication as airborne equip¬ 
ment. The likelihood that the DBA or DBH 
installed on aircraft will give good accuracy 
was not promising. 

Homing-type direction finders operated 
satisfactorily at any frequency on an aircraft, 
yet this survey reveals no homing direction 
finder existing or under development for fre¬ 
quencies below 18 me. This was probably not 
due to electrical difficulties in the design of 
such equipment, but to the fact that the Ser¬ 
vice found major difficulties in using homing 
direction finders for obtaining bearings on 


Ntia; 




122 




SURVEY OF AIRBORNE DIRECTION FINDERS-PROJECT 13-109 


123 


distant signals. To take a bearing with a 
homing direction finder, the plane’s heading 
had to be varied, and at the time the bearing 
was taken, the plane’s heading and position had 
to be recorded. This process was considered 
unduly confusing for the navigator. When the 
homing direction finder was employed to home 
to within visual sight of a transmitter, this 
difficulty did not arise, and it was for such 
purposes that the homing direction finders in 
this list were probably intended. 

Even in the lower very high frequencies, it 
was expected that the aircraft structure would 
cause serious errors in any direction finder 
which provided indication for 360°. It was for 
this reason that the accuracy of the CXGJ-2, 
CXGJ-5, and CXHM was doubtful in the lower 
portion of their frequency range. 

From 100 me upward, gaps in coverage by 
automatic direction finders appeared to be due 
only to lack of sufficient Service interest in the 
past. Upon completion of developments under 
way, the only frequencies between 100 and 


5,000 me not covered by airborne automatic 
direction finders would be from 160 to 225 me, 
and the manually rotatable C-2100 covered this. 
There was a need for an airborne unit to cover 
100 to 156 me which does not have the large 
wind drag of the CXGG-2. 

Above 5,000 me there appeared to be no 
development problems peculiar to airborne 
equipment, and any system which would oper¬ 
ate on land or on shipboard could be adapted 
for airborne installation. 

It appeared that new approaches were 
worth consideration for developing automatic 
direction finders in the frequency band from 
2 to 30 me, where there was need for equip¬ 
ment which could take bearings on communica¬ 
tions transmitters without the difficulties in¬ 
herent in homing direction finding. 

From 30 to 100 me, there was a similar 
though somewhat less pressing demand for 
automatic direction finders. Provided the 
CXGJ-2 proved satisfactory, research was in¬ 
dicated only in the h-f band. 











































PART II 

APPARATUS DESIGN 


| 
























































































































































Chapter 8 


U-H-F RADIO-SONDE DIRECTION FINDER 


Development of a simple direction finder for observ¬ 
ing the flight of meteorological balloons. 1 Using an 
Adcock antenna and a single-dipole antenna system 
with a corner-type reflector mounted on a tripod, an 
accuracy of %° in determination of azimuth and from 
0° to a few degrees in elevation was attained when 
measuring the direction of balloon transmitters operat¬ 
ing on 183 me. Gold plating the reflector wires im¬ 
proved the shielding of the reflector materially. 

81 OBJECT 

0 meet the need for a simple and dependable 
method for observing the flight of meteoro¬ 
logical balloons under any and all weather con¬ 
ditions, a simple, easily portable radio d-f equip¬ 
ment was developed." 1 The instrument meas¬ 
ures both the azimuthal and the vertical bear¬ 
ings of a small radio transmitter sent aloft on 
balloons, thus avoiding the problems incidental 
to the maintenance and synchronization of two 
ground stations which would be necessary if 
only the azimuthal bearings were observed. 

82 APPARATUS 

The transmitters employed as a source of 
signals for the experiments are the type used in 
radio-sondes. They transmit a vertically polar¬ 
ized wave signal at 183 me. 

The radio direction finder is comprised of an 
Adcock antenna for measuring azimuthal 
angles and a single dipole antenna for measur¬ 
ing vertical angles shielded with a corner-type 
reflector to make the dipole free from the effects 
of reflected waves from the ground. The reflec¬ 
tor system is in fact a secondary radiating sys¬ 
tem which, when placed in an electromagnetic 
field, produces a secondary radiation field such 

a Project C-33, Contract No. OEMsr-217, California 
Institute of Technology. It is understood that a radio¬ 
sonde d-f system developed independently by the Air 
Forces made it unnecessary for the Signal Corps to do 
any more work on the instrument developed under this 
project. 


that, when properly oriented and placed with 
respect to the dipole, it neutralizes the effect of 
the original field at the dipole. 

A simple sketch of the instrument is shown 
in Figure 1. Referring to the figure, the ele¬ 
ments marked 1 and 1', which are self support¬ 
ing rods of duralumin or other suitable ma¬ 
terial are each A/4 long. Rods 1 and 1' are co¬ 
axially supported, with their adjacent ends 
spaced approximately 1 cm apart, by insulating 
supports 2, which are in turn supported by the 
tubular spacer 3 so as to maintain the rods 1 
and 1' in a plane normal to axis X-X' with the 
pairs of rods parallel and spaced A/2. Rods 1 
and 1' have their inner ends connected together 
respectively by the line 4. This assembly is a 
directional antenna of the Adcock type, and is 
used to determine the azimuth of the incoming 
wave in a manner to be described later. 

Rods 5, similar to rods 1 and 1', are similarly 
coaxially supported by insulator 6. Rods 5, 
constituting a dipole antenna, feed the line 7. 
The dipole 5-5 together with the shield 8 con¬ 
stitute the antenna assembly for the determin¬ 
ation of the vertical angle of incidence of the in¬ 
coming wave. 

The shield 8 is of the corner-reflector type 
which shields the dipole from reflected waves 
from ground without impairing the receptive 
and directive characteristics of the dipole in its 
reception of the direct waves from the trans¬ 
mitter. 

The reflector wires 9, approximately 0.6a in 
length are supported so as to be mutually 
parallel, and at the same time parallel with the 
dipole 5-5 in two planes, whose intersection is 
the line D-D'. The included angle between the 
planes ABC is 60°. The dipole 5-5 lies in a plane 
which bisects angle ABC at a focal distance p 
of slightly greater than A/2 from the line D-D' 
and parallel thereto. The reflector wires should 
not be more than a/60 apart, and preferably 
closer. 



127 



128 


U-H-F RADIO-SONDE DIRECTION FINDER 


Y 


i 



























RESULTS 


129 


The two antennas are connected through 
their feed lines 7 and 10 to a d-p, d-t switch, of 
suitable design for the frequencies employed, 
permitting connection at will of either antenna 
to the line 12 which feeds the receiver 13. Line 
10 connects to the electrical midpoint, which is 
also the geometrical midpoint if carefully con¬ 
structed, of line 4. Lines 4, 7, 10, and 12 are 
made with two parallel No. 18 copper wires 
separated by victron spacers. 

The two antenna systems are so mounted as 
to be rigidly held in fixed positions relative to 
each other and constitute the directional an¬ 
tenna assembly. The axes X-X', Y-Y', and Z-Z' 
intersect at angles of 90° each with the other. 
The dipole 5-5 is parallel to axis X-X'. Lines 7, 
10, and 12 have a length of A. 

The directional antenna assembly may be ro¬ 
tated about axes Y-Y' and Z-Z'. One means of 
turning and controlling the rotation of the as¬ 
sembly about axis Z-Z' is illustrated where the 
worm reduction gear 14 is turned by means of 
the hand wheel 15. Rotation about axis Y-Y' 
may be produced manually or by some mechani¬ 
cal device. The angular positions due to rotation 
about axes Y-Y' and Z-Z' may be indicated and 
measured by any suitable system. The simple 
device shown in the drawing consists of a 
graduated quadrant 16 and fixed pointer 17. 
The fixed graduated circle 18 and its associated 
pointer, 19, indicate and measure angular rota¬ 
tion about the Z-Z' and Y-Y' axes respectively. 

The complete unit is shown mounted upon a 
tripod, 20, so that the receiver, 13, is one wave¬ 
length above ground. For best results the axis 
Z-Z' should preferably be more than two wave¬ 
lengths above ground. 

The superheterodyne receiver 13 has an out¬ 
put meter to indicate the signal intensity and a 
pair of earphones for audible indication. It 
must be well shielded to eliminate stray pickup. 
The receiver is supported by a tubular support 
21 , so that it rotates with the antenna assem¬ 
bly as an integral unit with it, and definite 
advantages result since it makes better shield¬ 
ing possible and there is no possibility of the 
characteristics of the transmission lines being 
altered even though the assembly is continu¬ 
ously rotated in the same direction. In this way 
the relative position of the operator with re¬ 


spect to the two antenna systems will remain 
unchanged during operation thus eliminating 
any error in the measurements caused by the 
changing position of the operator with respect 
to the antenna system. 

83 RESULTS 

Various types of reflector systems were 
tested as shields of a dipole antenna from 
ground-reflected waves when used to measure 

o* 



Figure 2. Azimuthal response of X/2 antenna 
with 60° corner reflector spaced A/60. 


the elevation angles of incoming electromagne¬ 
tic waves. A single rod reflector, reflector and 
director combination, cylindrically parabolic 
sheet or wire reflectors, cylindrical sheet or 
wire reflectors, and corner reflectors were ex¬ 
amined. 

The corner reflector in conjunction with a 
simple A/2 dipole was found to be most satis¬ 
factory for the purpose. Figure 2 is a polar 
diagram of the strength of the received signal, 










130 


U-H-F RADIO-SONDE DIRECTION FINDER 


in terms of the i-f voltage on the plate of the 
last i-f stage of the receiver, for various hori¬ 
zontal angular settings of the shield. The circle 


Figure 2. Curve C shows that the shielding is 
quite effective and fairly uniform. 

In Figure 4 are shown response curves indi- 



if) 8 


represents the uniform signal received without 
any shield. Maximum shielding is shown where 
the open side of the shield is 180° from the 
transmitter. In this setup the A/2 antenna and 
the reflector remained vertical, and the reflec¬ 
tor was rotated about the antenna. The curves 
show the results for different focal distances p 
at a spacing of A/60 between reflector wires. 

It was found that a focal distance of 86 cm 
gave the best results with a good ratio of shield¬ 
ing to gain as the reflector was swung through 
180°. Representative curves A, B, and C show 
the response for p’s of 85, 86, and 100 cm 
respectively. 

Figure 3 shows the results of tilting the A/2 
antenna with and without the shield, toward a 
stationary transmitter located on Mt. Wilson 
(about 7 miles away) about a horizontal 
axis. Curve B is the response of the antenna 
alone without any shield. Curves A and C are 
with the shield in place facing toward the 
transmitter, and opposite to it, respectively. 
These positions correspond to the respective 
angles of 0° and 180° in the polar diagram, 


i 

180 ° 



Figure 4. Azimuthal response of dipole with 60° 
corner reflector made of wires of different lengths. 

















RESULTS 


131 


eating the effect of the length of the wire ele¬ 
ments of the reflector at A/7.5 spacing on its 
shielding properties. Because of the congestion 
of the curves near the zero-angle region, only 
representative curves are drawn in the figure. 
It is seen that best shielding occurs at the 
length of 0.65a. This value was used as the 
optimum length of the wire elements in the 
later experiments on the wire spacings of the 
reflector. 

1 


I 8 0° 



Figure 5. Relation between spacing between 
wires in corner reflector and azimuthal response. 

Figures 5 and 6 show the effect of wire spac¬ 
ing on the shielding properties of the reflector. 
It is seen that the closer the spacing of the wire 
elements, the better is the shielding, although 
it is not too critical when the spacing is smaller 
than A/7.5. Nonetheless, A/60 spacing seems to 
be the best in the group. 

These tests were all made close to ground. It 
was later found that the results thus obtained 
do not quite hold when the corner reflector is 
mounted on the instrument at a height of 3 a 
above ground, and in the vicinity of the Ad¬ 


cock antenna and other metal supports of the 
instrument. 

Experiments made with an incoming radio 
wave emitted from a transmitter at Mt. Wilson 
at a vertical angle of 7%° showed that with¬ 
out the reflector the deviation from the true 
direction is over 22°, while with a reflector of 
A/30 spacing the deviation reduces to about 1° 
(see Figure 7). From Figure 7 it is seen that 
for a A/60 spacing the null point is much 
sharper. It is to be remembered that at this 


180 ° 



azimuthal response for spacing values not shown 
on Figure 5. 

grazing angle of 7%°, the intensity of the re¬ 
flected wave from the ground is extremely 
strong. This suggested the necessity of further 
decreasing the spacing and the results shown 
in Figure 8 using A/120 and A/240 spacing are 
quite satisfactory. In both cases the deviation 
is only which is of the same order of mag¬ 
nitude as the experimental error of the instru¬ 
ment. It is also seen from Figures 7 and 8 that 
the small humps which appear in the case of 
larger spacings are smoothed out in the case of 
A/120 and a/240 spacings. 











132 


U-H-F RADIO-SONDE DIRECTION FINDER 



Figure 7. Vertical response characteristics of X/2 dipole with 60° corner reflector with different spacing 
between reflector wires. 


co 

_i 

o 

> 

U_ 

I 


X 

I 

5 

cr 

f— 

CO 


_J 

< 


o 

10 











RESULTS 


133 


Use of Copper Screening as Shield 

For the A/60 spacing there were about 120 
wires which had to be individually fastened into 
proper place with the right spacing, and for a 
A/240 spacing there were 480 wires fastened on 
to the reflector frame. Some difficulties were 
experienced in putting on and changing all 
these wires on a light wooden frame with the 
spacing so close and yet without the wires 
touching each other, when they had to be fast¬ 
ened on to the reflector frame which is about 
20 ft above ground. The A/240 spacing is al¬ 
ready so close that further decreasing of the 


180 ° 



Figure 9. Azimuthal response of antenna with 
bronze screen reflector. Curve A, whole sheet of 
screen; curve B, screen cut crosswise into 4 pieces; 
curve C, screen in 8 pieces; curve D, screen in 12 
pieces; curve E, screen in 16 pieces. 


spacing is impractical in the present method of 
mounting. In view of this fact shielding prop¬ 
erties of fine copper wire screen were experi¬ 
mented. Figure 9 shows the response curves of 
the reflector using various numbers of pieces of 
copper wire screen as the reflecting elements. 


It can be seen from Figure 9 that the more the 
wire screen is cut, the better is the shielding. 
This shows that the presence of the horizontal 
members of the wire screen decreases the effect 
of shielding. 

Different sizes of wires and tubings ranging 
from No. 32 wire to 1,4-inch tubing were tried 
as the reflecting elements and no appreciable 
difference in the efficiency of the shielding prop¬ 
erty of the reflector was observed. 

Determination of Azimuthal and Elevation 
Angles of an Incoming Wave 

Directional measurements were made on in¬ 
coming waves emitted either from a stationary 
transmitter located on top of Mt. Wilson 7 miles 
away, or from a transmitter sent aloft on 
meteorological balloons. 

Table 1 shows the results of azimuthal meas¬ 
urements with the Adcock antenna made in an 


Table 1. Azimuthal angles measured by Adcock 
antenna. 


Observations made by 
Adcock antenna, 
in degrees 

Visual 
observation, 
in degrees 

Vertical angle (at time 
azimuthal observation 
was made), in degrees 

88 

88 

34 

87f 

88 

321 

88 

88 

29 

88 

88 

20 

7 

71 

45 

9 

91 

42 

i 

4 

I 

45 

29 

29.9 

251 

7 (Mt. Wilson) 

7.3 

81 

1 (Mt. Wilson) 

0 

7 4 

0 (Mt. Wilson) 

0 

71 


open field with the transmitter supported by a 
captive balloon. This illustrates the indepen¬ 
dence of the azimuthal measurements from the 
vertical incidence of the incoming wave. The 
last three low-angle measurements were made 
at different times, at different locations, on the 
Mt. Wilson transmitter. The accuracy obtained 
in the azimuthal measurements is within 14 0 . 

Repeated measurements made on the eleva¬ 
tion angles of the direction of the incoming 
wave emitted from a transmitter on top of Mt. 
Wilson are within %° from the true direction 













134 


U-H-F RADIO-SONDE DIRECTION FINDER 


(whose true elevation angle is 71 / 2 °). Observa¬ 
tions were also made on transmitters sent aloft 
on captive balloons at various altitudes and ele¬ 
vation angles. The results of these observations 
made at various times are shown in Table 2. 


Table 2. Measurements of elevation angles by a 
dipole antenna with corner reflector. 


Determined by dipole antenna, 
in degrees 

Determined visually, 
in degrees 

32 

32 

28 

28 

24 

24! 

58 

58 

50 

50 

20! 

20 

32 

32 

49 

49 

58 

57! 

65| 

64! 

66! 

66 

71 

69 

68 

68! 

67! 

68! 

77! 

77 

51 

50! 

35! 

35 

35 

34! 

33! 

33f 

25! 

25! 

26! 

29! 

30! 

31! 

28 

29 

28 

27! 

281 

25f 

30! 

31! 

31 

28! 

30! 

29 

29! 

29! 

29! 

28! 

31 

34 

31 

33! 

32! 

33 

32 

36! 

32! 

38! 

32! 

37! 

40 

34! 

40f 

36! 

41! 

39! 

41! 

39! 

41f 

36! 

41! 

40! 

41! 

39! 

40 

38! 

39! 

38 

40! 

35 

43 

42! 

43! 

43! 

43! 

44! 

43! 

43! 

42! 

39! 

43! 

43! 

42! 

40! 


Table 3 shows the measurements made on the 
elevation angles when the reflector was re¬ 
moved. This demonstrates the great deviation 


Table 3. Measurements of elevation angles with¬ 
out reflector. 


Determined by dipole antenna, 
in degrees 

Determined visually, 
in degrees 

3 

8 

12 

16 

7 

14 

19 

14 - 

25 

31 

28 

17 

30 

29 

40 

32 

60 

30 

68 

34 

90 

76 

98 

72 


in the readings from the true direction caused 
by ground-reflected waves. 

While some of the results of the elevation 
angles obtained by using the direction finder 
are very good and agree within %° with the 
readings obtained visually, there are readings 
which differ quite appreciably from those meas¬ 
ured visually. This is probably due to the fact 
that the antenna swings badly, changing the 
plane of polarization of the incoming waves. 
This in turn affects the magnitude of the emf 
induced in the receiving antenna and causes the 
fluctuation in the output of the receiver. When 
the indicator needle swings badly, it is hard to 
determine the null point with any accuracy. 

Effect of Shield Oxidation 

Another factor responsible for the deviations 
in the readings which sometimes amounted to 
as much as a few degrees is probably the de¬ 
crease in the efficiency of the shielding system 
on account of the formation of a poorly con¬ 
ducting layer on the surface of the copper wire 
elements. This layer is due to oxidation, caused 
by the constant exposure of the shielding sys¬ 
tem to various weather conditions. Since at 
ultra-high frequencies, practically all the cur¬ 
rent flowing in the wire is concentrated on the 
surface of the wire, any contamination of the 
surface will decrease the efficiency of the wire 
elements in their secondary radiation. This is 











RESULTS 


135 


especially important in the present case be¬ 
cause the secondary radiation field due to these 
wires serves to neutralize the effect of the 
ground-reflected waves at the antenna. Any 
deterioration in the efficiency of the shielding 
system would decrease the intensity of the sec¬ 
ondary radiation field thus causing the effect 
of the ground-reflected waves still to be mark¬ 
edly noticeable at the antenna. 


neers using fixed and movable target transmit¬ 
ters which were either set up on top of one of 
the laboratory buildings or carried around by 
an Army jeep on the field. The results obtained 
were quite satisfactory. Later a free balloon 
flight with a buzzer-modulated transmitter was 
made and the flight lasted about half an hour 
with a range of approximately 25 miles. The 
results obtained show that an error in both the 



Figure 10. Effects of gold plating the reflector. 


A new corner-type reflector was made of No. 
30 copper wire which was gold plated. Tests 
made using the new reflector on an incoming 
wave from Mt. Wilson show a marked improve¬ 
ment over the old reflector whose elements had 
been badly oxidized. This is shown in Figure 10. 

Experiments Made at Fort Monmouth 

The direction finder was shipped to Fort 
Monmouth September 18, 1942, where it was 
set up on the ground in front of one of the 
laboratory buildings of the Signal Corps Field 
Laboratory No. 2 at Eatontown, N. J. Exten¬ 
sive tests were made by the Signal Corps engi- 


azimuthal and elevation angles ranged from 
y 2 ° to 31/2°. 

During one of the flights, the theodolite ob¬ 
server lost the balloon in a heavy cloud bank, 
but 25 minutes later the balloon was relocated 
in the theodolite with the help of the settings 
obtained by the direction finder. 

It is believed that by using proper damping 
devices to keep the antenna from swinging ap¬ 
preciably during the flight, by properly shield¬ 
ing the transmitter and by further increasing 
the efficiency of the reflector system such as by 
decreasing the reflector spacing, etc., the ac¬ 
curacy in the determination of the elevation 
angles can be greatly increased. 







Chapter 9 


DEMOUNTABLE SHORT-WAVE DIRECTION FINDER 


Development of equipment (SCR-502) giving instan¬ 
taneous bearings on signals in the region 1.5 to 30 me, 
easily transportable in an Army trailer, capable of 
being erected in a few hours, with wave collectors as 
free from polarization errors as possible, the cathode- 
ray oscilloscope patterns which show bearing also 
giving indication of the quality of the bearings and 
the condition of operation. As a single-band system 
(SCR-291) this apparatus was widely used by the Air 
Transport Service. 1 

9.1 INTRODUCTION 

At the time this project 21 was started, the 
±\_ principal short-wave direction finder in 
use was the elevated H Adcock system. It was 
manually rotated by the operator and employed 
aural null indication. A typical device of this 
type was the SCR-551-T1. Some preliminary 
work had been done on a fixed land-station 
direction finder for these frequencies, desig¬ 
nated the DAJ, and made for the Navy. No 
portable short-wave direction finder was avail¬ 
able which would give reasonably accurate 
bearings under conditions of sky-wave recep¬ 
tion, principally because of errors caused by 
horizontally polarized components of the re¬ 
ceived signal. In a large number of cases it was 
impossible to take bearings with existing sys¬ 
tems because of the inability of the operator to 
follow the null mechanically. Tests on an ele¬ 
vated H Adcock showed that its performance 
was greatly affected by ground conditions and 
that the order of balance required to secure pro¬ 
tection against horizontally polarized waves 
was beyond all practical limits. 

The DAJ equipment showed that it was pos¬ 
sible to secure materially improved results by 
using cathode followers at the antennas and by 
burying the cables to reduce the effects of cur¬ 
rent in them to a small degree. 


“Project C-34, Contract No. OEMsr-263, Federal 
Telephone and Radio Corporation. 


92 ACCOMPLISHMENTS 

The SCR-502, a two-band system covering 2 
to 30 me and which was produced in quantity 
as a single-band system for 2 to 10 me (SCR- 
291), gave bearing accuracy of 2 per cent on 
perhaps 75 per cent of the received signals 
when the apparatus was properly set up, the 
octantal error corrections made and the opera¬ 
tor trained to its use. 

Compared to existing systems, the SCR-502 
had a great improvement in accuracy, gave the 
operator the ability to see the nature of the 
signal and to interpret its probable worth as an 
indication of bearing, and was reasonably port¬ 
able since it could be set up in about 2 hours 
on a suitable level site without the necessity of 
burying any cables or indulging in airplane 
calibration procedures. It was comparatively 
easy of maintenance because of the cathode fol¬ 
lowers separating the antennas from the con¬ 
necting cables to the apparatus. 

The principal characteristics and advan¬ 
tages of the system are: 

1. Bearings can be taken on very short sig¬ 
nals. For instance, a good bearing can be 
taken by an ordinary operator in 2 seconds 
maximum (including sense finding). 

2. The reliability of the bearing is known. 
The indicator shows whether the signal is 
mixed with interference or with other signals. 
It shows if the bearing is steady and reliable, 
or shifting due to propagation conditions. 

3. The signal is audible during the taking of 
bearings. 

4. The accuracy is independent of the fre¬ 
quency within the limits allowed by the quality 
of the antenna and goniometer designs. 

5. Only one receiver is used in a conven¬ 
tional manner. The receiver differs from a 
standard type in that the input circuit is de¬ 
signed to match the balanced output of the 
goniometer and the output of the rectified i-f 


136 



PRINCIPLE OF OPERATION 


137 


circuit is connected to the external oscilloscope 
amplifier. 

6 . Remote indication is possible. An oscillo¬ 
scope indicator installed at a distance will indi¬ 
cate the same image as the original. The trans¬ 
mission can be effected on two wire line pairs. 

7. Sensitivity of the direction finder for a 
selectivity of 3 kc and a signal-to-noise ratio of 

8 to 1 varies between 6 and 12 /xv per meter 
within the frequency range covered. Under 
these conditions, a good readable pattern is ob¬ 
served on the indicator so that an inexperienced 
operator can take a bearing accurate to within 
2 per cent and an experienced operator can take 
a bearing accurate to within 1 per cent. Cor¬ 
rection curves of the system installed at a 
proper site are not larger than ±4 degrees. 

The sensitivity on sense is the same as on 
direction finding; on downcoming waves with 
elevation angles up to 30°, the sensitivity is 
practically as indicated above; half the above 
figures hold for angles of 45° and one-quarter 
the values for angles of 60°. 

8 . Polarization errors as measured by means 
of a transmitter on a 90-ft tower are no larger 
than those of the best fixed antennas that 
could be installed. The monopole antennas are 
25 to 50 times less sensitive to horizontally 
polarized waves than to vertically polarized 
signals. Standard wave errors are of the order 
of 1.5 to 3° maximum. 

9. Installation of the complete system plus 
adjustments requires 47 man hours; a trained 
crew of five men can place the direction finder 
in operation in two to three hours. 

9 3 DESCRIPTION OF THE EQUIPMENT 

The direction finder comprises: 

1. Two sets of five monopole vertical an¬ 
tennas, only one set to be utilized at a time with 
the corresponding set of transmission lines. 
One set covers the range 2 to 10 me; the high- 
band collectors covering the range 10 to 30 me. 
The two wave collectors differ in the spacing of 
the receiving elements, and therefore, in the 
lengths of transmission lines and the dimen¬ 
sions of the ground mats. 

Each set of wave collectors comprises four 
fixed monopole antennas used in conjunction 
with a crossed-coil goniometer plus a fifth re¬ 


ceiving element identical to the four others of 
the group, installed in the center of this group. 
This fifth element is used for sense finding. The 
two wave collectors can be installed in several 
ways providing a minimum distance of about 
60 yards exists between the two. 

2 . Two remote motor-driven goniometers in¬ 
stalled in the field near the antennas and de¬ 
signed to operate in the wave range of each 
antenna. 

3. A receiver covering the entire range in 
several bands. 

4. A CRO indicator. 

5. A power-supply system normally designed 
to be operated from 110 volts alternating cur¬ 
rent ; if this current is not available the equip¬ 
ment can be operated from a storage battery 
feeding a rotary converter, although the bat¬ 
tery drain is high. 

6 . A remote indicator, all electronic, repro¬ 
ducing at a distance of 7.5 miles the pattern 
obtained on the local indicator by means of two 
W110-B type field wire lines. 

The receiver, indicator, and power-supply 
circuits are installed in a trailer. This trailer is 
arranged to carry all parts of the antennas, 
reels of cable, ground mats, and electronic 
equipment. 

94 PRINCIPLE OF OPERATION 

The signals received by a wave collector made 
up of five receiving elements are converted to 
balanced outputs by cathode followers, mixed 
and scanned with a goniometer, the search coil 
of which is rotated by an electric motor at a 
constant speed of 30 rps. The signals are then 
amplified and rectified in the receiver. 

The output circuit is arranged so that, in the 
absence of signal, a continuous current is ob¬ 
tained. The presence of the signal carrier re¬ 
duces the current, which approaches zero for 
signals of sufficient amplitude. The output cur¬ 
rent is applied to a deflection coil system which 
rotates about the neck of the oscilloscope syn¬ 
chronously with the search coil of the goniome¬ 
ter. 

In previous designs the goniometer was in¬ 
stalled on the same shaft as the rotating coil, 
thus avoiding the synchronization problem. In 
this design, the goniometer is remotely rotated 



138 


DEMOUNTABLE SHORT-WAVE DIRECTION FINDER 


by a synchronous motor and the synchronism is 
automatically controlled. 

In the absence of signal, the deflected spot 
traces a circle on the screen. In the presence of 
signal, the deflection is decreased and the spot 
tends to come back to the center of the screen. 
Because of the synchronization with the search 
coil, a fixed image like a double arrow, which 
can be sharpened or flattened depending on the 
amplitude of the carrier current at the detector, 
is seen on the CRO screen. 

9 41 Sense Circuit 

Sense indication is obtained as follows: the 
central antenna is connected to the output 
transformer of the goniometer through a high- 
frequency line. The resulting cardioid diagram 
or one intermediate between cardioid and figure 
eight causes an image as shown in Figure 1 to 
appear on the screen. To make the sense read- 



Figure 1. Sense and azimuth patterns superim¬ 
posed on CRO indicator. In practice, azimuth pat¬ 
tern is suppressed when taking sense bearing. 


ing easy, the position of the figure on the screen 
is rotated 90° by connecting the output of the 
CRO amplifier to another set of deflecting coils 
displaced 90° with respect to the normal set. 
This operation is performed by a relay at the 
same time as the emf from the central receiv¬ 


ing element is applied to obtain a unidirec¬ 
tional diagram. 

The CRO indicator used locally (type AS) is 
highly accurate. The remote indicator (type E) 
is less accurate but performs satisfactorily the 
operation desired. 

9,42 Operation of the Remote Indicator 

At the main station a pulse is generated 
synchronously with the rotation of the type AS 
indicator. This pulse, sent over one of the tele¬ 
phone lines to the remote indicator, synchro¬ 
nizes an oscillator which is used to produce a 
circle on the CRO screen. The diameter of the 
circle is controlled by the coupling tube bias, 
which is modulated by the output current of 
the receiver sent on the second telephone line. 
A pattern similar to that observed on the type 
AS indicator is produced on the remote indi¬ 
cator. 

A voice telephone circuit can be connected to 
the first line. 

The remote indicator is locally supplied with 
power not required to be exactly of the same 
frequency as at the main station. 


943 Wave Collectors 

Quite a number of antenna systems were 
studied before the one used in this system was 
selected. Vertical- and horizontal-spaced loops 
were compared with the well-known vertical¬ 
spaced antenna system and it was found that, 
for efficiency and sensitivity, the monopole an¬ 
tennas were much better. 

However, when using conventional monopole 
antennas, with or without solid ground mat, 
and with direct shielded crossed connection be¬ 
tween each pair of antennas (buried or not), 
the results obtained with respect to polariza¬ 
tion errors were rather poor. 

A thorough study of the operation of the 
monopoles showed that the direct connection 
between antenna bases through a solid ground 
mat of small dimensions, or through the shield¬ 
ing of the cross-connecting line was responsible 
for a very large part of the polarization errors 
observed. 

Therefore, a new system of connection was 
established, in which the lines approach each 





PRINCIPLE OF OPERATION 


139 


pair of monopoles at an angle of 90° from the 
position of the cross-connecting lines employed 
in the old designs. 

If, now, the two parallel lines leaving one 
antenna pair are prolonged to infinity, no 
parasitic induction will take place at the null 
point for any polarization of the sky wave. 

However, a practical length of these lines 
before cross connection has been found to be 
twice the spacing between monopoles. There¬ 
fore, the lines and cross connections have the 
shape of a U lying on the ground. 

Study also showed that: 

1 . Independent ground mats of a radius of 
about 20 to 30 per cent of the height of the 
monopole are the most satisfactory. The use of 
a solid ground mat of small dimensions im¬ 
mediately jeopardizes the quality obtained with 
the U connections. 

2. The connection of the monopoles as ex¬ 
plained above, with cables lying on the ground, 
gives much better results than direct cross con¬ 
nections even though the latter are buried 12 ft 
in the ground. 

The monopoles are coupled to the high-fre¬ 
quency lines through coupling units made of a 
cathode-follower phase-inverter circuit. In this 
manner: 

1. Only one tube is used to transform the un¬ 
balanced input into a balanced output. 

2 . The circuit efficiently transfers energy 
from the antenna to the low-impedance line, for 
any value of the antenna impedance through¬ 
out the rather large band required. 

3. The same coupling unit is used for fre¬ 
quencies from 1 to 30 me. 

4. There is a loss in the voltage transfer 
from antenna to line of about 50 per cent, but 
a gain of power transfer of over 20 db. This is 
in excess of an ideal transformer. 

5. Tube noise is negligible due to the de¬ 
generation present. In this respect the circuit 
works like a transformer, the fluctuation noise 
still being originated in the first circuit and 
tube of the receiver. 

6 . Any length of high-frequency line can be 
used due to proper impedance matching at the 
transmitting end. 

7. Due to the amount of degeneration and the 
absence of voltage gain, the transfer is quite 
stable and the d-f operation can be performed 


in spite of supply voltage variations as great as 
15 per cent. 

As a whole, this new means of coupling the 
antenna to the lines, jointly with the new sys¬ 
tem of U cross connection between antennas, 
represents a complete redesigning of the old 
Adcock antenna. 

The use of a remotely rotated goniometer 
requires the use of only one high-frequency line 
between the goniometer and the trailer con¬ 
taining the equipment, with consequent reduc¬ 
tion in the problem of balancing the electrical 
lengths of the cables. 


9 4 4 Sense Circuit 

Determination of sense has always been a 
delicate feature of the old Adcock systems, espe¬ 
cially when the frequency band was large. Am¬ 
plitude and phase adjustments were needed, 
and the results were doubtful and required too 
much time. A new sense circuit has been de¬ 
veloped which does not require an extra ampli¬ 
fier, tuned circuit, or phase and amplitude ad¬ 
justment. In this new circuit the sense mono¬ 
pole antenna is connected to cathode followers 
each of which is connected through a trans¬ 
mission line to a common dummy goniometer. 
The secondary of this goniometer is coupled to 
the secondary of the normal goniometer 
through a sense relay. The dummy goniometer 
is electrically equivalent to the rotating 
goniometer. Its purpose is to introduce within 
the frequency band a phase shift exactly equal 
to the phase shift introduced by the normal 
goniometer. The two transmission lines are of 
different lengths; the difference being elec¬ 
trically just equal to the spacing between the 
two monopoles of one directive antenna pair. 

The result is that the phase of the sense an¬ 
tenna signal is shifted by exactly 90° through 
practically all the frequency band that the di¬ 
rective monopoles can cover. Moreover, the 
amplitude of the signal from the sense antenna 
is automatically equal to the amplitude of the 
directive signal for all directions and fre¬ 
quencies without adjustment. This sense circuit 
avoids any sense-circuit modification of a stand¬ 
ard commercial receiver in its use in the d-f 
system. 



140 


DEMOUNTABLE SHORT-WAVE DIRECTION FINDER 



Figure 2. Field layout of demountable d-f set SCR-502. 


Stand-by reception is provided by cutting off 
the plate voltage of the coupling units of the 
four directive monopoles, leaving the sense 
monopole operating alone. 


The cable layout and the area covered by a 
two-wave d-f system of this type are shown in 
Figure 2. Other arrangements of the two an¬ 
tenna systems may be employed provided that 


STRAP 


^ v V ^ ^ V 1 rr 



-. £ 


COPPER MESH 


C7 

SECTION A-A 


n> 


§> 


o 



a 


<ti 



5’-0"- >\ IO>- 


6'-IO 


II 


io'-io- 


HIGH FREQUENCY MAT LOW FREQUENCY MAT 

Figure 3. Dimensions of ground mats used with two sets of antennas. 


The goniometers are of low impedance and 
without shielding between primary and sec¬ 
ondary. They are connected to the receiver 
without slip rings, through a rotating coupling 
transformer. 


a minimum distance of 60 yards exists between 
them. To make the antenna installation easy, 
a small compass and transit on a tripod are 
furnished, together with chains of fixed length 
to determine the spacing between monopoles. 












































































PRINCIPLE OF OPERATION 


141 


A relay in the coupling unit shorts the an¬ 
tenna to ground in absence of plate current so 
that any parasitic reception coming from this 
antenna is avoided, thus permitting a quick 
check-up of the individual antennas, high-fre¬ 
quency cables, etc. 

The ground mats (Figure 3) are made of 
flexible copper screening material. The dimen¬ 
sions are fairly critical. 

The high-frequency lines are flexible solid 
dielectric in type, balanced, shielded and 
vinylite covered, made up of two coaxial cables 
of about 60 ohms impedance. They can be 
operated in all conditions of humidity or under 
water. 

9 4,5 Goniometer Drive Units 

The goniometers (two of which are required 
for a two-wave collector system) are remote 
from the antennas and are contained in a 
goniometer drive unit which also comprises a 
synchronous motor with synchronization con¬ 
tacts and a junction box for connecting the ar¬ 
ray to the operating equipment. The relay for 
connecting the sense antenna line to the pri¬ 
mary of the goniometer output transformer is 
also placed in the goniometer drive unit. 

9 ' 4 * 6 Synchronization System 

The synchronous motor rotating the gonio¬ 
meter can take four different positions with 
respect to the synchronous motor rotating the 



Figure 4. Functional diagram of synchronizing 
system to keep indicator and goniometer motors in 
synchronism. 

indicator coils around the CRO tube in the 
trailer. To avoid a possible ambiguity of 90° 
an automatic synchronization scheme was de¬ 
veloped (see Figure 4). Since the stability of 
position of the two motors when they are run¬ 


ning is better than 1°, the purpose of the syn¬ 
chronizing circuit is only to place the two 
rotors in the same position without any am¬ 
biguity. 

To provide for this result, rotating contacts 
have been placed on the indicator shaft and on 
the goniometer motor shaft. The contactors 
are wired in series with one end grounded. The 
other end of this circuit is connected to the 
input grid of a two-tube amplifier. This grid 
is ordinarily biased to cutoff. The contacts are 
so adjusted that the indicator contactor is 
closed for about 270° of the rotation and the 
goniometer contacts close for about 30° of 
rotation while the indicator contacts are open. 
In normal synchronized operation the two con¬ 
tactors never close at the same time and thus 
the input grid of the amplifier is biased to cut¬ 
off. Under this condition, grid and cathode 
of the second of the amplifier tubes are at the 
same potential and current flows through this 
tube. Plate current flowing through a resistor 
produces a voltage drop which is applied to a 
thyratron as a negative bias preventing this 
tube from firing. 

If, however, the two motors are out of syn¬ 
chronism, there will be a period during the 
rotation cycle in which the two contactors close 
together, grounding the grid of the first ampli¬ 
fier permitting it to draw current. Each time 
the contactors close, a pulse of voltage appears 
across the plate load resistor and is applied to 
an RC circuit. After several pulses the poten¬ 
tial across the capacitor of the RC circuit 
reaches the flashing voltage of a neon lamp. 
Current from the lamp through a resistor puts 
a negative voltage on the grid of the second 
amplifier tube, cutting off its plate current and 
removing the cutoff bias from the thyratron. 
When the thyratron fires, it opens the power 
circuit to the goniometer drive motor allowing 
the motor to slip one pole but stay at syn¬ 
chronous speed. A potentiometer controls the 
time during which the motor power is inter¬ 
rupted. 

These synchronizing cycles will continue 
until the motors pull into step, usually a matter 
of from 1 to 3 seconds. 

The synchronizing unit also contains a multi¬ 
pole switch which selects the proper circuits for 




























142 


DEMOUNTABLE SHORT-WAVE DIRECTION FINDER 


operation from either the low- or high-fre¬ 
quency array. It switches phase inverter 
power, goniometer drive, and sense relay cir¬ 
cuits, contactor synchronizing pulses and re¬ 
ceiver r-f input. 

9 - 4 - 7 Local Indicator 

The local indicator consists of the following 
components. 

1 . A synchronous 1,800-rpm motor operating 
from the 60-cycle 110-volt supply. The motor 
shaft is provided with the synchronizing contact 
operation described. 

2. A set of magnetic deflection coils mounted 
in a rotating housing which is also driven by 
the motor. Provision is made for adjusting the 
instantaneous angular position of the deflection 
coils with respect to the motor armature. 

3. A 5-in. CRO tube of the electrostatic de¬ 
flection type which is positioned inside the 
rotating deflection coils and their housing and 
whose beam is therefore deflected by the mag¬ 
netic deflection coils. 

4. An optical system which consists of an 
illuminated scale and a mirror so positioned 
that the reflection of the scale appears to coin¬ 
cide with the pattern obtained on the cathode- 
ray oscilloscope. 

5. A control box containing circuits and con¬ 
trols for positioning the image on the cathode- 
ray tube screen and obtaining good focus and 
correct intensity for easy operation. 

6 . Housings and brackets for maintaining 
the mechanical, optical, and electrical parts of 
the indicator in correct alignment for accurate 
operation. 

948 Remote Indicator 

The remote indicator, b an entirely electron¬ 
ically operated unit, displays the same pattern 
as the local (AS) indicator, and is used in 
conjunction with the same receiver and goni¬ 
ometer, but does not use any moving parts. It 
is therefore particularly well adapted for use 
as a remote indicator. 

The circular trace of the CRO spot is ob¬ 
tained by applying to the deflection plates two 


b Not supplied with SCR-502. 


sinusoidal voltages in phase quadrature gener¬ 
ated by a local 30-cycle oscillator the phase of 
which is synchronized with the goniometer 
rotation by means of synchronizing pulses. 
These synchronizing pulses are taken from a 
rotating contact on the goniometer shaft. 

When a signal is being received, the CRO 
spot is deviated toward the center of the screen 
by a reduction in amplitude of the 30-cycle 
voltage. This reduction is accomplished by 
plate modulation of the tubes which amplify 
the 30-cycle voltage. The current which causes 
the plate modulation of the amplifying tubes is 
obtained from the rectified signal voltage in the 
receiver. 

The phase shift of the indicator pattern for 
purpose of sense determination is obtained by 
a 90° phase shift of the synchronizing pulse 
from the goniometer shaft. 

When the remote indicator is used at a dis¬ 
tance from the radio receiver, the rectified 
signal can be transmitted over a standard tele¬ 
phone line into a d-c amplifier of sufficient gain 
to compensate the attenuation in the line. 

In the remote indicator assembly are the 
following circuits: 

1. Synchronizing-time phase shifter. This 
shifts the phase of the 30-cycle voltage to rotate 
the position of the arrow on the cathode-ray 
screen. 

2. Oscillator. This is the initial source of 
the emf for the circular trace. A modified 
transitron oscillator is employed with output 
taken from the plate circuit. This form of 
oscillator is particularly adapted to pulse syn¬ 
chronization over a fairly wide frequency band 
with the particular virtue that changes in 
speed of the goniometer will not cause loss of 
synchronism. 

3. Smoothing amplifiers. Since the oscillator 
wave form is not sinusoidal, a conventional 
resistance-capacitance low-pass filter in con¬ 
junction with an amplifier is necessary to ob¬ 
tain a pure sine wave of sufficient amplitude. 

4. Phase splitter. To obtain the circular 
trace, two 30-cycle voltages in phase quadrature 
must be available. Accordingly, a phase splitter 
with semivariable control is incorporated in 
the circuit design. 

5. Dual push-pull modulated amplifiers. To 
avoid trapezoidal distortion on the cathode-ray 




CHOICE OF THE SITE FOR DIRECTION FINDING 


143 


tube it is necessary to operate with balanced 
input to the deflection plates. To avoid tran¬ 
sients after modulation over a frequency range 
of 0 to 10 kc, a push-pull d-c amplifier is em¬ 
ployed. 

6. Modulator. Two tubes in parallel, a low- 
mu triode and a high-mu pentode, are used to 
give an arrow pattern which is sharply pointed 
at the circumference of the circle. 

7. Second anode modulator. Because the de¬ 
flection plates change in potential about 450 
volts, and the focus of the cathode-ray tube is 
dependent upon the difference in the voltage 
between second anode and deflection plates, the 
second anode voltage is modulated propor¬ 
tionately with the deflection plate voltage. 

8. Audio amplifier. A conventional single- 
stage amplifier is included so that when the in¬ 
dicator is used as a remote indicator, the 
operator may hear the radio signal the bearing 
of which is being indicated. 

9. Two conventional power supples are used. 
Voltage regulation is used in the power supply 
for the vacuum tube plates but no regulation is 
employed on the CRO power supply. 

Accuracy of the type E indicator is depen¬ 
dent upon purity of wave form in the 30-cycle 
circuits, freedom from distortion in the modu¬ 
lation and amplifying circuits, and balance in 
the push-pull circuits. An accuracy of ±3.5° 
was obtained in the laboratory model. 

9 5 CHOICE OF THE SITE FOR 

DIRECTION FINDING 

Considerable experience was gained during 
this and other d-f projects on the matter of 
proper choice of sites for d-f operations. The 
gist of this experience follows. 

The choice of the site is not necessarily the 
point which would be closest to the transmit¬ 
ter. In fact, long-distance bearings are more 
accurate than bearings taken at the skip dis¬ 
tance. The following simple rules will be found 
useful in picking a d-f site: 

1. Soil. 

a. Must be flat (not more than 1 ft change 
in elevation in 50 yd). 

b. Must be as homogeneous as possible, 
capable of supporting grass or plants 
most of the year, and as wet as possible. 


Outcroppings of rocks, or large boulders 
at less than 6 ft from the surface, are 
undesirable. 

c. A flat prairie, or a cultivated field or 
pasture is perfectly suitable. 

d. Rocky seashores, rock islands, or rocky 
hills are unsuitable. 

e. A flat area behind a beach is suitable if 
the beach ridge meets the following 
specifications. 

2 . Obstacles around the direction finder. 

a. From the antenna site the angle between 
the top of any obstacles and horizon 
must not exceed 2°. (Obstacles are 
mountains, hills, trees, houses which 
mask the view of the horizon.) 

b. An obstacle of 2° cannot be tolerated 
closer than 200 yd for many antennas. 

c. Long-distance obstacles like mountains 
can be tolerated if they subtend 5° or 
less at 5 miles, but it must not be forgot¬ 
ten that they will act as a perfect 
screen for direct-ray short-wave recep¬ 
tion from a transmitter located on the 
other side of the obstacle. 

d. These obstacles may also affect the in¬ 
tensity and direction of low-angle (10°) 
sky waves coming from long-distance 
transmitters. 

e. Such obstacles will not generally affect 
the reception of sky waves making an 
angle of 25° or more with respect to 
ground and coming from medium-dis¬ 
tance transmitters. 

3. Power and telephone lines. All incoming 
lines should be laid on the ground and leave 
the trailer in such a way that they are as re¬ 
mote as possible from the wave collectors. No 
high-voltage power line can be tolerated at less 
than half a mile. Such power lines supported 
by tall steel towers should be at least 1 mile 
away. The same distance applies to a railroad 
or a trolley line. On one side of the installation 
one telephone and/or one low-voltage power 
line can be tolerated at distances greater than 
300 yd from the nearest antenna. 

4. Trees. Tall trees (40 to 50 ft) or forest 
growth must be farther than 300 yd. Small 
groups of trees not exceeding 20 ft in height 
can be tolerated if farther than 30 yd from the 
nearest antenna. 



144 


DEMOUNTABLE SHORT-WAVE DIRECTION FINDER 


*6 CALIBRATION 

In calibrating the direction finder it is im¬ 
portant to operate the target transmitter at a 
sufficient distance from the collector antenna so 
the effects of supply and h-f cables are com¬ 
pletely avoided. If the calibration is made at 
too short a distance from the antenna array, 
errors are observed which are much larger and 
do not correspond at all to the errors observed 
with waves coming from distant transmitters. 

Figure 5 shows a calibration made on 8 me, 
indicating that when the target transmitter is 


on the h-f array with the target transmitter on 
a 75-ft radius. The maximum error observed 
is 5° in the north direction and is still due to 
effects of obstacles around the direction finder. 

The Great River (Long Island) experimen¬ 
tal field where the calibrations were made is far 
from ideal for such studies with its many an¬ 
tennas and underground cables. 



DISTANCE FROM ARRAY IN FEET 


Figure 5. Calibration curve of low-frequency 
array on 8 me. 

near or above the cables, large errors are ex¬ 
perienced. In making this calibration the tar¬ 
get transmitter was 200 ft from the center of 
the array at each target position. 

To investigate the deviations as a function 
of the distance of target transmitter, the latter 
was moved in the direction shown in Figure 
6 . The diagram on the same drawing indicates 
for 8 and 3.5 me how the error varied between 
75 ft and 500 ft from the array. 

Figure 7 shows a calibration made at 20 me 



Figure 6. Variation of deviation between 75 and 
500 ft; 3.5 and 8 me. 


These calibration tests show that, first, 
calibrations at relatively short distances, where 
cable effects are magnified by the lack of homo¬ 
geneity of the field of the target transmitter, 
have to be completely discarded; secondly, 
calibrations made at a reasonable distance in¬ 
dicate a very fair accuracy of the direction 
finder (small instrumental error) but no con- 


























INTERPRETATION OF PATTERNS 


145 


stancy in the results because these calibrations 
are not yet free from effects of obstacles. 

Bearings taken on distant stations, 10 miles 
or farther, show much better accuracy than the 
best accuracy noted on the calibration curves 
shown. 



Figure 7. Calibration of high-frequency array 
on 20 me. 

About 50 per cent of the bearings taken 
under any conditions were within 1° and 2° 
accuracy and only a very small percentage of 
bearings, less than 2 or 3 per cent, showed 
errors of more than 10°. These errors were 
attributed to propagation irregularities or par¬ 
ticularly bad interference effects. 

** INTERPRETATION OF PATTERNS 

The patterns on the CRO can be read easily 
by an inexperienced operator. With some train¬ 
ing or skill, however, an operator can obtain 
much information through the interpretation 
of the patterns which are superimposed upon a 
fixed scale graduated in degrees from 0 to 360. 
The equipment is installed and adjusted so 


that the 0° or 360° point corresponds to a 
bearing of true north. 

In operation, a station is tuned in on the 
receiver and the gain is adjusted so that a read¬ 
able pattern is secured. The preferable pattern 
is that composed of a double arrow (Figure 
8 A), its points resting on the scale and its 
central point in the exact center of the scale. 
The width of the pattern is not important 
except as this determines the sharpness of the 
points. Such a signal is usually obtained from 
a local transmitter having an unkeyed and un¬ 
modulated carrier and where reception is well 
above the noise level of the receiver. The bear¬ 
ing of such a transmitter is easily determined 
by reading the scale so as to determine the 
exact point to which the arrow points. Through 
operation of the sense key, it is possible to de¬ 
termine which of the arrows to read. If the 
signal is modulated, the point of the arrow is 
slightly broadened, but the bearing can be read 
to the same degree of accuracy. The same is 
not true for a signal which is so weak that it 
is mixed with receiver noise. The operator is 
able to determine whether or not the bearing 
is reliable by observation of the pattern partic¬ 
ularly in regard to the following points: 

1. Are the arrows sharply pointed? If not, it 
is possible that the gain of the receiver is not 
properly adjusted or that the propagation char¬ 
acteristics between the transmitting and receiv¬ 
ing aerials are for that moment unfavorable 
to the determination of the bearing. 

2 . Are the points of the arrows fixed? If the 
CRO pattern is shifting, the propagation char¬ 
acteristics are changing with time. This may 
be due to changing characteristics in the upper 
atmosphere, or it may be due to reflections 
from other sources or absorption in the path 
of the transmission. If the pattern is changing, 
it will usually be found that not only have the 
positions of the indicated bearings changed, but 
also the shape of the arrows is changed. It 
usually happens that the arrows are broadened 
and also that the shape of the pattern is dis¬ 
torted from that desired. The oscilloscope gives 
the actual instantaneous conditions and there¬ 
fore will indicate the quality of the wave prop¬ 
agation. If the arrow points are sharp and 
steady, the operator can always be sure that 
the correct bearing is indicated. 





















146 


DEMOUNTABLE SHORT-WAVE DIRECTION FINDER 




Figure 8. Typical patterns illustrating different 
of noise and signal. 

A. Very strong signal at 90° (or 270°) giving 
sharp arrow at indications. By manipulation of 
gain control, arrow may be widened or sharpened 
but no change of bearing is caused by this opera¬ 
tion. 

B. Good keyed signal showing circle, smaller 
than scale circle, when carrier is off air and good 
indication when carrier is on. Circle can be in¬ 
creased in size by decreasing bias on deflection 
amplifier. 

C. Signal weaker than A and B, but still giving 
good bearing indication. Arrow points are sharp 
although maximum signal at center does not cause 
trace to pull in toward center as in A and B. 

D. Much weaker signal in which gain control of 
receiver has been turned up so high that noise is 
shown on pattern. Bearing is still readable to 
within ±3° and possibly higher, since time ex¬ 
posure caused blur of moving arrow points. 

Because of the type of indicator employed the 
operator is able to take bearings on signals of 
very short duration, or on signals which are 
rapidly changing at the point of reception. He 
is provided with a continuous accurate picture 
of the arrival of the waves and in a short time 


types of images secured under different conditions 


E. Modulated signal, strong and without noise, 
but showing modulation envelope all around out¬ 
side of pattern. Bearing is not changed by this 
modulation and is readable to accuracy of ±1°. 

F. Time exposure of very weak fading signal. Ex¬ 
cept for center, entire screen is covered by noise 
patterns but bearing is still readable to ±5°. Sig¬ 
nal is not changing in apparent direction; has 
little or no polarization error. 

G. Time exposure when no signal is being received 
and with gain control turned to maximum. Al¬ 
though noise appears to have directional char¬ 
acteristics, this would not be visible and is shown 
here by long time exposure required. 

H. Appearance of two keyed signals on same fre¬ 
quency. The one at 90 to 270° is strong and gives 
good pattern. The one at 42 to 122° is weaker and 
shows noise envelope as well as slight movement. 

should be able to take very reliable bearings 
under conditions which have hitherto made 
readings impossible. 

If a bearing is shifting and also fading, it 
will usually be found that the nulls are more 
rounded at one indication than at others. It 










INTERPRETATION OF PATTERNS 


147 


will also probably be found that one indication 
is given with very good nulls. This indication 
may last only a fraction of a second., but the 
operator should always remember that the 
sharpest nulls correspond to the most nearly 
correct bearing. They always correspond also 
to the strongest reception during the fading 
cycle. Round nulls indicate that the bearing is 
uncertain or in error. 


Seldom will no bearing be possible. Bearings 
will differ only in the degree of accuracy with 
which they can be read. The most common im¬ 
possible bearings will be when the noise level is 
higher than the signal. (Figure 8, F and G.) 

A number of patterns are shown in Figure 8 
to illustrate the different types of images which 
may be secured under different conditions of 
signal and noise. 



Chapter 10 

DIRECTION FINDING BY IMPROVISED MEANS 


A study 3 to determine if effective direction finding 
could be done by the Armed Forces in war theaters, 
using only a radio receiver with no special measuring 
equipment and only such antennas as could easily be 
improvised in the field. Methods using loops and low 
horizontal wires were developed. A scheme using low 
horizontal radial wires each radial having two wires 
one above the other gave reasonably accurate locations 
for angles up to about 80° with the horizontal. The 
text that follows is condensed from the contractor’s 
final report. 1 

INTRODUCTION 

wo general types of antennas were tried, 
(1) simple loops and (2) various arrange¬ 
ments of low horizontal wires. Table 1 sum¬ 
marizes the results that may be obtained with 
six schemes briefly described in the table. 

The preliminary work indicates that: (1) 
For strong ground-wave signals, the loop 
scheme is indicated; (2) for weak ground- 
wave signals, a rough location may be obtained 
with a single wire at or near the ground, 
walked around a central radio receiver. A 
more accurate location may be obtained using 
eight radial wires at or near the ground, each 
wire a wavelength or more in length; (3) for 
sky waves coming from distances of 150 miles 
or more, scheme (2) is suitable. For sky waves 
coming from distances between about 50 and 
150 miles, a scheme involving eight radials, 
each having two wires, one above the other, is 
indicated; (4) the work on the loop schemes 
might well be extended to determine methods 
of locating stations sending sky waves from 
distances of 150 miles or more. At present, 
direction but not sense can be determined. 
Further work on the scheme involving double¬ 
wire radials is indicated. A mathematical anal¬ 
ysis would be useful to determine a further 
program of experiments which might lead to 
refinements and a more accurate determination 
of the precision of location. 

With the loop antenna, the test procedure 
is to turn the loop for a minimum signal and 
then reverse the loop by 180° and again find a 
minimum signal. The best estimate as to sta¬ 
tion direction is obtained by bisecting the angle 

3 Project 13-101, Contract No. OEMsr-1410, Western 
Electric Co. 


between the two minimum-signal positions. A 
special connection described below permits ob¬ 
taining the sense on ground-wave transmis¬ 
sions. 

All schemes described require an a-m re¬ 
ceiver with a beat-frequency oscillator (for 
producing an audible tone from the carrier) 
and with the automatic volume control, if any, 
disabled. 

Fairly extensive tests were made of the sys¬ 
tem listed under item 6 in Table 1. Part of the 
tests consisted of locating a mobile station, the 
direction of which was unknown to the test 
crew. The distance of the mobile station 
ranged from 50 to 112 miles. The power into 
the transmitting antenna was only about 2 
watts at 4.8 me and 6.425 me. Five tests were 
made. The average error was about 8° and 
the maximum error was 22°. The received 
signal was entirely sky wave. b The transmit¬ 
ting antenna consisted of a half-wave hori¬ 
zontal wire about 2 ft above the earth. 

io.2 EXPERIMENTAL WORK 

Loop Antenna 

Two-turn untuned loops, shown in Figure 1, 
were found to be satisfactory for direction 
finding on vertically polarized ground waves 
between 2 and 20 me. The 2-ft loop is usable 
in the 2- to 10-mc range and the 1-ft loop in the 
5- to 20-mc range. In the 5- to 10-mc range it 
was found to be more advantageous to use the 
smaller loop provided the received signal is 
sufficiently strong. 

At first one terminal of the loop was con¬ 
nected to the antenna post of the set and the 
other terminal was connected to the ground 
post of the set. This was found unsatisfactory 
since there was no sharply defined null when 
the loop was placed broadside to the direction 
of the transmitting station. (Normally a loop 
is operated into a balanced circuit.) However, 
if one terminal of the loop was connected to 

b The tests were made in the daytime. It is believed 
that F-layer reflections were involved at both these fre¬ 
quencies. See IRPL-E1, issued September 1944, Figure 
15, and TM 11-499, page 45 (both obtainable from 
Office of the Chief Signal Officer). 






EXPERIMENTAL WORK 


149 


the antenna post of the set and the midpoint 
of the loop was connected to the ground post 
of the set with the other terminal of the loop 
open, there was a sufficiently defined null when 
the loop was placed broadside to the direction 
of the transmitting station. The broadside null 
for one orientation of the loop was within about 
10° of the null for the 180° loop reversal. The 
average of the nulls gave an indicated direction 
within a maximum of about 5° of the true 
direction from 2 to 20 me when the transmit¬ 
ting station was about one-half mile away over 
level terrain. 


Not only the direction of the station but the 
sense also may be obtained. The sense is ob¬ 
tained by placing the loop in the plane of 
maximum signal, a position at right angles to 
the average of the null, and then touching with 
the finger or with a pair of pliers held in the 
hand the free terminal of the loop. With the 
transmitting station in the direction shown in 
Figure IB there will be an increase in the 
signal. If the transmitting station were in a 
position 180° from that shown with the loop in 
the same position, then touching the free end 
of the loop would result in a reduction of the 


Table 1. Field of use of improvised direction-finding schemes. 




Coverage 

Estimated error § 

Scheme 

Kind of antenna 

Freq. 


Sky wave * 


Sky wave 

No. 

(See Figures 1, 2, and 3) 

range* 
in me 

Ground 

wave"*" 

o 

00 

1 

o 

o 

Below 

70° 

Ground 

wave”! 

o 

o 

1 

o 

Below 

70° 

1 

Simple 2-turn loop," midpoint con¬ 
nected to radio-set ground post, one 
end to antenna post, other end float¬ 
ing. 

2 to 20 

Yes 

No 

Data in¬ 

±10° 




complete 










2 

Single horizontal wire on ground X or 
more 11 in length, walked around cen¬ 
tral radio receiver. 

3 to 20 

Yes 

No 

No 

± 40° 






3 

Single horizontal wire A/4 in length 
and supported 3 feet high, walked 
around central radio receiver. 

2 to 20 

Yes 

No 

No 

±30° 






4 

Fixed system of four wires on the 
ground or up to 3 ft in the air, X or 
more in length, radially at 90° in¬ 
tervals, with central receiver. 

2 to 20** 

Yes 

No 

Yes 

±23° 


±23° 



5 

Fixed system of eight wires, X or more 
in length, radially at 45° intervals, 
with central receivers. 

2 to 20** 

Yes 

No 

Yes 

±8° 


±8° 




6 

Fixed system of eight double wires, 
radially at 45° intervals, with central 
receivers; lower wire of each pair 
0.9 to 1.1A in length.t* 

2 to 8t+ 

Yes 

Yes 

Yes 

1+ 

00 

o 

±8° to 
±20° 

±8° 


* For sky wave, do not include frequencies above maximum usable frequency in each particular case, 
t At ranges where ground wave and sky wave are of nearly the same magnitude, direction cannot be determined. 

t Angles given are vertical angles of arrival of the wave front. Directly overhead = 90°. Below 70° corresponds to distances of 100 to 150 miles or 
more. 

§ Both direction and sense are obtainable for all schemes listed, by means described in the test. In the low wire schemes there is no “standard-wave 
error” as defined by R. H. Barfield in the I.E.E. Journal, London, Vol. 76, p. 423. 

" Loop is untuned, hence strong signal is required. 

II X is a wavelength on the ground return circuit (not in free space); for X in feet see text. 

** Tests indicate difficulties above 10 to 15 me if wires are raised above the earth, 
tt Higher frequencies not tested. 

+t Order of magnitude of distance to unknown station may be obtained from observations at a single d-f station. Spread of estimated error between 
70 to 80 degrees due to increasing “lateral deviation,” a phenomenon of the ionosphere. 
































150 


DIRECTION FINDING BY IMPROVISED MEANS 


signal. Tests in the 2- to 7-mc range showed 
that touching the free end of the loop with a 
wire connected to a vertical antenna produces 
the same effect as the operator’s finger, but the 
height of the vertical antenna must be adjusted 
for each frequency. 

Three tests (at 3.4925, 4.7975, and 6.425 
me) were made when the transmitting station 
was within ground-wave range and in direc¬ 
tions unknown to the operator. The average 
bearing error on these three tests was about 
10 °. 


» A 11 

2 -0 



A 



Figure 1. Side and perspective views of loops em¬ 
ployed in 2- to 10-mc range. For 10- to 20-mc range, 
loop should be 1 ft square. 

In the case of ground wave with an ap¬ 
preciable horizontally polarized component, the 
loop in broadside position was found to give 
lower minimum signal when the free turn was 
on the side toward the transmitter. 

The loops, when operated in a vertical plane 
as shown in Figure 1, could be used to find the 
direction but not the sense on signals arriving 
by sky wave from transmitters located at dis¬ 
tances exceeding 150 miles. The loop gave a 


lower minimum signal on sky wave when the 
free turn was on the side toward the transmit¬ 
ter. However, the results of a few tests made 
with the loop tilted off vertical and with the 
free turn on top indicated that a single null 
could be found which might give both the 
direction and distance of the transmitter. Time 
was not available to investigate completely this 
phase of the problem. 

A test was made on WWV at 5 me where 
the free end of the larger loop was connected 
to an eight-wire crowsfoot counterpoise, each 
toe being about 10 ft long. The radio receiver 
and associated power supply were not grounded 
other than through own capacitance to ground, 
but the connection of the loop to the receiver 
was through a short length of twisted pair in 
addition to the coaxial cable. With the loop 
and WWV in the relative positions shown in 
Figure IB a pronounced null was found. When 
the opposite edge of the loop was pointed at 
WWV there was a maximum. No broadside 
null existed; the typical cardioid pattern fa¬ 
miliar in the case of a loop combined with a 
vertical antenna was obtained. There was about 
10 db average difference between null and 
maximum. This scheme, with the same counter¬ 
poise, did not work on another station at a 
different frequency. These tests are mentioned 
to indicate that the field has not yet been fully 
explored. 

When the loop was connected with its two 
outer terminals to the antenna post and ground 
post, respectively, of the receiver, no null of 
any sort was obtained on WWV or on any other 
sky-wave signal. This was also true when the 
loop was connected to a balanced preamplifier, 
except that in the latter case London and San 
Francisco stations were found to produce some 
broadside null. 

There is some evidence that the special con¬ 
nection of the loop to the set, described above, 
tends to reduce its response to the horizontally 
polarized component of a wave, particularly in 
the case when the free turn is on the side of the 
loop toward the oncoming wave. 

For sense location the receiver must be 
placed at ground level, not inside a vehicle, 
with the loop directly over the set. To facilitate 
turning the loop any simple mechanical con¬ 
struction may be used. A simple method is to 












EXPERIMENTAL WORK 


151 


suspend the loop by a string from a light scaf¬ 
fold directly over the receiver with lead-in 
consisting of a flexible coaxial cable extending 
vertically below it to the receiver. The height 
of the bottom of the loop above the ground 
should not be over about 3 ft. Army-Navy type 
No. KG 8/U cable is satisfactory. Twisted pair 
(for example, W-110B) will not work well for 
sense indication but may be used for direction 
indication up to about 7 me if coaxial cable is 
unavailable. 


Low Horizontal Wires 

Low horizontal wires may be used as direc¬ 
tional antennas. When the length is %a or 
more, a stronger signal is received when the 
wave is traveling from the free end of the wire 
(front) toward the receiver than is received 
when the wave is traveling in the opposite 
direction (back) or from the side. The front- 
to-back ratio will amount to from 5 to 15 db 
for a wire one wavelength or more in length. 
For a given length of wire (>A) the front-to- 
back ratio is greater with higher attenuation. 
That is, there is a greater front-to-back ratio 
with a wire on the ground than with one sup¬ 
ported at some distance above the earth (not 
over three feet in connection with this work). 

The following rule may be used for deter¬ 
mining the physical length of one-wavelength 
wire: 


No. of feet 
In the clear, 


900 



, l }/2 to 3 feet above ground. 


No. of feet = , on short grass. 

J me 


A a/4 horizontal wire is not appreciably 
directional in itself, but becomes directional 
when supported in the air and associated with 
a vertical down-lead. That is, a A/4 wire on 
the ground is not usefully directional as con¬ 
cerns front-to-back ratio, but when raised 11/2 
to 3 ft in the air, with a vertical down-lead at 
the receiving end, it becomes directional. In 
this case, the stronger signal is received when 
the wave is traveling parallel to the wire from 
the receiver end of the wire toward the free 
end. A A/4 wire supported up to 3 ft in the air 
gave poor discrimination in azimuth. 


A fixed one-wavelength wire did not give 
good results on horizontally polarized ground 
waves. 

Calculation and test show that the %A wire 
has a larger front-to-back ratio than the A 
wire when the wave approaches the wire from 
a high vertical angle. However, it has a smaller 
front-to-back ratio than the A wire when the 
wave approaches at a low angle. The simple A 
wire gives a good front-to-back ratio on low- 
angle direction of the wave or on ground wave. 
This ratio is of the order of 6 db. A wire 
longer than A gives a still higher front-to-back 
ratio. The ratio is affected by attenuation, as 
noted above, and therefore is affected by 
ground constants as well as by wire height 
above ground. The %A wire is poor in azi¬ 
muthal discrimination and therefore is not rec¬ 
ommended for use. As discussed later, the 
A/4 wire is useful as a walked wire in ground- 
wave direction finding but is not successful 
when used in the fixed-wire radial scheme. The 
down-lead 1 % ft to 3 ft high also has an effect 
on the front-to-back ratio of the %A wire or 
A wire, but the effect may be ignored below 
about 8 to 10 me. 

Walked Wires 

A one-wavelength wire may be used for di¬ 
rection finding on vertically polarized ground 
waves. The terrain requirements are satisfied 
by an open field with short grass or weeds of 
fairly uniform height. The walked wire re¬ 
quires a steady signal, hence it is not useful 
when fading is present. This limits its use to 
ground-wave signals. 

A full-wavelength wire cut for use on the 
ground is laid out over grass or weeds and the 
radio receiver is connected to one end of it. The 
unknown signal is tuned in, and the wire is 
walked around to a direction 90° from the first 
position. At this point the wire is again laid 
back on the ground and another observation is 
taken. By progressing 90° at a time, one di¬ 
rection or two directions at about 90° to each 
other will be found where the signal is fainter 
than in the other two directions. Further 
moves, making smaller angular adjustments in 
the general minimum direction, will disclose a 
position that gives the faintest signal. In this 





152 


DIRECTION FINDING BY IMPROVISED MEANS 


position the outer end of the wire is pointing 
away from the “unknown” station. 

This scheme is cumbersome at the lower fre¬ 
quencies because of the length of the wire, 
which becomes difficult to handle and takes a 
relatively long time to move through an ap¬ 
preciable angle. The scheme works best when 
a steady carrier is present. With short bursts 
of carrier, as on average push-to-talk phone 
operation, the system does not give good re¬ 
sults. With c-w or m-c-w telegraph, with 
steady sending, fair results can be obtained 
on ground waves. 

Quarter-Wavelength Wire. The raised A/4 
wire with iy 2 to 3 ft vertical down-lead is less 
cumbersome than the full-wave wire. In this 
case the wire must be held in the air and as 
nearly parallel to the earth as possible while 
walking the outer end around. It is preferable 
to insulate the wire from the walker’s hand. 
When minimum signal is received, then the 
outer end of the wire is pointing toward the un¬ 
known station. 

This scheme will work with either vertically 
or horizontally polarized ground waves. Where 
the wave is horizontally polarized, two posi¬ 
tions giving low signals will be found; one 
with the wire pointing away from the station 
and one with the wire pointing toward the sta¬ 
tion. The one with the wire pointing toward 
the station will be found to be the lowest. 

Fixed Multiple Antenna Systems 

The use of fixed multiple wires (four or 
eight), A or more in length, around a central 
receiver, in connection with a key which per¬ 
mits rapid switching from one wire to another, 
may be used for direction finding on ground 
waves or on sky waves. The wires may be laid 
on the ground or supported in the air up to a 
height of 3 ft. Above 10 me better results are 
obtained with wires on the ground, unless they 
are longer than A. A length of 2a or more and 
a height of iy 2 ft is satisfactory in the 10- to 
20 -mc range. 

Experiment and calculation showed that 
where the signal was due to sky waves arriving 
at the receiving site at an angle greater than 
about 70° above the horizon, the system was 
inclined to fail. Fortunately, a failure is re¬ 


vealed in the measurements; hence there is 
avoided the possibility of taking the “bad” mea¬ 
surements seriously. The failure is revealed 
by the following symptoms in the test results. 

1. The combination of opposite wires giving 
the greatest front-to-back ratio does not con¬ 
tain the wires which have on them the strongest 
signals. 

2 . There may be no consistent front-to-back 
ratio. 

In an example of (1) the station was due 
north transmitting into a A/2 horizontal wire 
2 ft high. The greatest front-to-back ratio was 
SW-NE with SW greater than NE. E and W 
were greater than SW, and were about equal. 
As an example of (2), all wires were about 
equal or they changed back and forth during a 
fading cycle. At the receiving site near Flor- 
ham Park, N. J., it was impossible to determine 
the direction of Floyd Bennett Tower on Long 
Island at about 7 me by the above method. The 
airline distance is about 40 miles. It was pos¬ 
sible to determine the direction of WWV in 
Washington, D. C., at 5 me. The airline dis¬ 
tance is about 180 miles. Successful direction 
finding was also done by the above method from 
6 to about 15 me on Montreal, Halifax, Toron¬ 
to, London, San Francisco, and on Bound Brook 
and Wayne, N. J. The last two stations came 
in on ground wave. Successful direction finding 
was also done on the project transmitting sta¬ 
tion when it was about 150 miles distant and 
transmitting successively at 4.7975 me and 
6.425 me. 

The above method may employ either bare or 
insulated wires on the ground provided (1) 
there is no grass or weeds or (2) the grass or 
weeds are short and uniform where the wires 
are located. If there are weeds or undergrowth 
of nonuniform height, the wires should be 
supported in the air from 1 y 2 to 3 ft high. 
All the wires in a given layout should be placed 
at the same height and all should be of the 
same kind of wire. Test indicates difficulties 
above 10 to 15 me if the wires are raised above 
the earth. 

The receiving site should be level. A 5-degree 
slope, however, will not give trouble. The an¬ 
tennas may be placed in a forest provided 
raised wires are used (1 y 2 to 3 ft high) and 
growth is cut away below the wires to heights 



EXPERIMENTAL WORK 


153 


of not over 4 to 6 in. and laterally to a distance 
of at least 5 ft. Sites near overhead wire lines, 
wire fences, other antennas or other metallic 
structures which will distort the field pattern 
should be avoided. The character of the terrain 
under all the wires and the growth near them 
should be approximately the same. For exam¬ 
ple, it is unwise to put half the wires in the 
woods and the other half in the open. 

When using this method on a fading signal, 
a phenomenon is present which results in a 
change of front-to-back ratio during the fading 
cycle. For example, when listening alternately 
on wires having about equal signal strength, 
such as the two wires that are nearly at right 
angles to the direction of travel of the wave, 
first one wire and then the other may have a 
stronger signal. It may be necessary to make 
the listening comparison for some time to de¬ 
termine which wire gives the stronger signal 
most of the time. The front-to-back ratio on 
wires most nearly pointing toward and away 
from the transmitter is consistently in the same 
direction, but may vary over a range of from 
1 to 10 db. This may be quite disconcerting to 
the operator at first, but it was found that 
practice in measurements leads to quickness in 
their interpretation. This practice may be ob¬ 
tained by first testing on known stations. 

Double Wires. When the angle of arrival of 
the sky wave is greater than about 70° above 
the horizon, (beyond ground-wave range but 
less than 150 miles) experiment showed that 
some other scheme than those described above 
was necessary. It was found that if two wires, 
one above the other, were put out on each of 
the eight legs and a slightly different test pro¬ 
cedure were used, a great improvement was 
obtained in short-distance sky-wave direction 
finding. In this arrangement each leg consisted 
of a A wire li/2 ft high and a A/2 wire 3 ft 
high directly above the first wire. In the fol¬ 
lowing discussion the A wire will be called the 
low wire and the A/2 wire will be called the 
high wire. The receiver was rapidly connected 
in succession between two opposite low wires. 
The associated high wires were connected 
through and connected to the ground post of 
the receiver which was also grounded to three 
ground rods connected in parallel. The other 
wires of the system remained open and clear. 


Now when receiving a high-angle signal which 
is moving in the direction of the plane contain¬ 
ing the wires, the connection of the high wires 
as noted above increases the signal from the 
low wire pointing toward the station in ques¬ 
tion and decreases the signal from the low 
wire pointing away from the station. Thus in 
a case which, without connection of the high 
wires, would give a front-to-back ratio of unity 
or less, the connection of the high wires gives 
a front-to-back ratio greater than unity, i.e., 
the low wire pointing toward the station (free 
end toward station) has more signal than the 
low wire pointing away from the station. To 
facilitate quick changes, all wires were brought 
in to a breadboard having Fahnstock terminals 
and with the switching key screwed to the 
board. The arrangement is illustrated in Fig¬ 
ure 2. 


A 


RADIO RECEIVER AND 
SWITCHING ARRANGEMENT 


L» _ HW \ J _% _ HW A '''jig ♦ 

///////////yZ/ ,, x / //yy °" 

SIDE ELEVATION OF TWO OPPOSITE DOUBLE WIRE ANTENNAS 



PERSPECTIVE SHOWING TWO OPPOSITE DOUBLE WIRE ANTENNAS 



DIAGRAM SHOWING SWITCHING BREADBOARD WITH FAHNSTOCK 
CLIP TERMINALS TO WHICH LOW { LW) AND HIGH WIRE(HW) 
ANTENNAS ARE CONNECTED BY SPACED WIRE TRANSMISSION 
LINES. TELEGRAPH KEY AND HIGH WIRE SHORT AND GROUND 
SHOWN AS CONNECTED FOR EAST-WEST COMPARISON. 

Figure 2. Double-wire direction-finding scheme. 
A. side elevation of two opposite double-wire an¬ 
tennas; B. perspective showing double-wire ar¬ 
rangement ; C. switching arrangement with key and 
high wire short connected for east-west compari¬ 
son. 

























154 


DIRECTION FINDING BY IMPROVISED MEANS 


With this arrangement it was possible to 
“locate” a mobile station out to 20 miles. There 
was then a blind ring out to about 60 miles, 
and from 60 miles on out it was again possible 
to determine the direction of the mobile sta¬ 
tion. The extent of the blind ring would vary 
in different cases depending on frequency, 
earth constants and on the type of antenna 
used at the transmitter. The mobile station in 
question used a low horizontal A/2 antenna. 
Had the mobile station used a whip or other 
vertical antenna, the ground-wave range un¬ 
doubtedly would have been extended. 

The theory of this system has not yet been 
completely worked out, but successful use was 
made of it over a period of about two weeks. 

Five tests were made where the mobile sta¬ 
tion went to locations that were unknown to 
the measuring crew. The only stipulation was 
that the distance should be greater than 50 
miles. Two frequencies were tested at each 
location, 4.7975 me and 6.425 me. The receiv¬ 
ing antennas were arranged to be quickly 
changed to the right lengths by means of in¬ 
sulators and jumpers. Both frequencies indi¬ 
cated the same direction, excepting for the last 
test, where it turned out that the mobile trans¬ 
mitter was 50 miles distant. The frequency 
6.425 me gave indeterminate results for this 
location. Table 2 shows the results. The power 
into the transmitting antennas was about 2 
watts. A Hammarlund radio receiver was 
used. 


Table 2. Tests on double-wire system.* 


Location 

Distance 
in miles 

True bearing 
in degrees 

Measured bearing 
in degrees 

1 

63 

322 

300 

2 

112 

335 

330 

3 

104 

15 

22.5 

4 

90 

359 

355-0 

5 

50 

359 

359 


* Average error = 8°. 


It is usual to interpolate the bearing between 
the 45° legs in 15° steps. However, in the case 
of the reading taken with the transmitter in 
Location 5, NW was thought to be just slightly 
stronger than NE and N was stronger than 
either of these. Hence it was concluded that 


the station was either due north or very slightly 
west of north. As can be seen in the table, it 
was really 1° west of north. 

A number of tests were made with the trans¬ 
mitter at 40 miles. This is in the blind ring. 
In about half the cases tested it was possible 
to obtain a bearing on the transmitter, most 
of them at 4.7975 me. However, in these cases 
the measured bearing usually came out too 
large by about one-half the angle between 
adjacent legs, i.e., 22.5°. 

In another set of tests the mobile transmitter 
was sent, as nearly as possible, due west. On 
the outward trip tests were made at 20, 40, 80, 
120 , and 200 miles. On the return trip tests 
were made at 160, 140, and 80 miles. Errors 
comparable in magnitude with those shown in 
Table 2 were obtained for all distances ex¬ 
cepting 40 miles. The readings for the latter 
distance were ambiguous. 

On this series of tests a meter was used in 
the receiver output and it was noted that for 
the 80-mile transmission, opening the high 
wire and clearing it from ground reversed the 
front-to-back ratio from a condition of west 
wire stronger than east wire by an average of 
about 4 db, to east wire stronger than west by 
about 1 db. At 120 miles with the high wire 
open and clear there was a 0-db (unity) front- 
to-back ratio, and with the high wire connected 
there was a front-to-back ratio of about 6 db 
with west stronger than east. At 200 miles 
with the high wire open and clear there was a 
2- or 3-db front-to-back ratio in the right di¬ 
rection (west greater than east) and with the 
high wire connected, the front-to-back ratio 
was about 8 db in the right direction. These 
comparisons were made at 4.7975 me. Obser¬ 
vations on a broadcast station in Toronto, Can¬ 
ada (about 500 miles), at 6+ me showed that 
there was no appreciable difference between 
front-to-back ratio with the high wire con¬ 
nected or disconnected. The above results sug¬ 
gest that, after further checks of the phe¬ 
nomena, these comparisons might well be used 
as a criterion of the order of magnitude of the 
distance of the unknown station. 

The high wire and low wire were brought 
from the terminal stake of each of the antennas 
to the switching breadboard as spaced trans¬ 
mission lines not over 5 ft long. The wires in 











THEORETICAL DEVELOPMENT 


155 


each transmission line were separated by about 
3 in. Using twisted pair for lead-in was not 
tried. See Figure 2B. 

The above method permits direction finding 
on relatively weak signals and on signals which 
are very close in frequency to other unwanted 
signals. This is due to the selective action of 
the ear in being able to identify and concen¬ 
trate on a tone of a particular pitch in con¬ 
trast to tones of other pitches or in contrast to 
noise. A full-wave wire 1% ft in the air 
delivers a stronger signal in the 2- to 8-mc 
range than a 15-ft whip when the wave is 
moving parallel to the wire from the outer end 
over average earth. 

103 GENERAL OPERATING NOTES 

In the case of any of the methods described 
above, a radio receiver outside of a vehicle 
must be used. The receiver must be placed at 
ground level for the a/ 4 wire. For sense loca¬ 
tion, in the case of loop direction finding, the 
receiver must also be at ground level with the 
loop directly over the set. For use of any of 
the full-wave (or more) wire schemes the re¬ 
ceiver may be mounted at table top level. The 
receiver and power supply is not grounded 
through other than its own capacitance to 
ground, excepting in the case of the double¬ 
wire scheme. In this case, the ground post of 
the set should be connected to a low-impedance 
driven ground located near the set. This scheme 
will work without the driven grounds but it 
worked better with them. 

Power should be supplied by battery and 
vibropack, both located close to the receiver. 
A gasoline-driven generator with rubber- 
covered line on the ground from generator to 
receiver would probably result in intolerable 
noise in the antennas. A hand-driven generator 
probably could be used provided a short length 
of line between generator and receiver were 
employed. 


104 SUPPLEMENTARY TESTS 

Tests were made using various other com¬ 
binations of single- and double-wire antennas, 


but in the time available no combinations were 
found that were more satisfactory than the 
ones described above. 

No tests were made above 20 me. In the 
range above 30 me it is likely that the best re¬ 
sults would be obtained by use of a reflector- 
director or other directional antenna. The 
method would not be applicable to f-m re¬ 
ceivers, on account of the limiter action and the 
lack of a beating oscillator. 

10 5 POSSIBLE REFINEMENTS 

The fixed-wire methods could be extended in 
a way that might afford a field of usefulness 
in other than forward area military direction 
finding. The refinements would require more 
apparatus, for example means to connect two 
antennas through some goniometer coupling 
device to an oscilloscope for purposes of phase 
as well as magnitude comparisons. 

Modifications can be imagined that would 
permit rapid direction finding as in many of the 
existing commercial Adcock systems or spaced- 
loop systems. Such modifications, applied to the 
double-wire scheme with eight or more radials, 
might have particular advantages for high- 
angle sky-wave direction finding where the Ad¬ 
cock and spaced-loop systems run into difficul¬ 
ties. 

The use of the loop with free turns should 
also be investigated. It is possible that a single 
loop arranged in this way to suppress the hori¬ 
zontally polarized component might be used in 
place of two spaced loops. 

10 6 THEORETICAL DEVELOPMENT 

The proportions of the antennas described in 
this report were determined by cut and try 
methods. The general theory of loops and long- 
wire antennas was used as a qualitative guide. 
It appeared that this method of attacking the 
problem would be more efficient than propor¬ 
tioning the antenna structures on the basis of 
predetermined calculation formulas. The ex¬ 
perimental work has given clues to the physical 
approximations which are justified and which 
are essential to obtaining reasonably compact 
calculation formulas. 






156 


DIRECTION FINDING BY IMPROVISED MEANS 


A calculation formula has been worked out 
for a simple case. This formula will be dis¬ 
cussed together with the physical approxima¬ 
tions leading up to it. The method of analysis 
could be extended to cover the more complicated 
cases. Figure 3 shows the calculation formula 
for the case of two collinear horizontal wires 
at or near the ground and free from ground 
at both ends. The open-circuit voltages to 
ground at the inner ends of the two wires were 
calculated. If the antenna terminal of a 
grounded radio-receiving set (which in itself 
does not pick up any voltage from the radio 
field) is switched from one wire to the other, 
the relative amplitude of the sounds heard at 
the output of the receiving set will be propor¬ 
tional to the relative magnitudes of the open- 
circuit voltages. Therefore, the magnitude of 
the ratio of these two calculated voltages gives 
the observed front-to-back ratio. It is assumed, 
of course, that the radio set does not have 
automatic volume control. 

The radio set does, in itself, pick up voltage 
from the radio field if the down-lead is regard¬ 
ed as a part of the set. Experiments indicated 


that if the down-lead were not more than about 
14 o of the length of the horizontal wire, the 
effect was unimportant. The theory could be 
extended to include the effect of voltages in¬ 
troduced in the down-leads. 

The experiments also indicated that it was 
satisfactory to assume that each horizontal 
wire was a ground-return transmission line 
with uniformly distributed constants and 100 
per cent reflections at the open ends. Since the 
wires are near the ground, the transmission 
line may be regarded as having uniformly dis¬ 
tributed resistance. This is caused largely by 
losses in the ground; the radiation resistance 
is relatively unimportant. The magnitude of 
the voltage induced per unit length in the 
horizontal wire may be assumed to be the same 
at all points of the wire. The coupling between 
the wires forming different radials may be 
ignored. 

As noted in Figure 4, the front-to-back ratio 
would be unity if the transmission line had no 
attenuation. If the attenuation is large, a very 
simple expression for front-to-back ratio is 
obtained. 


Table 3. Front-to-back ratio for pair of collinear wires at or near the ground. 


(Propagation constant of wire ground circuit = y = a + jfi, (3 = 2‘7r/\, \=*k\, 
\ o = wavelength for propagation in air.) 







Front-to-back ratios for wires of length 

Elevation 

Azimuth 

Attenuation 








angle in 

angle <f> in 

in 


X 0 in 






degrees 

degrees 

nepers 

k 

meters 

X/8 

X/4 

X/2 

%\ 

X 

10 

0 

0.32 

0.92 

50 

1.00 + 

1.01 

1.16 

1.26 

1.34 

10 


0.64 

0.92 


1.00 + 

1.02 

1.34 

1.54 

1.78 

10 


0.67 

0.70f 


1.00 + 

1.02 

1.26 

1.73 

1.56 

80 


0.32 

0.92 


1.00 + 

1.00 + 

1.03 

1.38 

1.06 

80 


0.64 

0.92 


1.005 

1.01 

1.05 

1.85 

1.14 

80 


0.67 

0.70f 


1.00 + 

1.00 + 

1.04 

1.67 

1.13 

80 

0 

0.64 

0.92 

50 

1.005 

1.01 

1.05 

1.85 

1.14 


30 




1.00 + 

1.00 + 

1.04 

1.69 

1.13 


45 




1.00 + 

1.00 + 

1.04 

1.60 

1.11 


60 




1.00 + 

1.00 + 

1.03 

1.41 

1.11 


75 




1.00 + 

1.00 + 

1.01 

1.20 

1.11 


90 




1.00 

1.00 

1.00 

1.00 

1.00 

10 

0 

0.64 

0.92 

50 

1.00 + 

1.02 

1.34 

1.54 

1.78 


30 




1.00 + 

1.02 

1.29 

1.62 

1.63 


43 




1.00 + 

1.01 

1.22 

1.75 

1.49 


80 




1.00 + 

1.01 

1.15 

1.99 

1.33 


75 




1.00 + 

1.005 

1.08 

2.12 

1.17 


90 

1 




1.00 

1.00 

1.00 

1.00 

1.00 



* Per cent of the speed of a wave in apace, 
t Wire on the ground, wire-ground speed. 
























THEORETICAL DEVELOPMENT 


157 



1 

\ww\n\n\\\n 

Normal to Wave Front of Arriving E lliptically Polarised Plane Wave 


I C I 

Front to back ratio = — = I if «*= o 

e e. 


lit • l fe Kt «, 

0-9, ^ 


propagation constant per meter of wire-ground transmission fine 
0—angle of elevation with respect to the horizontal, 
jangle of agi rnuth with respect to antenna wire. 

(3 o = phase change constant for propagation through air = ^ raciians/meter. 
(3, = Q> 0 cos e cos <f>. 

r v and r^ are vector reflection coefficients for vertically or 
horizontally polarised plane waves. 


A~-r w sin e cos <j> (j- r v £* ^ ^ r h f—--) 


- 2 J 0 O H sin 


'•**7 | - e : *«•- — 

‘7 i- £ - 2 ^ 


®i~ -K^ - co»J,r£- j llsinh ar.fi.) =-k(£ +J ^ cosh yl- sinh yfi.) 
e t = KCe^-cosh yl+J 3inh y£) = K(£ J<J ‘ A - coshy£+Ji. s -, n h y£) 

aAy -f'^ r !° e '^ 

o-r^)(y z + 


2. 

K * 


for @1= air. 


e, 


cos@£- cosh *JL- s i n ^i. 


coss£- c.oshxJl + CisInhc<-& j sj^eJi 


Figure 3. Calculation of front-to-back ratio; sky-wave reception. 































158 


DIRECTION FINDING BY IMPROVISED MEANS 


The formulas may be used for calculating 
ground-wave front-to-back ratios. In this case 

A = F v tan T cos </> 

where T = the tilt angle of the ground wave. 
Since 

0 = 0 

Pi = Po cos <j>. 

Table 3 gives calculations of front-to-back 
ratios for various assumptions regarding atten¬ 
uation and length of the wires. The values of 
attenuation were chosen as the result of previ¬ 
ous calculations and measurements on low hori¬ 
zontal wires. The attenuation per wavelength 
(of the wire ground circuit) is roughly constant 
(2:1 variation) over a range of about 2 to 10 
me, for heights from about 2 cm to about 1 
meter and over a range of ground con¬ 
ductivities from about 0.001 to 0.01 mho per 
meter. Values between 0.6 and 0.7 neper per 
wavelength were chosen for calculation. Since 
these values seem somewhat on the high side, 
a value of about one-half as much was also 
used to indicate the effect of reduced attenua¬ 
tion. 

The order of magnitude of the calculated 
ratios is the same as those measured except 
that the experiments have a reversed front-to- 
back ratio for A/4 wires 3 ft above the ground. 
The experiments showed these to be due to 
voltages induced in the down-leads. Computa¬ 
tions of front-to-back ratios were first made 
for zero azimuth angle, i.e., for collinear wires, 
one of which points at the transmitter. For 
such wires the angle <f> = 0. For lengths of wire 
which gave a sizable front-to-back ratio, com¬ 
putations were also made for values of <f> from 
0° to 90°. These correspond to collinear radial 
wires not pointing at the transmitter. It will 
be seen from Table 3 that the front-to-back 
ratio does not shrink to unity rapidly as <f> in¬ 
creases from zero. In one case the maximum 
front-to-back ratio is obtained on the pair of 
wires not pointing toward the transmitter. 
Such false indications were noted during the 
experiments when arrangements of wires not 
of the optimum length were used. Since the 
measurements of only front-to-back ratios do 
not give a sensitive indication of the direction 
of the transmitter, it is necessary to compare 


open-circuit voltages at the inner ends of the 
wires which are not in line. 

Taking as a reference a wire pointing to¬ 
ward the transmitter and for which = 0 and 
considering other wires of angular displace¬ 
ment d=</>, an inspection of the formulas shows 
that the magnitude of the open-circuit voltage 
for any wire is approximately proportional to: 

J ga | I * +m ~ cosh yl — ~ sinh yl \. 

In the above expression, a is neglected in the 
terms having (3 in the denominator, i.e., p is sub¬ 
stituted for a + jp. Since p 1 = cos <f> cos 0 it 
is a function of <£. A is also a function of <f>. 

For ground waves A is proportional to cos <j>. 
For high-angle sky waves (0 = 70° to 90°), 
the reflection coefficients r v and r h are about 
equal in phase and magnitude. Assuming them 
alike and assuming F v and F h have the same 
rms values for a time interval of a few seconds 
but are related at random as to instantaneous 
magnitude, A is proportional to 

a/ sin 2 0 cos 2 (f> + sin 2 0 . 

For low-angle sky waves, the resultant elec¬ 
tric force due to arriving and reflected waves is 
greater for vertical polarization than for hori¬ 
zontal polarization, that is, 

t> = (1 -v" 2iM/sin *j 
is greater than 

h = (1 — r h e~ 2j ®" Hsin e ) 

For 6 me, ground conductivity of 0.008 mho 
per meter and ground dielectric constant of 10, 
the following values of |v| and \h\ result if the 
wires are 3 ft above ground: 


0 

r v 

Th 

M 

1*1 

10 ° 

0 . 3 / 100 ° 

0 . 96 / 2 ° 

1.10 

0.084 

50 ° 

0 . 66 / 16 ° 

0 . 79 / 9 . 6 ° 

0.50 

0.37 

80 ° 

0 . 725 / 12 . 5 ° 

0 . 74 / 12 ° 

0.46 

0.45 


For angles where \h\ is appreciably less than 
\v\ the coefficient A is proportional to: 

V | v 2 1 sin 2 0 cos 2 0 + | h 2 1 sin 2 0 . 








THEORETICAL DEVELOPMENT 


159 


Using these assumptions and approxima¬ 
tions, the ratio of the voltage on a reference 
wire pointing toward the transmitter to the 
voltage on a wire of angular displacement ±<£ 
may be computed. Results of computations for 
%A and A (wire-ground speed) 3 ft above the 
earth are given in Figure 4. A frequency of 6 
me and values of elevation angle 0 of 10°, 50°, 
and 80° were used, a was taken as 0.014 neper 
per meter (0.64 neper per wavelength). 

It will be seen that %A wires which give 
relatively large front-to-back ratios for high- 
angle sky waves give poor azimuthal discrimi¬ 
nation. It will also be seen that A wires have 
good azimuthal directivity for low- or medium- 
angle sky waves. 



Figure 4. Voltage ratios for wires near ground. 










Chapter 11 


PORTABLE RADIO ASSAULT BEACON 


Development of a radio beacon to guide an infantry¬ 
man to an objective for a distance of 2,500 yd with an 
accuracy of ±3° using equipment already available. 
Choice of antenna systems, modulation methods, fre¬ 
quency, solution of key-click troubles are described. 
The greater part of the contractor’s final report 1 on 
this project is contained in this summary, the chief 
deletions being in descriptions of Army requirements, 
changes in requirements and in. description of methods 
of aligning antennas in the field. 

111 INTRODUCTION 

A t the beginning of this project portable 
l radio assault beacons were in use by the 
British for the guidance of tanks, the assemb¬ 
lage of paratroopers, etc. The British beacon 
used two Beverage antennas at right angles. 
Each antenna was several hundred feet long 
and was stretched along the ground, usually 
supported a few inches above the ground. 

In this country, some attempt was made to 
use a crossed-loop antenna system as a beacon 
for guiding troops to pill boxes, and other tar¬ 
gets through fog, smoke, jungle, or at night. 
The crossed loops, however, gave results which 
were inferior to the British system. This proj¬ 
ect studied the performance of the British 
beacon under various conditions of terrain, 
weather, antenna length, frequency, obstruc¬ 
tions, size of antenna wire, angle between the 
antennas, height of the wires above earth, in¬ 
equality of antenna currents, and the polariza¬ 
tion error under various conditions. 

Existing Army radio transmitters and re¬ 
ceivers were incorporated into a beacon similar 
to that of the British but provided with means 
for steering the defined course over an arc of 15° 
or 20° to obviate the necessity of laying out the 
long antennas very accurately. 

The original requirements that the course ac¬ 
curacy be ±y 2 ° were manifestly impossible of 
attainment because average site errors in d-f 
systems are greater than this figure. The re- 

a Project 13.1-100; Contract No. OEMsr-1261, Ray¬ 
mond M. Wilmotte. 


quirement was modified to be ±3°. Original 
instructions that the equipment was to be the 
best possible was also changed to a request 
that every effort be made to utilize equipment 
already available in the field and to make as 
few changes in this equipment as possible. 

These requirements limited the field of study 
considerably and finally the development was 
centered around the use of the SCR-536 re¬ 
ceiver and SCR-284 transmitter. 

1111 Selection of Type of Beacon 

Six types of beacons were considered. 

1. Crossed loops set at 45° to the required 
direction. 

2. Crossed loops set at 0° and 90° to the re¬ 
quired direction. 

3. Crossed Adcock antennas. 

4. Spaced antennas. 

5. British type using Beverage antenna. 

6. Modified British type using Beverage an¬ 
tenna. 

The d-f systems most commonly used are the 
crossed coil and the Adcock system. They have 
been used very successfully for aircraft naviga¬ 
tion. Because of portability requirements, it is 
clear that the Adcock type of antenna could 
only be used at very high frequencies. Because 
of site errors the use of very high frequencies 
was discarded. No work, therefore, was carried 
out with Adcock antennas. Work with crossed 
loops was successful but the results proved to 
be less satisfactory than with the British and 
modified British systems described below. 

11 2 LOOP ORIENTATION TO 

ELIMINATE KEY CLICKS 

It was soon found that one of the major 
difficulties was the elimination of key clicks 
which were likely to be so strong as to seriously 
reduce the sensitivity of operation. It was sug¬ 
gested that the crossed loops be located at 0° 
and 90° with respect to the desired direction 


160 



SELECTION OF FREQUENCY AND POLARIZATION 


161 


instead of at 45° angles with the direction in 
the center. The current in the 90° loop would 
have its current reversed in direction to 
produce the switching of the antenna pattern. 
The current in the 0° loop would remain 
constant so that the signal strength of the 
signal along the desired direction would not 
change during the switching period. This meth¬ 
od was found successful. 

“•» BRITISH AND MODIFIED 

BRITISH SYSTEMS 

The two systems which seemed to give the 
most promise were the British system and a 
modification of it. The British system consists 
of two Beverage antennas set at 45° to the 
required direction. The antennas are stretched 
a few inches above the ground and act in a 
manner very similar to the ordinary crossed- 
coil system. Like the crossed-coil system, the 
British beacon suffers from troubles due to 
key clicks. That problem was not solved for a 
considerable time, however. Eventually a relay 
was developed which reduced the key clicks to 
such low intensity that the British system was 
found to be accurate and sensitive. 

The modified British system was an attempt 
to eliminate the key clicks in a manner similar 
to that in which they are eliminated in the 
crossed-loop system, i.e., by locating the ground 
antenna at 90° to the required direction and 
by use of a vertical antenna. This system was 
found successful and from an operating point 
of view was almost identical in sensitivity to 
the British system but in certain respects of 
installation was somewhat more complicated. 

11 4 SELECTION OF MODULATION 

As regards modulation the British reported 
a hand-operated system in which an operator 
switches from one antenna to the other and 
simultaneously speaks into a microphone saying 
“left,” “right,” etc., as he switches. The listener 
then judges the relative intensity of the words 
“left” and “right” and goes to the left or right 
accordingly until the intensity of the two words 
appears approximately equal. Some British 
reports have indicated remarkable accuracy 
with this system of modulation. Experiments 
under this project, however, did not show the 


degree of accuracy claimed in those reports 
for normal operating conditions. Other forms 
of modulation systems such as the dot and dash 
systems were tried but were not found to be 
as accurate or as sensitive as the results ob¬ 
tained with the standard A and N system used 
on aircraft radio ranges. 

The modulation can be carried out in one of 
two ways. When the listener is away from the 
required direction he must hear a difference in 
intensity between the signals as the transmit¬ 
ting antenna is switched. This difference in 
intensity may be obtained either by a change in 
intensity of the radio frequency or a change in 
the percentage modulation. In practice it would 
be preferable to change the intensity of modu¬ 
lation because in that case the automatic vol¬ 
ume control [AVC] of the receiver could be 
used to its full extent without decreasing the 
sensitivity. When the r-f intensity is changed, 
however, it is essential that the AVC be elimi¬ 
nated, or that the time of the dots and dashes 
be short compared with the time constant of 
the a-v-c circuit, or that the intensity of the 
signal be sufficiently weak so that the AVC 
does not operate, or operates only partially. 

Since it was eventually decided to try to 
develop a system using equipment available in 
the field without making any internal modifica¬ 
tions, it was clearly not possible to use the sys¬ 
tem in which the percentage modulation was 
changed. The system eventually developed was 
based on the compromise of using a signal 
which was sufficiently weak so that the AVC 
of the receiver is only partially operative, 
thereby permitting the detection of changes in 
signal intensity. One method of detecting small 
changes in modulation was discussed but even¬ 
tually discarded because it would have required 
careful operation by the infantryman. This 
method consisted in using a limiter in the re¬ 
ceiver, such as is available in f-m receivers, and 
adjusting the level of the signal at the limiter 
so that small changes of signal intensity 
produced a large change in receiver output. 

ii s SELECTION OF FREQUENCY 
AND POLARIZATION 

The original requirement of an accuracy of 
±y 2 ° was greater than the accuracy normally 




162 


PORTABLE RADIO ASSAULT BEACON 


obtained with direction finding. Since it was 
also clear that any known beacon system would 
suffer from errors similar to d-f errors, it ap¬ 
peared that even though an accuracy of ± y%° 
might be obtained the absolute direction would 
probably not be known with an accuracy of 
better than ± 2° for most conditions, and 
under some conditions, a considerably greater 
error might be obtained. A few tests indicated 
that it was probable that a system could be 
developed which would give a high degree of 
sensitivity. Therefore, it became apparent that, 
for practical purposes, it was essential to keep 
site errors to a minimum. Originally it had 
been suggested that frequencies between 40 and 
48 me be used. This suggestion was based 
largely on experience with radio ranges at air¬ 
ports and because at those frequencies it might 
be possible to use spaced antennas thereby 
providing greater sensitivity than would be 
possible with the comparatively blunt type of 
directional pattern that a loop or Adcock an¬ 
tenna provides. Conditions in the field, how¬ 
ever, were found to be substantially different 
from the conditions found at airports. It was 
also decided that the vast amount of informa¬ 
tion available on d-f errors should be used in 
deciding which group of frequencies would 
reduce the site errors to a minimum or at least 
to practical values. Dr. Smith-Rose indicated 
from his experience, which also checked with 
the experience of the contractor, that the very 
high frequencies would produce greater errors 
than lower frequencies. He suggested using 
frequencies around 300 kc and lower, and that 
if such frequencies could be used it was proba¬ 
ble that site errors as low as 1° might be ob¬ 
tained. However, no equipment was available 
in the field for these low frequencies and the 
size of equipment for these frequencies was 
likely to be excessive if reasonable efficiency 
was to be obtained. Moreover, the Army in¬ 
dicated that these low frequencies might not be 
available for this use in the field. Smith-Rose 
pointed out that the errors increased with in¬ 
crease in frequency and reached a minor maxi¬ 
mum between 3 and 10 me for the reason 
that at those frequencies an average tree is 
approximately A/4 long and by its resonance 
causes comparatively large errors. He indi¬ 
cated that errors of the order of 2° could be 


expected in this range and suggested the use of 
frequencies around 15 me with an expectation 
of reducing the average site error by about i/2°- 
The frequency eventually selected was 5 me be¬ 
cause equipment was available in the field at 
those frequencies. 

Originally NDRC indicated an interest in 
studying the difference in operation between 
horizontally and vertically polarized waves. The 
selection of a frequency of 5 me eliminated the 
use of horizontally polarized waves, for within 
a few hundred feet of a horizontally polarized 
antenna most of the horizontally polarized 
waves seem to be eliminated and only the re¬ 
maining vertically polarized portion is received. 
The suggestion had been made because it was 
believed that the horizontally polarized waves 
would be able to travel farther through wooded 
territory and produce less error than the 
vertically polarized waves, since trees are main¬ 
ly vertical. No work was carried out on this 
angle of the project because of the decision to 
use a frequency in the h-f band instead of the 
v-h-f band, and because it was believed im¬ 
practical for an infantryman to carry a non- 
directional horizontally polarized antenna. 

11 6 EXPERIMENTAL RESULTS 
Crossed-Loop Beacon 

The crossed-loop system is shown in Figure 
1. The loop at right angles to the course is con- 



Figure 1. Crossed-loop type of course indicator. 

nected to the transmitter through a keyer 
which reverses the polarity of the currents in 
this loop in accordance with an A and N. It 





















EXPERIMENTAL RESULTS 


163 


was hoped that by performing the switching 
in that loop which has a null in the direction 
of the course, the occurrence of key clicks on 
course would be prevented. To avoid detuning 
the other loop whenever the keyed loop was dis¬ 
connected in the process of switching, it was 
necessary to resonate separately each loop. If 
the two loops were tuned by a single capacitor, 
the unkeyed loop was detuned to such an extent 
when the keyed loop was disconnected during 
the keying process that the keying was fully re¬ 
produced in this loop. The final circuit shown 
in Figure 1 was more complicated than the 
British system and the tuning procedure re¬ 
quired considerable care. 

The field strength obtained with a loop 
2 ft square was about % that of the British 
beacon and averaged about 12 ^v at li/ 2 miles. 
The width of the course was ±2° for a 1-db 
difference between the A and N signals. Ex- 


direction OF COURSE 



Figure 2. Antenna system used in British assault 
beacon. 


perience with loop antennas amply indicated 
that no appreciable improvement could be ob¬ 
tained by increasing the size of the loops and 
it was also felt that larger loops would be un¬ 
desirable from the standpoint of portability. 
Because of the low field strength, the greater 


course width and the greater difficulty of 
tuning this system, the crossed loops were 
abandoned. It is possible that the crossed-loop 
beacon would give satisfactory results at fre¬ 
quencies of the order of 40 to 48 me and also 
near 20 me. 

The British Type of Beacon 

The transmitter used was the SCR-284 which 
has a maximum output power of 5 watts. The 
antenna was connected as shown in Figure 2 
with a keyer switching from one antenna to the 
other. The first tests were carried out using 
the words “left” or “right” spoken in the micro¬ 
phone in accordance with the British method. 
In the first tests each antenna was about 220 
feet long following the British recommenda¬ 
tions. It was found, however, that shorter an¬ 
tennas could be used just as effectively and 
eventually antennas as short as 100 feet were 
used. 



Figure 3. Course of British beacon with voice 
modulation. Solid line for receiver having A VC; 
dotted line for receiver without A VC. 


Results with “Left-Right” Modulation 

Using antennas 220 ft long, tests were made 
on the width of the course as detected by a non¬ 
technical person who had been given some 
training in listening to the signals. A Navy 
RBZ receiver was used. The frequency used 
was 5.8 me. 















164 


PORTABLE RADIO ASSAULT BEACON 


Tests were made with A VC both on and off. 
The signal was sufficiently weak in most cases 
that the A VC did not have any large effect. 

The results, of which those shown in Figures 
3 and 4 are typical, indicate that the course 
defined in this manner of operation is very 
wide, ranging up to 8° for medium and long 



DISTANCE IN YARDS 

Figure 4. Width of course of British beacon. 

Receiver having A VC; voice modulation. 

distances and has very poor definition within 
the first 100 or 150 yd. Although these tests 
are highly subjective they do give an indication 
of the bluntness and inadequacies of this type 
of beacon compared with the requirements of 
the Army for this equipment. While the course 
could be followed more accurately by well- 
trained personnel carefully controlling the in¬ 
put and output levels of the receiver, it was 
apparent that the required accuracy could not 
be attained under normal operating conditions. 

One reason for the bluntness of the course 
according to this system was the difficulty of 
distinguishing differences in loudness between 
two dissimilar sounds occurring at different 
times and probably spoken with different 
degrees of loudness. It was expected, therefore, 
that considerably increased sensitivity would 
be obtained by using tone modulation. 

Results with A-N Modulation 

The use of tone modulation (1,000 cycles) 
produced a great improvement in the sharpness 
of the course. An important limiting factor 
appeared to be the key clicks. It was found 
difficult to compare accurately the loudness of 
the A and N signals in the presence of strong 


key clicks. These key clicks were frequently 
so much stronger than the tone signals that 
the observer had difficulty in eliminating from 
his mind the clicks and concentrating on the 
tone. The result was often very confusing ex¬ 
cept to the highly trained personnel. Much 
work was carried out on the elimination of key 
clicks. It was found that they could be elimi¬ 
nated or reduced to negligible amounts either 
by using the modified British system or by a 
relay of special design. 

After the key clicks had been substantially 
removed it was found that the speed of keying 
could be increased to 64 characters a minute 
and still be comfortably read by an untrained 
observer. Under those conditions the width of 
the course obtained was about ± 1° for a 1-db 
difference in signal level but aurally the course 
width was ±y 2 ° to trained observers. The site 
errors were considerably more than this, being 
of the order of 2°. The course could be fol¬ 
lowed with a receiver having A VC such as the 
SCR-536 and the Navy type RBZ. However, to 
attain good sharpness of the course with such 
receivers it was necessary to retract the anten¬ 
na, slightly detune the receiver, or otherwise 
maintain the receiver input sufficiently low to 
minimize the a-v-c action. This was particular¬ 
ly important at the closer distances, say the 
first 400 yd. It was also found important not 
to overload the receiver by permitting excessive 
input voltages, since this could cause an ap¬ 
parent reversal of the A and N quadrants when 
considerably off course. 

Factors Affecting Operation 

The system was studied in detail by ana¬ 
lyzing the effect of changing some of its para¬ 
meters and sources of error. The factors 
studied were: 

1. Length of antenna. 

2. Size of wire. 

3. Angle between the antennas. 

4. Height of antenna. 

5. Effect of obstructions. 

6. Effect of unequal currents. 

7. Effect of polarization. 

8. Effect of sky wave. 

9. Effect of weather. 






EXPERIMENTAL RESULTS 


165 


Figure 5 shows that not much is gained by 
making the antenna longer than 100 ft. 
Although measurements showed that sharper 
courses would be obtained with longer anten¬ 
nas, the course with the 100-ft antenna is only 
±y 2 ° wide which is sufficiently sharp. 



Figure 5. Field strength versus antenna length; 
W-110B wire on ground. 


The current distribution was measured to see 
if standing waves were appreciably reduced 
when using long antennas. The current in a 
ground antenna as long as 230 ft was a stand¬ 
ing wave having an attenuation of only 25 per 
cent per wavelength and a velocity of propa¬ 
gation of 0.8 the velocity of light. 

No appreciable variation of the velocity of 
propagation was noted when No. 18 enamel- 
covered wire, Army wire W-110B, No. 14 
stranded insulated wire or No. 14 solid copper 
rubber-covered wire was used. Although the 
desirability of using thinner wire to obtain a 
higher velocity of propagation was evident, 
experiments with such wire showed it to be im¬ 
practical for field use. 

Effect of Angle Between Antennas 

The 90° angle between the antennas of the 
British system is not essential. A system using 
a 60° angle was tried and the course obtained 
was fully equivalent to one using the 90° an¬ 
tennas except that there apparently was a 
slight amount of coupling between the two an¬ 
tennas in the 60° position so that the field from 
the energized antenna was diminished about 


3 per cent along the direction of the course. 
The only advantage in using a 90° angle is the 
greater ease of accurately laying out this angle. 

The degree of interaction of the two an¬ 
tennas is shown in Figure 6. Here one antenna 
was energized and its field measured at a point 
on a line making an angle of 45° thereto, 
while the unenergized antenna was swung from 
a position parallel to one perpendicular to the 
energized antenna. The maximum variation of 
the field under these conditions was about 8 
per cent. The field strength was 157 (in rela¬ 
tive units) when the unenergized antenna was 
entirely removed. When it was placed perpen- 



0 y ANGLE BETWEEN ANTENNAS 
IN DEGREES 

Figure 6. Field strength of energized antenna as 
function of angle of unenergized antenna. 

dicular to the energized antenna and raised 
from the ground to a height of 8 ft, the field 
strength varied only from 162 to 164. It may 
be concluded from these measurements, and 
from the many 360° surveys of the courses 
which were made, that a deleterious interaction 
of the antennas will not occur under any con¬ 
ditions likely to be encountered. 

Effect of Antenna Height 

Measurements indicated that raising the an¬ 
tenna from the ground to a height of 2 ft 
increased the field strength in the direction of 
maximum radiation only about 25 per cent, and 
that a height of 6 in. produced only a 10 per 
cent gain in field strength over the case of the 


% 











166 


PORTABLE RADIO ASSAULT BEACON 


antenna lying on the ground. Therefore it was 
decided to adopt the simple practice of stretch¬ 
ing the antenna along the ground unsupported. 

Effect of Obstructions 

For this type of beacon obstructions near the 
transmitter appear to have very little effect on 
site errors. Such obstructions as a full-scale 
model airplane 25 ft from one of the antennas 
did not appreciably affect the course. An auto¬ 
mobile placed within 5 ft of one of the anten¬ 
nas also had no measurable effect. A small 
building having electrical wiring about 20 ft 
from the apex of the antennas, small trees, a 
wooden tower 20 ft tall, and another wooden 
tower 40 ft tall, close to the antennas produced 
no noticeable deviations of the course. 

Obstructions near the receiver site were 
found to be quite important. Bends in the 
course resulting in errors as high as 4° were 
noted in the neighborhood of overhead power 
lines, and multiple courses were also noted in 
their neighborhood. In one case a course was 
traversed by a power line at an acute?angle at 
a distance of 400 yd along the course. There 
was also at this point on the course a building 
150 ft long and 40 ft high having electrical 
wiring. The course was bent about 4° in the 
vicinity of the power line. However, the course 
was found to resume approximately its correct 
direction about 100 yd beyond this line. The 
effect of these obstructions was undoubtedly 
increased by their location on high ground. 

There appeared to be some correlation be¬ 
tween hills and deviations of the course, but it 
was inconclusive because of the invariable 
presence of other site factors. The course was 
found to be straight through woods. A barbed 
wire fence across the course at an angle of 
about 90° did not appear to bend it measurably. 

Effect of Unequal Currents 

The course of the British beacon lies along 
the bisector of the angle between the antennas 
when the antenna currents are equal. When the 
current in one antenna is greater than the 
other, the course is deflected toward the other 
antenna. This effect is utilized in directing the 
course. Capacitors C 1 and C 2 in series with the 


antennas, shown in Figure 7, vary the currents 
in the antennas and thus determine the direc¬ 
tion of the course. These capacitors (maximum 
capacity 100 /x/xf) are varied differentially by a 
single control knob, which when turned to the 
right steers the course to the right, and when 



Figure 7. Modulator-keyer unit used with Brit¬ 
ish system. 

turned to the left steers the course to the left. 
The course may be steered ±20°. The course 
may also be steered by potentiometers placed in 
series with the antennas at their sending ends. 
It is considered more advantageous, however, to 
use capacitors because of the ease of attaining 
smooth operation, the avoidance of loss in the 
potentiometers, and the greater durability of 
condensers. 

Effect of Polarization 

The ground antennas in addition to radiating 
the desired vertically polarized field also radiate 
a horizontally polarized field. At a distance of 
200 yd, at an angle of 45°, with the antenna 







































EXPERIMENTAL RESULTS 


167 


3 ft above the ground, the vertically polarized 
field was 3.5 mv and the horizontally polarized 
field was 0.47 mv. The horizontally polarized 
field is capable of producing an error of 2° at 
100 yd and 1° at 200 yd. The polarization 
error at 300 yd is y^° and at greater distances 
it becomes negligible. These errors represent 
the maximum deviations which can be obtained 
with the SCR-536 receiver at heights of about 
2 ft above the ground and were determined by 
locating the apparent course with the receiver 
held horizontally and at right angles to the 
course, and then turning the receiver 180° in 
the horizontal plane and relocating the course. 
The difference between these two course deter¬ 
minations is called the horizontal polarization 
error. 

Sky-Wave Effect 

The course was tested at night and no sky- 
wave difficulties were noted. The critical fre¬ 
quency of the F layer at Washington at the 
time was lower than 5.5 me. Therefore, sky- 
wave propagation at this frequency could occur 
only by means of sporadic E-layer clouds. The 
radiation of the ground antennas in a vertically 
upward direction is so small that it seems very 
unlikely that an appreciable signal can ever be 
received via the ionosphere. The course showed 
the same accuracy at night as during the day. 
A loop direction finder placed on course at a 
distance of 2,500 yd showed no evidence of 
polarization error. There was also no evidence 
of fading at this distance, using a meter in the 
output circuit as an indicator. 

Effect of Weather 

This British type of beacon was tested under 
various weather conditions including heavy 
rainfall and while the ground was covered 
with a light snow. Such weather conditions 
did not appreciably shift the course. The course 
shift from a dry day to a succeeding rainy day 
measured at a distance of 700 yd was only %°, 
which is within the limits of experimental error. 

Modified British System 

The equipment for this system differs from 
that of the British system chiefly in the an¬ 


tennas. The antennas for the modified British 
system consist of a ground antenna at 90° to 
the desired course, and an antenna for radiat¬ 
ing vertically polarized waves of the proper 
phase with respect to the field from the first 
antenna. The second antenna may be the verti¬ 
cal rod antenna normally used with the SCR-284 
transmitter. Three antenna systems for the 
modified British beacon are shown in Figure 8. 
In Figure 8A the ground antenna at right 
angles to the course is a dipole type of Beverage 




COURSE AT RIGHT ANGLE TO 
60 FT GROUND ANTENNAS 

Figure 8. Arrangements of antennas with modi¬ 
fied British system. 

antenna. The antenna system in Figure 8B con¬ 
sists of a pair of single-ended Beverage-type 
antennas arranged in a straight line at right 
angles to the course. The course is obtained by 
























168 


PORTABLE RADIO ASSAULT BEACON 


reversing the current in the 90° antennas and 
thus switching the radiation pattern. 

The modulator and keyer unit for the modi¬ 
fied British beacon is similar to that of the 
British beacon. A circuit diagram is shown in 
Figure 9. 

The course of the modified British beacon can 
be shifted by suitable means. One such means 
which has been thoroughly tested is shown in 
Figure 8C and consists of a pair of antennas 
about 15 ft long arranged along the ground at 
right angles to the main ground antennas. 
These antennas are fed through a pair of uni- 
controlled capacitors which vary the current in 
these antennas. By means of these capacitors 
and a reversing switch (SW 2 in Figure 9) the 
course may be shifted about 7° to the right or 
left. 



Figure 9. Connections of antennas and relay with 
modified British system. Connections to micro¬ 
switch and hummer is as shown in Figure 7. 


Purpose of Modified British System 

The original purpose of the modified British 
beacon was to eliminate key clicks by perform¬ 
ing the required antenna switching in those 
antennas which have a null in the direction of 
the course, and thus prevent the switching from 
affecting the field in this direction. 

Another purpose was to eliminate those 
errors which are caused by factors that may 
cause the ground antennas of the British 
beacon to radiate fields of different intensities 
or different directional characteristics. Such 


factors are variations in height of the antennas, 
length, ground over which the antennas are 
stretched, and current distributions of the an¬ 
tennas. The independence of the modified sys¬ 
tem of these factors affecting the intensity of 
the radiated field arises from the fact that the 
course in this system is determined only by the 
location of the null of the radiation pattern and 
not by the absolute magnitude of the field or the 
shape of the radiation pattern. 

Factors Affecting Operation. The modified 
British system was studied by analyzing the 
several parameters and factors which might 
cause errors. 

Ground antennas of various lengths were 
studied, both of the dipole and single-ended 
types. These studies showed that a dipole 120 ft 
long, and a single-ended antenna 60 ft long had 
a sharp null and a high ratio of vertical to 
horizontal polarization. Figure 10 shows the 
radiation patterns of several types of antennas 
in the neighborhood of the nulls and exhibits 
the superiority of the 120-ft dipole at 5.5 me. 
Similar tests showed that 60-ft and 100-ft 
single-ended antennas had satisfactory radia¬ 
tion patterns. These curves, of course, are ap¬ 
plicable only to antennas utilizing the type of 



Figure 10. Radiation pattern in vicinity of nulls 
of several antenna types. 


wire and arranged at the heights above ground 
used in these tests. 

The effect of the size of wire and the height 
above ground has already been discussed, and 
the same findings which apply to a single-ended 
antenna also apply to a dipole type of ground 
antenna. 

The effects of obstructions both at the trans- 
































DESIGN OF SWITCHING RELAY 


169 


mitter and receiver sites are the same as for the 
unmodified British system. 

Tests made to determine the effect of unequal 
antenna currents on the position of the course 
showed that even a ratio as great as four to 
one between the currents in the main ground 
antennas had no effect on the location of the 
course. This is very important because unequal 
antenna currents always occur. 

The effect of horizontal polarization in caus¬ 
ing errors was found to be the same as in the 
British system. 

The height of the vertical rod antenna and the 
current in this antenna determine the range of 
the beacon and the sharpness of the course. With 
ground antennas 60 ft long a 9-ft vertical an¬ 
tenna gives approximately the same range and 
sharpness of course at a frequency of 5.5 me 
as the 45° antenna of the British beacon. 

117 DESIGN OF SWITCHING RELAY 

It was early recognized that one of the chief 
problems in perfecting the British beacon was 
that of eliminating key clicks. One solution was 
the use of the modified British system, the 
other was the design of a special relay. 

Many variations of relays were tested. The 
first attack on the problem was to study the 
cause of the clicks. It was found that the clicks 
resulted from the fact that during the time of 
commutation the r-f current in the antennas 
was reduced to a low value and that the in¬ 
tensity of the clicks was dependent on the time 
during which the current remained low. The 
designs were, therefore, directed toward a relay 
in which the actual time of commutation was 
reduced to a minimum. In commercial practice 
this result has been achieved by the use of very 
large magnets operating small moving parts. 
Relays weighing as much as 20 lb have been 
used for this purpose. In the present case such 
relays were not practicable. It was also found 
undesirable to develop a relay which would 
reduce the current in one antenna gradually 
before increasing the current in the other an¬ 
tenna. The effect of such a shift was to reduce 
the accuracy with which the course could be 
detected. To reduce the time of commutation it 
was realized that the moving contact would 
have to be made as light as possible and that 


commutation should take place only after this 
moving contact had reached a substantial veloc¬ 
ity. It was also found important that the 
inertia of the magnetic armature carrying the 
moving contact should cause as little delay on 
the action as possible. 

The first relay tried, while suitable for the 
voice-modulation type of British beacon, was 
found entirely unsuitable when tone modulation 
and A-N keying were applied. An antenna 
change-over relay having a 1/300-second time 
of throw was also found entirely inadequate. 
Snap-action switches such as a microswitch 
produced pronounced clicks. Two switching ar¬ 
rangements having make-before-break action 
gave no improvement. 

A double-contact switch making contact first 
through a resistor and then making contact 
directly was tested. The resistors were varied 
from 0 to 2,000 ohms and the least click ap¬ 
peared to occur when the resistors were entirely 
out of the circuit. This scheme appeared to have 
no promise. 

An Allied Control Company Type AK relay 
modified similarly to one used by the Naval Re¬ 
search Laboratory for the same type of beacon 
and loaned for study was tested and found to 
have a change-over time of approximately 
1/2,000 second. This relay would probably give 
very little key click. The result was obtained by 
an excessive current in the magnets causing 
them to become extremely hot. Since this type 
of relay was no longer manufactured, no work 
was done to incorporate it in the experimental 
models finally submitted. 

The relay shown in Figure 11 was built along 
the principles of having a light contact reaching 
a substantial velocity before commutation. It 
could be adjusted to have a change-over time of 
about 1/4,000 second. To avoid chatter it was 
necessary to dampen the vibration of the mov¬ 
able contacts by a packing of Airfoam sponge 
rubber. This relay was connected across a 12- 
volt battery which was switched from coil to 
coil by a microswitch. The current taken was 
1.5 amperes. The relay armature and movable 
contacts would normally occupy an inter¬ 
mediate rest position where they do not con¬ 
nect to either fixed contact during the throw 
of the microswitch, because the microswitch 
takes a comparatively long time to move from 



170 


PORTABLE RADIO ASSAULT BEACON 


one of its contacts to the other and the coils of 
the relay are unenergized for a sufficient time 
to allow the relay movable contacts to return to 
this center position. This condition was at first 
remedied by adjusting the magnetic circuit so 



fixed contacts, (4) movable contact, (5) armature 

stop screw, (6) adjustable bearing, (7) armature. 

that there was sufficient residual magnetism to 
hold the armature in one position until the coil 
in the other position was energized. Later the 
relay circuit was modified so that both coils are 
energized during the change-over of the micro¬ 
switch and hence the armature is positively held 
in a given position until the coil in that posi¬ 
tion is short-circuited. 

The relay is adjusted as follows. The fixed 
contacts are screwed in about 0.003 in. beyond 
the point where they just touch the movable 
contacts. During operation the movable con¬ 
tacts acquire such a high velocity just before 
making contact on the other side that the closed 
movable contact cannot, by virtue of its spring, 
remain closed. The best adjustment of the relay 
appears to be one in which the change-over time 
is about 0.00025 second. When so adjusted both 


antennas are simultaneously connected to the 
transmitter for the very brief time of about 
0.0001 second, but this appears to have no bad 
effects. Armature stop screws maintain a suffi¬ 
cient gap between the relay armature and pole 
pieces to prevent the armature from locking in 
one position. 

The relay can easily be set to have a change¬ 
over time of 0.0003 second. It showed no de¬ 
terioration of performance after 24 hours of 
continuous operation. These results require a 
positioning of the fixed contacts with a toler¬ 
ance of the order of 1/1,000 in. The models 
delivered to the Signal Corps were laboratory 
models and might not be able to maintain this 
degree of tolerance under hard field use. The 
Navy Department advised, however, that such 
a degree of tolerance can be maintained satis¬ 
factorily in the field. It is believed, therefore, 
that with suitable mechanical improvement in 
the design there will be no difficulty in having 
reliable relays for field operation. 

n s SETTING UP THE ANTENNAS 
IN THE FIELD 

Several methods of installing the antennas in 
the desired directions were developed and 
mechanical aids were delivered with the equip¬ 
ment. These included a magnetic compass with 
a pair of sights and aligning bars. The antennas 
can be aligned by placing them approximately 
in the correct directions and then by adjusting 
the currents so that “steering” occurs. These 
methods are described in greater detail in the 
contractor’s report on the project. 

EQUIPMENT DELIVERED 

Three sets of the final model of the equip¬ 
ment for the British beacon together with an 
antenna aligning device were delivered August 
5, 1944. Each set of equipment consisted of an 
instruction book and a carrying case containing 
two reels of antenna wire, two antenna ground 
stakes, cables, and a modulator-keyer unit. 

The modulator-keyer unit was housed in a 
waterproof box 10x7x9 in. The 1,000-cycle 
tone used for modulating the transmitter was 
generated by a General Radio Type 572-B hum¬ 
mer. The two coils of the keying relay were 















































































CONCLUSION 


171 


connected across a 12-volt battery. A cam- 
operated microswitch short-circuited the coils 
alternately. The cam was cut to produce inter¬ 
locked A and N characters and was rotated by a 
Haydon timing motor. The speed of the motor 
was adjusted to produce a keying rate of 64 
characters a minute. The motor is capable of 
keying up to 128 characters a minute by a suit¬ 
able adjustment of a resistor. A pair of lOO-^/xf 
straight-line frequency capacitors were used for 
setting the course within 20° of either side of 
the bisector of the angle between the antennas. 
The capacitors were coupled together so that 
they were rotatable by a single knob and were 
arranged so that they varied differentially but 
had equal capacitances at a midposition. 

The unit was powered by the 12-volt battery 
used for the transmitter. Its power consump¬ 
tion was 3.7 watts. It was capable of producing 
100 per cent modulation of the transmitter at 
the maximum power output of the transmitter. 
The weight of the complete unit in a heavy 
steel box was 18 lb. This weight could be re¬ 


duced to 9 lb by the substitution of an alu¬ 
minum box and a reduction in size of the unit. 

1110 CONCLUSION 

Of the three types of beacons studied experi¬ 
mentally the British and modified British bea¬ 
cons were found superior to the crossed-loop 
beacon. The experimental models of the British 
and modified British beacons gave substantially 
equal results. The modified British system is 
less subject to certain possible sources of error, 
but it is slightly more complicated if a steering 
adjustment is required. From a designer’s 
point of view the modified British system has 
also the advantage that the course can be 
readily broadened or sharpened at will by alter¬ 
ing the ratio of the currents in the vertical and 
horizontal antennas. As the course is broad¬ 
ened the signal intensity on the course is in¬ 
creased and vice versa. 

A comparison of the three types of beacons 
is presented in Table 1. 


Table 1 . Comparison of beacon systems. 



Crossed loops at 0° and 90° 
to required direction 

British beacon 

Modified British beacon 

Field strength at 1.5 miles 

12 /xv/m 

60 to 100 /xv/m 

60 to 100 /xv/m 

Width of course for ± 1 db 

± 2° with best installation 

±1° 

±1° 

Tuning procedure 

Least simple 

Simple 

Simple 

Steering of course 

Readily done 

Readily done 

Readily done 

Key-click elimination 

No special relay required 

Requires very rapid, well- 
adjusted relay 

Current 

Ratio Error 

No special relay required 

Unbalance between A and N currents 

Causes no error 

f 2° 

h 9° 

i 25° 

Causes no error 

Time required to install and tune 

More than 10 minutes 

Less than 10 minutes 

Less than 10 minutes 

Weight exclusive of transmitter 

Approx. 40 lb 

Approx. 30 lb 

Approx. 30 lb 

Polarization error 

Not measured 

±2° at 100 yd 
± 1° at 200 yd 

0° at 400 yd 

±2° at 100 yd 
± 1° at 200 yd 

0° at 400 yd 

Variation of ground under the 2 antennas 


Shift possible 

0° 

Unequal length of antennas 


Lengthening 220-ft an¬ 
tenna 16 ft caused shift 
of 3° at 100 yd, 5° at 
200 yd. 

Lengthening 90-ft antenna 
10 ft produced no meas¬ 
urable course shift. 
















Chapter 12 

U-H-F DIRECTION-FINDING ANTENNA STUDY 


Development of a direction-finding system 1 covering 
the range 140 to 600 me, providing instantaneous bear¬ 
ing indications for vertically polarized signals. Two 
wave collectors utilize a common receiver and indicator. 
One antenna consists of an Adcock system with output 
fed into a capacitive goniometer; the other antenna 
(for 300 to 600 me) is a rotating element in front of a 
reflector, the position of the antenna being synchronized 
with the CRO indicator. 

121 INTRODUCTION 

HE object of this project* was, briefly, to 
develop a d-f system operating in the u-h-f 
region of 140 to 600 me. It was hoped that 
much of the experience gained and the means 
developed in previous development programs 
on d-f systems for lower frequencies (1.5 to 30 
me) could be drawn upon in this project. It was 
found, however, that while the experience was 
useful, the methods employed in the lower-fre¬ 
quency systems so usefully could not be effective 
in the u-h-f region. 

12 2 PROBLEM DEVELOPMENT 

In the systems developed for the 1.5- to 30-mc 
region, aperiodic thermionic (cathode follower) 
coupling between the high impedance of the 
antennas and the low-impedance lines connect¬ 
ing the antennas to the receiver was quite 
effective in making it possible to space the re¬ 
ceiver at some distance from the antenna, and 
to provide an impedance match between an¬ 
tenna and line. An attempt to use this method 
in the higher-frequency region failed for the 
simple reason that tubes available at the time 
provided no more energy transfer when the 
tubes were operating normally than when they 
were cold. The major contribution to transfer 
existed in the capacitances within the tubes. 

It was found also that an inductive goniome¬ 
ter had to be abandoned because the transfer 
through it was largely capacitive and because 
of its low impedance. 

An electronic goniometer depended upon 

a Project C-80, Contract No. OEMsr-961, Federal 
Telephone and Radio Corporation. 


obtaining identical transfer characteristics 
through four separate tubes at all points of a 
modulation cycle. The difficulty of matching 
tubes made it impossible to obtain equality of 
transfer with modulation or to obtain adequate 
transfer of energy over the wide frequency 
range contemplated. This system had to be 
abandoned. Since the inductive goniometer be¬ 
haved better as a capacitive than as an induc¬ 
tive instrument, further work was concentrated 
on the development of a truly capacitive goni¬ 
ometer with the result that adequate transfer 
was obtained. The final model of the direction 
finder employed such a unit. 

Using the design principle which had previ¬ 
ously proved adequate in the frequency range 
1.5 to 30 me, a ground plane carrying four 
monopole antennas, acting in pairs to give 
crossed figure-eight diagrams, was constructed. 
Since the thermionic coupling means were 
proved to be unsatisfactory the antennas were 
terminated resistively. 

The receiver research for this project passed 
through three stages. The preliminary receiver 
was constructed having one r-f stage, an oscil¬ 
lator and mixer each tuned by means of coaxial 
lines the movable elements of which were 
ganged to a single control. The r-f input of this 
receiver was applied through a 50-ohm coaxial 
transmission line. 

The first modification was alteration of the 
input circuit to obtain balanced input. The sec¬ 
ond and final modification consisted of a com¬ 
plete mechanical redesign to avoid the necessity 
for having the cumbersome tuning method of 
the previous models. 

12.3 SYSTEM EXPERIMENTS 

The first experiments with the complete d-f 
system were conducted using a capacitive 
goniometer mounted on the Type A indicator 13 

b The Type A indicator utilizes a cathode-ray tube 
and circular trace. The trace is obtained by mechan¬ 
ically rotating magnetic deflection coils about the neck 
of the tube. The rectified received signal is fed into the 
coils to change the circular trace to the typical pro¬ 
peller-shaped direction pattern. 



172 




SYSTEM EXPERIMENTS 


173 


in place of the normally used low-frequency 
goniometer. The antenna output was connected 
to two balanced transmission lines, one for each 
antenna pair, and applied to the two sets of 
stator plates of the capacitive goniometer. The 
first system tested was composed of the most 
satisfactory elements determined from the pre¬ 
liminary research. The monopole antennas were 
resistively terminated. Use of two 40-ft bal¬ 
anced transmission lines enabled the collector 
system to be placed at a distance from the re¬ 
ceiving and indicating equipment. It was im¬ 
mediately determined that very poor nulls were 
secured, that the nulls were not reciprocal and 
that the overall sensitivity of the system was 
very poor. A modification program was insti¬ 
tuted leading to the following changes: 



IOO 200 300 400 500 600 

FREQUENCY IN MC 


Figure 1 . Characteristics of capacitive 
goniometer. 

The first capacitive goniometer used induc¬ 
tive means for coupling the rotor plates to the 
receiver input. This output transformer gave 
very poor transfer and immediate steps were 
taken to increase the efficiency. One goniome¬ 
ter was constructed in which the inductive out¬ 
put device was replaced by slip rings. A con¬ 
siderable gain in transfer was apparent but 
due to the use of a continuously rotated gon¬ 
iometer, the slip rings required frequent main¬ 
tenance. A capacitive-output coupling system 


was then constructed which gave reasonably 
good transfer characteristics. The character¬ 
istics of the capacitive goniometer are shown in 
Figures 1 and 2. 



Figure 2. Transfer characteristic of goniometer. 


One of the principal reasons for poor nulls 
and for nonreciprocal bearings was the fact 
that the transmission lines connecting the an¬ 
tennas of one pair were not properly shielded 



Figure 3. Direction-finder receiver in which tun¬ 
ing is accomplished by rotating a drum which sup¬ 
ports entire r-f and converter section, varying 
effective length of three coaxial lines and one 
quarter-wave open wire line (local oscillator). 
Four-stage 15-mc i-f amplifier with gain of 25,000 
and band width of 1 me follows the converter. 


and that there was direct pickup on these lines. 
It was found necessary to shield very 
thoroughly the transmission lines themselves, 
to provide additional shielding at the crossover 













174 


U-H-F DIRECTION-FINDING ANTENNA STUDY 


point, and to enclose the entire transmission 
line system within an additional shield. 

After the shielding means had been employed 
poor nulls were still observed over a consider- 


the balanced output of the goniometer to match 
the unbalanced coaxial transmission line. This 
modification not only enabled the distance be¬ 
tween the antenna and the receiver to be in- 



Figure 4. Inside view of 300- to 600-mc antenna system. 


able portion of the frequency range and large 
errors were introduced because of unbalance 
in the 40-ft transmission lines between the col¬ 
lector system and the goniometer and because 
of the differences in electrical length of these 
two lines. Therefore the capacitive goniometer 
was moved into close proximity with the an¬ 
tenna system. A further improvement was 
effected when the output of the goniometer was 
fed directly into a “balance box” transforming 


creased but in addition eliminated a great many 
of the poor nulls which had previously been 
observed. 

At the same time it was possible to begin 
tests with an improved model of the receiver 
(Figure 3) having square cross-section trans¬ 
mission lines as the tuning elements coiled on a 
drum which was rotated by the dial mechanism. 
This receiver used lighthouse tubes throughout 
and was more sensitive than previous models. 





FINAL DESIGNS 


175 


Some difficulty was encountered because of the 
use of sliding short circuits as tuning elements 
of the transmission lines. 

To localize any difficulties which might be 
contributing to errors or to poor operation, an 
extensive series of tests was instituted on the 
separate components of the collector system to 
determine the impedance characteristics of each 
over the frequency range and, if possible, to 
discover design criteria. The results obtained 
showed that the antennas would be extremely 



Figure 5. Calculated field pattern of two vertical 
monopoles mounted in front of infinite reflector. 
Monopoles are fed 180° out of phase with each 
other. At mean frequency, half-space between an¬ 
tennas was 5 in., and spacing to reflector, 5 in. 
Ratio of field intensity at 600 me to that at 100 me 
is almost 24/1 for equal fields set up by antenna. 

difficult to match to a transmission line and 
indicated why the capacitive goniometer ceases 
to function at about 300 me and in general 
show the difficulties which were encountered in 


an attempt to make a monopole system of this 
type operate over such a wide frequency range 
without drastic changes in design. 

12 4 FINAL DESIGNS 

For several months studies had been in 
progress on a collector system constituted by 
two oppositely connected dipoles spaced from 
each other and in front of a reflecting plane 
surface. To obtain automatic instantaneous 
indication from a system of this type, a collec¬ 
tor was constructed as illustrated in Figure 4. 
This rotating collector was driven by a large 
induction motor and the instantaneous position 
of the collector was repeated through a selsyn 
system so as to be shown on a CRO screen. The 
calculated directional pattern of the collector, 
the measured pattern and the resulting indica¬ 
tion are shown in Figures 5 and 6. The system 
operated with satisfactory results between 300 



Figure 6. Measured field pattern (left) and result¬ 
ing indicator pattern of monopole-reflector system. 


and 600 me, thereby supplementing the per¬ 
formance which had been obtained using the 
fixed monopole system and the capacitive 
goniometer. 

As a final step in the development, the low- 
frequency system (140 to 300 me), consisting 
of the five monopole antennas and the capaci¬ 
tive goniometer, and a high-frequency system 
(300 to 600 me), consisting of the rotating an¬ 
tenna, were incorporated for use with a single 
control unit consisting of the receiver, an indi¬ 
cator and the necessary power supplies. 


























176 


U-H-F DIRECTION-FINDING ANTENNA STUDY 


12.5 


PERFORMANCE 


Figure 7 shows that the sense performance 
of the 140- to 300-mc monopole system is not 
adequate. A considerable amount of redesign 
and further development would be necessary to 
obtain results which would permit a produc¬ 
tion-type system to be built. Figure 8 shows the 
directional accuracy of the monopole antenna 
collector system with the capacitive goniome¬ 
ter in the frequency range 140 to 300 me. This 


and 600 me, nulls are always sharp and in every 
way the operation of this system is much more 
satisfactory than that of the fixed-monopole 
system. 











p 





140 MC 



P - POOR 

0 - ZERO 

1 - R- REVERSE 

I = - 140 MC 

-20 

P 






160 MC 

P 












“ 150 MC - 

+ 20 

0 

-20 



P 










bd 

f- - 

170 MC 



G 

Z F 

2 g 


2 F 

* 8 


G - GOOD 

F ~ FAIR 


170 MC 


180 MC - 


ZERO SENSE 


190 MC - 


220 MC 


230 MC 



45 


90 135 180 225 270 

BEARING IN DEGREES 


+ 40 
+ 2 0 


315 


Figure 7. Sense operation characteristics of 140- 
to 300-mc Adcock. 


performance could also be considerably im¬ 
proved. 

The rotating antenna system is not subject 
to the same type of errors as the monopole 
system. The accuracy is indicated as ±3° in all 
tests made. No sense ambiguity is possible 
with this type of collector system. Between 300 




45 90 135 ISO 225 270 315 

TRUE BEARING IN DEGREES 

Figure 8. Bearing accuracy of 140- to 300-mc 
Adcock antenna. 






































































































































































































LOW FREQUENCY ANTENNA SYSTEM RECEIVING SYSTEM 


PERFORMANCE 


177 


r 



L 


J 


l_ 


J 



Figure 9. Block diagram of 140- to 600-mc direction-finder system. 



















































178 


U-H-F DIRECTION-FINDING ANTENNA STUDY 


12 6 ELECTRICAL CIRCUIT THEORY 

The entire system as finally developed con¬ 
sists of three major units: Band I (140 to 300 
me) wave collector and goniometer; Band II 
(300 to 600 me) wave collector; receiving and 
indicating unit for remote operation. (See 
Figure 9.) 



Figure 10. Low-frequency (140- to 300-mc) an¬ 
tenna system. 



TO RECEIVER 


Figure 11. Elementary schematic of low-fre¬ 
quency portion of direction-finder system. 


diagram is shown in Figure 11 where only one 
Adcock pair is indicated for the sake of sim¬ 
plicity. The polar diagram of this array is a 
figure eight (Figure 12). 



Figure 12. Figure eight pattern of Adcock an¬ 
tenna. 

This pattern follows a cosine function. If the 
antenna system were rotated by hand only one 
pair of antennas would be required, the position 
of the nulls indicating the direction of the re¬ 
ceived signal. For instantaneous indication the 
capacitive goniometer scans the output of two 
pairs of Adcocks (four antennas). 


Band I Wave Collector 

As shown in Figure 10 the 140- to 300-mc 
Adcock wave collector consists of five vertical 
monopoles mounted on insulators over a copper 
ground plate. Directly below the plate and 
mounted in a wooden protective box are the 
capacitive goniometer, the driving motor, and 
the selsyn generator. The entire system block 


N-S STATOR E-W STATOR 



Figure 13. Elements of capacitive goniometer. 




























































ELECTRICAL CIRCUIT THEORY 


179 


Goniometer 

The rotor of the goniometer consists of two 
semicircular plates, A and B, in Figure 13, 
insulated from each other. The two pairs of 
stators are identical except that one is oriented 
90° with respect to the other. One output ring 
is connected solidly to rotor A, while the other 
is connected to rotor B. These are rotated inside 
two fixed rings to provide capacitive coupling 
to the rotor output. (See Figure 14.) 

The stator plates are so shaped that the 
capacitive coupling between rotor and indi¬ 
vidual stators varies as a cosine function with 
rotation. For example, assume both pairs of 
antennas connected to both stators and the 
signal being received in the N-S direction. The 


away from the previous ones because of the 
positioning of the E-W stator. In this manner 
the goniometer will indicate bearings of signals 
in line with the antennas. 

For the case where the signal direction is not 
in line with either array, assume the signal is 
received along the line o-b (Figure 12). This 
means that there will be o-a volts delivered to 
stator E-W, and o-b volts delivered to stator 
N-S. Therefore, across stator N-S there will be 
a voltage 

e cos 0, 

where e ~ voltage (o-c ), and across stator E-W, 
there will be a voltage 

e sin 0. 


INSULATED BEARING 


INPUT D.F. 
STATOR PLATES 


INPUT D.F 
STATOR PLATES 


BALL BEARING 



Figure 14. Photograph of goniometer. 


signal will be in the null of the E-W antennas 
so that no voltage appears across the E-W 
stator to be picked up by the rotor. For the N-S 
stator, as the rotor is turned slowly, the output 
will vary from a maximum when the plates A 
and B are parallel to the stator to a minimum 
of zero when the rotors are at right angles to 
the stators. Thus two nulls are produced 180° 
apart. 

Similarly, if the signal is in the direction of 
the E-W antenna, two nulls will again be pro¬ 
duced 180° apart, except that they will be 90° 


For any rotor position, there will be a voltage 
coupled from stator E-W to the rotor propor¬ 
tional to e sin 0 and equal to 
K [e sin 0]. 

Similarly, the voltage coupled from stator 
N-S will be equal to 

K[e cos 0]. 

If the rotor is lined up for maximum cou¬ 
pling to stator E-W, and then rotated through 
an angle f3, the voltage across it will be 

K[e sin 0] [cos /?]. 
















180 


U-H-F DIRECTION-FINDING ANTENNA STUDY 


The voltage coupled into the rotor from stator 
N-S will vary from zero to maximum as f3 is 
increased and the resultant voltage will be 

K[e cos 0 ] [sin /?]. 

There will be some position for a rotation of 
(3 degrees where 


K [e cos 0] [sin /3 ] 



A 



B 


Figure 15. A shows double null pattern from Ad¬ 
cock antenna; B shows pattern with sense antenna 
added to Adcock. 


must be fed in phase to the goniometer output 
to produce a cardioid pattern. Since the Adcock 
monopoles are cross-connected, an analysis of 
the voltage vectors will show that the sense 
antenna output is 90° out of phase with the 
Adcock antenna output. To shift it 90°, the out¬ 
put of the sense antenna is fed through two 
unequal transmission lines, through unbalance- 
to-unbalance converters, and mixed through a 
relay with the goniometer output. The line 



TO RECEIVER 

Figure 17. Block diagram of high-frequency an¬ 
tenna system. 


At this point, the voltages will cancel in the 
rotor for zero output in a null position. Solving 
the above equation, it will be found that 

0 = ~ P 

indicating that the position of the rotor indi¬ 
cates the actual bearing. This would produce a 
double null pattern as shown in Figure 15A. To 
establish sense, the output of the sense antenna 



Figure 16. High-frequency (300- to 600-mc) ro¬ 
tating antenna. 


lengths are so proportioned as to produce a 90° 
phase shift over the band. When the sense 
antenna is connected in the circuit the sense 
pattern would theoretically appear as shown in 
Figure 15B where the pattern indicates the di¬ 
rection of the bearing. Because of difficulties 
with balance in two coaxial lines, the goniometer 
output is fed into a balance-to-unbalance con¬ 
verter and then via a single coaxial line to the 
receiver. 



Figure 18. Antenna pattern from crossed high- 
frequency monopoles. 

The goniometer is rotated by means of a 
motor which also turns a selsyn generator. This 
selsyn generator is used to drive a selsyn motor 
in the indicating unit. 

Band II Wave Collector 

As shown in Figure 16, the 300- to 600-mc 
collector consists of a pair of vertical monopoles 



















ELECTRICAL CIRCUIT THEORY 


181 


in front of a reflector and rotated over a ground 
screen. A block diagram of the system is shown 
in Figure 17. 

The entire system is mounted in such a man¬ 
ner that the cylindrical-shaped balance box 
serves as a shaft for rotation. The output is 
taken off via a fixed line about which the bal¬ 
ance box rotates so that no rubbing contacts are 
used. The monopoles are cross-connected at the 
balance box so that the antenna patterns are 
approximately as shown in Figure 18. 

Since a sharp null is produced in the direc¬ 
tion of the received signal, the system is uni¬ 
directional and requires no sense, as in the 
Band I collector. As the collector is rotated, the 
operator will first find the signal over a rather 


Receiving Unit 

The receiving unit (Figure 19) consists of a 
140- to 600-mc tuned line receiver, a d-c ampli¬ 
fier, and switching circuits for operating Band 
I and Band II collectors. Tuning the receiver is 
accomplished by varying the length of a cir¬ 
cular transmission line by means of shorting 
bars. Receiver input is single-ended and is fixed 
at 90 ohms. The i-f channel is straightforward 
and has a bandwidth of 1,000 kc for passage of 
pulses. 

Motor switching circuits are interlocked so 
that only one system can be operated at a time. 
In operation only the r-f cable need be changed 
for a band change. 



broad lobe, pass through a sharp null, continue 
over another broad lobe of reception and then 
pass through approximately 180° of null. The 
collector is driven by a variable-speed motor 
which also drives a selsyn generator for syn¬ 
chronization with the indicator. 


Indicator Unit 

The Type B indicator (Figure 20) with two 
selsyn motors for driving, and speed control for 
antenna systems are mounted on the power- 
supply chassis. The Type B indicator consists 




























































182 


U-H-F DIRECTION-FINDING ANTENNA STUDY 


of a strip of alternate thin laminations of cop¬ 
per and polystyrene. The projecting ends of the 
laminations are ground to a flat surface and a 
uniform resistance strip is compressed on one 
side. This produces a commutator with a large 
number of equal resistance steps between bars. 
The strip is rotated by a pair of selsyn motors 
to produce the voltage needed to generate a 
circular trace in the cathode-ray oscilloscope. 

If a current is sent through the strip a sinu¬ 
soidal voltage will be generated across a pair of 
brushes mounted along a line perpendicular to 



Figure 20. Elements of Type B strip indicator. 

the rotational axis and equidistant from it. By 
mounting another pair of brushes at right 
angles to the first pair, two sinusoidal voltages 
are obtained with 90° phase difference. These 
voltages applied to the deflecting plates of the 
CRO tube produce a circular trace when the 
spot moves at constant velocity. 

By supplying the Type B strip current from 


the plate of a d-c amplifier following the re¬ 
ceiver detector, the receiver output can be made 
to vary the shape of the circle for an indication. 

When the receiver output is zero at the null 
(0°) the plate current in the d-c amplifier will 
be maximum and the spot will be at the outside 
of the circle. As the goniometer scans from 0° 
to 90°, the receiver output will increase to 
maximum, biasing the d-c amplifier until cutoff 
is reached, and no voltage will appear across 
the strip. Thus the spot will approach the 
center, rapidly at first because of the sharpness 
of null and then more gradually. 

For sense operation the same principles 
apply except that the cardioid pattern resultant 
produces a pattern with one broad null. 

To place the cardioid pointing in the same 
direction as the d-f pattern, it is necessary to 
turn the pattern by 90° on the cathode-ray tube. 
This is done by means of a four-pole double¬ 
throw relay which switches each brush to the 
adjacent cathode-ray tube deflection plate. Posi¬ 
tioning of the circle is effected by magnetic de¬ 
flection coils placed about the neck of the 
cathode-ray tube and operated from the low- 
voltage supply. 

Circle diameter is varied by cathode bias con¬ 
trol of the d-c amplifier. Speed control is incor¬ 
porated into the wave collector motor since it 
is necessary to bring the selsyn motors up to 
speed gradually. 



















Chapter 13 


LOCATING TANKS BY RADIO 


Problem of locating the position of friendly tanks 
with respect to a fixed station to an accuracy of ± 50 
yd in 5 miles using existing Signal Corps tank equip¬ 
ment by an audio-phase-measurement method. Investi¬ 
gation of the characteristics of existing tank equip¬ 
ment indicated that inherent phase instability would 
make impossible location of tanks to the requited 
degree of precision. 1 * 2 

131 INTRODUCTION 

T he basic idea involved in these two proj¬ 
ects' 1 was to place a constant audio tone on 
the carrier of a standard communication trans¬ 
mitter at a locator station. This signal would 
be received by the tank and the tone would be 
retransmitted by the tank on another radio 
frequency. Assuming constant time delay, or 
phase shift, through the transmission and re¬ 
ception networks, the measured phase shift 
in the audio tone as measured at the locator 
station would be a measure of the distance 
between the tank and the locator station. The 
location of the tank or group of tanks would 
be accomplished by a triangulation process. 

One requirement established was that exist¬ 
ing equipment be employed in these projects. 
Therefore, although the method for locating 
tanks by radio was considered basically work¬ 
able, whether the scheme would be successful 
would depend entirely upon the following two 
major factors: 

1. The accuracy with which the phase mea¬ 
surement could be made. 

2. The stability of the phase shift through 
the tank equipment under normal operating 
conditions. 

Tests, therefore, were made by the two con¬ 
tractors on the phase stability of two existing 
pieces of radio equipment, the SCR-506 in the 
2- to 41^-mc region and the SCR-508 in the 
20- to 30-mc region. 


a Project C-60, Contract No. OEMsr-787, Bell Tele¬ 
phone Laboratories; and Project C-61, Contract No. 
OEMsr-737, General Electric Co. 


13 2 TEST RESULTS 

13,21 Tests on SCR-506 

To measure the distance of the tank within 
d=50 yd at 5 miles requires an accuracy of 
0.57 per cent. Using an audio frequency of 
2,000 cycles per second would result in a phase 
shift of 38.7° for a 5-mile spacing between 
tank and fixed station. To measure this phase 
shift to an accuracy of 0.57 per cent would 
require that measurement to 0.22° would be 
necessary. 

Measurements on the SCR-506 (Project C- 
61) were accurate to about ±0.25°. It was 
found that the slope of the tuning curve of 
this receiver was about 1° per kc off tune. 
Using the beat-frequency method, this error 
might be held to 0.05°. Even when the local 
oscillator was adjusted by the zero beat method, 
a change of phase shift of 0.07° occurred per 
degree centigrade rise in ambient temperature. 
The average slope of the curve of phase shift 
versus percentage modulation was about 0.12° 
for a 1 per cent change in modulation. With 
the automatic volume control disconnected 
(manual gain control condition) severe phase 
shifts with changes in signal level occurred. 
In the a-v-c condition, no measurable phase 
shift occurred with a signal level change of 
10 to 1. A signal level of at least 1,000 /xv 
would be required for reliable readings. In the 
operating region, the slope of the volume con¬ 
trol setting curve showed a phase shift of ap¬ 
proximately 0.12° per degree rotation of the 
volume-control knob. 

In light of these measurements, it was de¬ 
cided that the instability in phase shift through 
the receiver alone under normal service con¬ 
ditions would make the audio phase shift 
method of measuring distance impractical. 

13 2 2 Tests on SCR-508 

Using the measurement of time as a concept 
of the measurement of distance, phase shift 


183 



184 


LOCATING TANKS BY RADIO 


would have to be measured within time inter¬ 
vals of 0.306 [x sec to accomplish the accuracy 
of 0.57 per cent required. Direction would 
have to be measured within 19.5 minutes. 

It was found that the inherent variations of 
phase shift in the SCR-508 (Project C-60), if 
uncontrolled and uncalibrated in the mobile 
tank at the time of measurement, would pro¬ 
hibit measurements within ±8 /zsec. For ex¬ 
ample, variations in temperature between 
—20 C and +50 C together with changes in 
humidity would produce oscillator drift as 
much as 50 kc. This alone makes it impos¬ 
sible to meet an accuracy of ± 5.4 /xsec or 0.5 
mile in 5 miles. Through inability of the re¬ 
ceiver’s pushbutton tuner to be reset at the 
same oscillator frequency by merely selecting 
the same pushbutton would produce an error 
of ±2.7 /xsec. These figures do not include 
the inherent differences between tank equip¬ 
ments of the same model numbers. 

So far as the SCR-508 was concerned, it 
was apparent that the a-f phase-shift measure¬ 
ment method of measuring distance could not 
be more accurate than about 25 per cent, or to 
within 2,200 yd of 5 miles instead of the re¬ 
quired 50 yd. 

Modification to Improve Accuracy 

Variations in the receiver’s pushbutton tun¬ 
ers gave errors in excess of 10° at 10,000 
cycles. To offset these errors together with 
the 50-kc oscillator drift would require a crys¬ 
tal-controlled oscillator in the receiver. 

By a technique which called for the trans¬ 
mission of two audio frequencies somewhat 
greater accuracy could be attained since dis¬ 
tance would now be determined by the total 
measured phase difference between the two 
frequencies rather than the absolute value of 
phase at either frequency. Assuming that the 
phase-shifting networks were individually ad¬ 
justed for each mobile tank installation and 
that each receiver had the necessary crystal 
oscillator modifications, an accuracy of ap¬ 


proximately 12 per cent or 1,000 yd in 5 miles 
would be possible. 

Elimination of all audio amplification, using 
the i-f voltage to drive the transmitter, and 
by making other changes to the receiver (such 
as changing the intermediate frequency) might 
result in a phase-shift time in the mobile unit 
of approximately 4.0 /xsec. The amplitude sta¬ 
bility of the SCR-508 equipments will not per¬ 
mit the adjustment of two voltages required 
for measuring phase by the sum-and-difference 
method to closer than 0.2 db with the result 
that an accuracy of measurement of 250 yd in 
5 miles is about the limit possible with the 
modified receiver suggested. 

Required Measurement Accuracy. A 1° ac¬ 
curacy when measuring phase will permit ap¬ 
parent errors of 90 yd in 5 miles at a modula¬ 
tion frequency of 5 kc. If the modulation fre¬ 
quency is 15 kc this 1° accuracy of measuring 
phase shift will permit measurements to within 
30 yd at 5 miles. Therefore any phase shift 
method must have an accuracy of 1° or better, 
particularly if any latitude is to be left for 
variations at the mobile tank. Such methods 
are known but they are not of such nature that 
they could be used in the field easily. Labora¬ 
tory methods exist which will provide an ac¬ 
curacy of 0.2°. 

13 2 3 Simplified Radar Method 

The final report on Project C-60 1 proposes 
a modified radar method in which the tank car¬ 
ries a repeater made up of a 90-db voltage am¬ 
plifier and a 50-watt 50-mc power amplifier. 
The fixed station transmits pulses of 1 ^sec 
duration. With a receiver band width of ap¬ 
proximately ±3 me an accuracy well within 
the prescribed 50 yd independent of the dis¬ 
tance measured is estimated. The tank unit 
being a repeater requires no tuning or crystal 
and could be readily adapted to equipment al¬ 
ready in the field. Thus it would be much 
simpler than the proposed a-f phase-shift 
method. 




Chapter 14 


U-H-F FRIENDLY AIRCRAFT LOCATOR 


A d-f system providing automatic and continuous 
indication of bearings of signals in the region 100 to 
250 me, with arrangements for remote display of the 
azimuthal distribution of received signals. 1 


them for tracing out the necessary patterns on 
the CRO screens for indicating the bearing. A 
block diagram of the apparatus is shown in 
Figure 1. 


141 INTRODUCTION 

A t the time this project a was started radar 
l was in its infancy but it was realized that 
means for identifying friendly aircraft were 
needed. It was believed that d-f methods giving 
the azimuth of the target would be useful, par¬ 
ticularly if two or more d-f stations could use 
triangulation techniques. 

Means were developed for taking bearings in 
a matter of about five seconds with an accuracy 
of approximately ±8° and for transmitting the 
bearing data over conventional telephone facili¬ 
ties. The system was operable on c-w, i-c-w, and 
pulse signals. Bearings were taken at nearly 
maximum signal level rather than at a null, and 
could be taken on two or more signals at the 
same time provided the bearings were not too 
close together in azimuth. There was no am¬ 
biguity regarding sense. The visual indicator 
(CRO) traced a polar diagram of the received 
signal, and an electrical marker system put 
markers on the CRO screen at 1° intervals. 

142 THE OVERALL SYSTEM 



Principal components of this direction finder 
consisted of a rotating directional and non- 
directional antenna assembly, a u-h-f receiver 
having two channels for amplifying the respec¬ 
tive antenna signals, line transmitter goniome¬ 
ter units to prepare the signals from the d-f 
channel of the receiver and signals from the 
goniometers which indicate antenna orienta¬ 
tion for transmission over a telephone line, and 
a line receiver indicator unit which obtained 
signals from the line transmitter (directly in 
the case of the monitor and over the telephone 
line in case of remote operation) and prepared 


a Project C-12, Contract No. NDCrc-193, Hazeltine 
Electronics Corp. 


Figure 1. Block diagram of aircraft locator. 

143 ANTENNAS 

The antennas provided (1) a directional lobe 
of the received signal for bearing purposes and 
(2) a nondirectional signal for audible moni¬ 
toring and for a-v-c purposes. The directional 
antenna consisted of a conical dipole A/4 from 
the origin of a parabolic reflector; as the an¬ 
tenna rotated, a varying signal was induced in 
the antenna producing a single-lobe pattern 
with the axis pointing toward the received sig¬ 
nal. The antenna rotated at 100 rpm producing 
rapidly recurrent patterns so that continuous 
indication of the received signal took place. 


185 


























186 


U-H-F FRIENDLY AIRCRAFT LOCATOR 


The nondirectional antenna consisted of a 
single-cone monopole and an artificial ground 
mounted on top of the parabolic reflector. 

Sheathed transmission lines matched to the 
characteristic impedances of the antennas con- 


and furnishing output to a headset for aural 
monitoring and a voltage for automatically con¬ 
trolling the volume of both channels. A com¬ 
mon heterodyne oscillator served both purposes. 

Provisions were made for handling either c-w 



DF SIGNAL FROM 
UHF RECEIVER 


Figure 2. Block diagram of line transmitter showing several frequencies employed to transmit information 
to telephone lines. 

nected the antennas to the inputs of the re¬ 
ceiver. Rotating transformers designed as 
band-pass filters coupled the transmission lines 
from the rotating structure to the receiver. 

14.4 U-H-F RECEIVER 

The equipment was designed for two fre¬ 
quency ranges, 100 to 156 me and 156 to 250 
me. The receivers were superheterodynes with 
two separate channels, one modifying the sig¬ 
nals from the directional antenna and applying 
its output to the line transmitter, the other 
amplifying the nondirectional antenna signal 


or pulse signals; the i-f bandwidth could be set 
at 250 kc for c-w or at 3.5 me for pulses by 
switching transformers in four of the six i-f 
stages. The circuits were designed to handle 
pulses having a repetition rate of from 625 to 
4,000 per second and having a pulse width of 
from 1 to 15 p,sec. 

145 LINE TRANSMITTER 

GONIOMETER UNIT 

This unit obtained electrical information as 
to the exact and instantaneous position of the 
antenna and prepared these signals and the 

































































LINE RECEIVER AND CATHODE-RAY INDICATOR UNITS 


187 


output signal of the d-f channel of the receiver 
for transmission over the telephone line. Three 
goniometer assemblies were required, each 
geared through a differential to the rotating 
antenna. One goniometer rotated at the same 
speed as the antenna, providing X and Y com¬ 
ponents for tracing out the angular position of 
the antenna on the quadrant-indicating CR 


A total of seven audio signals was used to 
transmit this information to the line receiver. 
Frequencies and amplitudes of these signals 
were proportioned to produce the least amount 
of distortion and crosstalk in the telephone 
lines. A block diagram of the line transmitter 
showing several frequencies employed to trans¬ 
mit information is given in Figure 2. 





50^ CTI FfTHP A, 


>J\J O C. L. C. \j 1 Un Or 

DETECTOR UNIT 



(HORIZONTAL DEFLECTION) 






GRID 



FROM TELEPHONE 
LINE OR FROM 
LINE TRANSMITTER 
< WHEN USED AS 
MONITOR) 



Figure 3. Block diagram of line receiver and CRO indicator tube. 


tube, another rotated at four times the antenna 
speed and produced the components for tracing 
out the angular position of the antenna on the 
bearing-indicator CR tube, and the third goni¬ 
ometer rotated at 12 times the antenna speed 
for producing phase-modulated signals for elec¬ 
trical markers on the bearing-indicator cathode- 
ray tube. 


146 LINE RECEIVER AND CATHODE- 
RAY INDICATOR UNITS 

The line receiver (Figure 3) separated and 
prepared the signals received from the line- 
transmitting unit as to antenna location and 
d-f signal output for tracing the polar dia¬ 
grams on cathode-ray tubes, one for indicat- 























































188 


U-H-F FRIENDLY AIRCRAFT LOCATOR 


mg the directional lobe of the received signal 
for approximately locating the signal and an¬ 
other bearing-indicator cathode-ray tube hav¬ 
ing an expanded scale such that one complete 
revolution on the screen was equivalent to 90° 
of antenna rotation. On this tube a portion of 
the directional lobe was also traced out. 

Because the lobe itself was not sharp enough 
to indicate the bearing accurately, circuits were 
provided for switching a deflection field at a 
rapid rate so that two intersecting patterns 
appeared on the face of the tube. The point of 
intersection of these patterns enabled the op¬ 
erator to determine azimuth accurately. 


two signals of the same frequency. The sig¬ 
nals that were blocked out were shown elec¬ 
trically on the quadrant indicator tube by dot¬ 
ted traces. Only solid traces shown on the 
quadrant tube were reproduced on the bearing- 
indicator tube. A sample indicator pattern is 
given in Figure 5. 




Figure 5. Sample pattern obtained when taking 
bearings. 


Figure 4. View of 156- to 250-mc direction finder 
with conical antennas. 

Electrical markers at 1° intervals with dis¬ 
tinguishing marks at 5° and 15° intervals were 
provided. Transient traces were blocked out 
so that clear patterns were obtainable. A quad¬ 
rant blockout control blocked out any two 
quadrants, a useful feature when examining 


Provisions were made for equalizing the tele¬ 
phone circuits. A pre-emphasis control was 
available for use where a Signal Corps line 
was connected between the d-f station and a 
telephone line, enabling the input to the Signal 
Corps line to be increased so that the signal 
arriving at the commercial facilities had the 
proper level. 

14 7 APPARATUS LIMITATIONS 

Effective service was accomplished on sig¬ 
nals having strengths of 50 /xv per meter or 
less. More modern techniques would enable 
this figure to be increased by a factor of five 
or more. The automatic volume control in the 
d-f channel of the u-h-f receiver obtained its 
voltage from the monitor channel so that the 
gain of the d-f channel was controlled in pro¬ 
portion to the input level of the monitor channel. 
Inasmuch as the monitor signal was not exactly 
constant as a funtion of antenna rotation, it 
was necessary to have a reasonably long time 
constant (approximately 1/2 second) in the a-v-c 
circuit for the d-f channel so that minor flue- 




















APPARATUS LIMITATIONS 


189 


tuations resulting from antenna rotation would 
not distort the d-f pattern and cause bearing 
error. Hence the d-f channel automatic volume 
control would, in general, respond to only rela¬ 
tively slow changes in signal level. Rapid 
changes caused a proportionate distortion in the 


d-f pattern which were indicated as instantane¬ 
ous bearing errors on the cathode-ray screen. 
Such rapid variations caused the indicated bear¬ 
ing to vary about the true azimuth. Averaging 
the bearings of several traces visually enabled 
the operator to obtain the correct bearing. 


■ ; VI. | 






Chapter 15 

ELECTRICAL DIRECTION-FINDER EVALUATOR 


Development of an electromechanical device which, 
from the bearings to a radio transmitter measured by 
any number of fixed radio direction finders, determines 
the most probable location of the transmitter and the 
boundary of the smallest region in which, to any pre¬ 
assigned probability, the transmitter can be presumed 
to be located. 


151 INTRODUCTION 

A t the time of this project 3 there were, in 
l use or available, a great number of radio 
direction finders capable of providing informa¬ 
tion which, if properly analyzed statistically on 
simultaneous bearings, could determine the 
location of a radio transmitter with much 
greater precision than had been obtained by 
methods of evaluation then existing. 

This report describes a device which, with¬ 
out mathematical approximations and almost 
instantaneously, can apply the method of least 
squares to the bearings of any number of direc¬ 
tion finders operating in a network. In con¬ 
junction with d-f networks organized to make 
optimum use of its properties, this electrical 
d-f evaluator was expected to place direction 
finding in an entirely new category of precision 
and dependability. 

13 2 STATEMENT OF PROBLEM 

A radio direction finder provides means for 
measuring the bearing to the source of a radio 
signal, and therefore two direction finders can 
provide sufficient information to determine the 
position of a radio transmitter, provided that 
the position of the transmitter is not on the line 
joining the two direction finders. 

The bearings from the two direction finders 
will determine a fix (point where the bearing 
lines cross) with an accuracy dependent upon 
the precision of the two direction finders. In 
common with all physical measurements, the 
bearings as obtained from a direction finder 

“Project 13-121, Contract No. OEMsr-1472, J. A. 
Maurer, Inc. 


deviate about the true value. And as with all 
physical measurements, if a number of values 
will be obtained and properly averaged, a 
resultant value will be obtained more dependa¬ 
ble than any of the individual values. 

The use of a number of direction finders 
instead of only two will provide information 
which, if properly averaged, will determine the 
location of a transmitter with greater precision 
than would the bearings from any two of them. 
In fact, the bearings from a large enough 
number of instruments can provide informa¬ 
tion for a fix of any desired accuracy. But the 
difficulty is in properly averaging the bearings. 
Unlike the measurements, for example, of the 
temperature at some location by a number of 
thermometers whose readings can be averaged 
by determining a simple mean value, the cor¬ 
rect bearing of a transmitter from each direc¬ 
tion finder of a network is in general a different 
value, and thus the mean value of the several 
bearings from direction finders located at dif¬ 
ferent positions has no significance. If a meth¬ 
od for correctly averaging their readings is 
used, the accuracy of a d-f fix is theoretically 
limited only by the number of direction finders. 
In Appendix A of the final report 1 the theory 
is fully expounded. 

13 3 VISUAL D-F EVALUATION 

The method usually employed in averaging 
the information obtained from a number of 
direction finders is to plot the bearings on a 
map of the region involved, and then, by visual 
observation, to estimate on the map the most 
probable location of the transmitter. This proc¬ 
ess makes use of various rules-of-thumb, geo¬ 
metrical constructions, and common-sense ap¬ 
proximations in an attempt to obtain the coor¬ 
dinates of the most probable location of the 
transmitter. The more direction finders there 
are in a network, the less likely is the result of 
visual evaluation to approach the correct solu¬ 
tion of the proper averaging process. The other 


190 



USE OF THE SUMS OF THE SQUARES OF THE DEVIATIONS 


191 


desired value: the boundary of the “search 
region,” that is, of the smallest region in which 
to any preassigned probability the transmitter 
can be presumed to be located, cannot even be 
estimated by the visual evaluation method com¬ 
monly employed. And yet this information may 
be very important in certain situations, such 
as, for instance, upon the reception of a distress 
signal, when the size and shape of the area 
most profitably to be searched by rescue craft 
should quickly be determined. 

15 4 GROUP D-F SYSTEM OF 

EVALUATION 

Another method to average the values from 
several direction finders has been attempted. 
This requires that a number of direction finders 
be located so close together that in effect they 
may be considered to have the same geographi¬ 
cal location yet they must be far enough apart 
to prevent electrical coupling and to allow the 
errors in each instrument to be entirely un¬ 
correlated. Thus if half of the direction finders 
are grouped at one location and half at another, 
the bearings within each group may be aver¬ 
aged by simply computing the mean value, and 
the resulting two bearings are used to obtain a 
fix on a map as if each were from a single 
direction finder, except that each mean bearing 
should be more precise than a bearing from a 
single direction finder. As experimentally 
tested, this group d-f system has been disap¬ 
pointing. Aside from the obvious limitation of 
having only two locations, it was found that 
when several direction finders were placed close 
enough to be treated as at one geographical 
point (not more than 2 miles apart) the 
deviations were not statistically random, and 
so in other words the errors were correlated, 
and the mean value of the bearings taken by a 
group was not much more dependable than the 
bearing from one direction finder alone. 

15 5 USE OF THE SUMS OF THE 

SQUARES OF THE DEVIATIONS 

The requirements for properly averaging the 
bearings from a number of separated direction 
finders may be represented geometrically in 
Figure 1 where the dotted lines represent the 


reported bearings from three direction finders 
as plotted on a map of the region, and the solid 
lines represent assumed bearings which meet 
in a common point T. The angles b lf b 2 , and b 3 



Figure 1. Location of transmitter by three d-f 
stations, dotted lines representing reported bear¬ 
ings, solid lines being assumed bearings which meet 
in common point T. 

are the deviations between the reported bear¬ 
ings and the assumed bearings to the common 
point T. If the deviations of each direction 
finder are normally distributed (this is de¬ 
scribed in Appendix A of the final report 1 ), 
then the most probable location of the transmit¬ 
ter is that position of T for which the sum of 
the squares of the deviations is a minimum. 

A method which has been developed for 
evaluating d-f fixes analytically on a map com¬ 
prises a series of computations of the sums of 
the squares of the deviation angles. In the 
neighborhood of the estimated location of the 
fix, a number of points in regular pattern are 
marked. By means of a transparent protractor, 
the deviation angle of each point from the re¬ 
ported bearing line of each direction finder is 
measured. These angles are then squared and 
added together for each of the points. The re¬ 
sulting values of the sums of the squares com¬ 
puted for each point give an indication of 
where the minimum sum would be located if an 
infinite number of points were used. 


CONFIDENT!\l f 







192 


ELECTRICAL DIRECTION-FINDER EVALUATOR 


15 6 BASIC PRINCIPLES OF THE 
ELECTRICAL D-F EVALUATOR 

The electrical d-f evaluator does not use any 
approximations nor are any computations re¬ 
quired during the actual evaluation process. 
Instead, it provides a mechanism whereby the 
common point T of Figure 1 can be moved to 
any position and simultaneously a reading pro¬ 
portional to the sum of the squares of the 
deviation angles is indicated on an electric 
meter. Thus by varying the position of T until 
the sum of the squares of the angles of devia- 


is a constant value (indicated by a constant 
meter reading) is a contour of constant proba¬ 
bility density for the location of the transmit¬ 
ter. For any number of direction finders and 
any desired probability a value of this sum may 
be determined. Actually, the value of the sum 
has been computed for various probabilities 
and is provided with the evaluator in the form 
of a table. 

In the development of the evaluator, tests 
were run on various d-f networks which verified 
the requirement that deviations of direction 
finders are approximately normally distributed, 



Figure 2. Electrical direction-finder evaluator. 


tion is a minimum the most probable location 
of the transmitter can be determined. The con¬ 
tour which bounds the smallest region in which 
to any preassigned probability the transmitter 
can be presumed to be located may also be 
determined from the sums of the squares of the 
deviations. Each curve along which this sum 


and thus the method of least squares is proper 
for these determinations. 

The above description assumes that the bear¬ 
ings as reported from each direction finder are 
equally dependable. In case it is known that 
the several direction finders have unequal 
precisions, the deviation angles (b lf b 2 , and b z 






GNOMONIC CHART DISTORTION CORRECTION 


193 


of Figure 1) are weighted by dividing each by 
the standard deviation of the respective instru¬ 
ment. In the evaluator the weighting is per¬ 
formed in a circuit in which the squares of the 
deviation angles are measured, and therefore, 
the weighting control is a measure of the 
variance , which is the square of the standard 
deviation. 

im BASIC MECHANISMS OF THE 
ELECTRICAL D-F EVALUATOR 

The electrical d-f evaluator, illustrated in 
Figure 2, is approximately the size and shape 
of the visual evaluating tables now in use by 
the Army Airways Communication System and 
the U.S. Coast Guard. It performs the opera¬ 
tions of determining the minimum value of the 
sum of the squares of the weighted angles of 
deviation by means of a number of protractors 
located at points representing the positions of 
the direction finders on a gnomonic chart of the 
region involved. Each protractor electrically 
measures the square of the angle between the 
reported d-f bearings and the great-circle line 
to the common point T of Figure 1. In the 
evaluator, this point can be manually moved to 
any position on the map, and is called the scan¬ 
ning point. Each protractor is a form of poten¬ 
tiometer carrying 60-cycle alternating current 
and is constructed basically of a resistance 
strip attached to the bearing disk which can be 
oriented to the azimuth reported by the direc¬ 
tion finder, and a wiper attached to a telescop¬ 
ing pointer arm which leads to the scanning 
point. From each protractor a separate pointer 
arm leads to the same scanning point. The 
resistance strip and wiper of the protractor 
are so designed that a voltage is obtained pro¬ 
portional to the square of the angle measured 
by the relative position of the pointer arm from 
the reference line on the bearing disk. Exter¬ 
nal to the protractor is a selector switch which 
permits the 60-cycle current to each protractor 
resistance strip to be so regulated that the volt¬ 
age from each can be weighted according to 
the variance of the direction finder represent¬ 
ing the protractor. The voltage from each pro¬ 
tractor is applied to the primary of one of a 
bank of “summation transformers.” The sec¬ 
ondaries of these summation transformers are 


in series, and the series output is applied to the 
grid of a vacuum-tube amplifier whose ampli¬ 
fication is variable in five steps. The output of 
this amplifier actuates the “summation meter,” 
and this is the meter whose reading is propor¬ 
tional to the sum of the squares of the weighted 
deviation angles. 

158 PANTOGRAPHS 

Because each protractor is a fairly large 
component (about 4 in. in diameter) and be¬ 
cause direction finders are occasionally located 
rather close together, it would not be practical 
to place all the protractors side by side on a 
chart. In the electrical d-f evaluator this diffi¬ 
culty is resolved by providing a number of 
decks, permitting the different protractors to be 
located at different levels, but each is directly 
below the point on the gnomonic chart repre¬ 
senting the position of its corresponding direc¬ 
tion finder. The scanning point appears as the 
reference point with a marking pencil at the 
end of a movable arm just above the map on top 
of the evaluator structure, but at each deck of 
the evaluator there is a duplicate scanning 
point attached by a horizontal pantograph and 
vertical shaft assembly to the scanning point 
so that it always remains directly below it. It 
is to the duplicate scanning points that the 
pointer arms from the various protractors are 
pivoted. 

15 9 GNOMONIC CHART DISTORTION 
CORRECTION 

The chart or map used with the evaluator 
must be a gnomonic projection because only 
with such a projection are all great circles 
represented by straight lines. This projection, 
however, has one property which presents dif¬ 
ficulties in measuring the angles of deviation 
at various parts of the map. This property is 
called nonconformality and because of it angles 
on the surface of the earth are not preserved 
in the flat projection. 

To overcome this difficulty, a corrector as¬ 
sembly is employed in each protractor by which 
the angle between the wiper and the resistance 
strip is altered by a cam to compensate exactly 
the gnomonic distortion. 



194 


ELECTRICAL DIRECTION-FINDER EVALUATOR 


15 10 D-F BEARING INPUT 

At the right end of the evaluator is a series 
of boxes called bearing-input boxes, one for each 
direction finder of the network. Each contains 
an internally illuminated translucent drum 
with an engraved scale reading 0-360° which 
can be rotated by means of a 36/1 ratio bear¬ 
ing knob to the bearing reported by the corre¬ 
sponding d-f station. A flexible shaft runs from 
the bearing knob of the bearing-input box to 
the bearing disk of its associated protractor 
through a 36/1 ratio worm so that drum and 
protractor rotate together. 

Each bearing-input box also has a 10-point 
switch by which the current to the resistance 
strip in the protractor can be varied to provide 
proper weighting for deviation angles accord¬ 
ing to the known dependability of the particu¬ 
lar direction finder. The weighting is depen¬ 
dent upon the statistical history of each direc¬ 
tion finder. 

Means are provided for visual evaluation in 
case a breakdown occurs of the electromechani¬ 
cal system. 

1511 METHOD OF OBTAINING 
THE DESIRED DATA 

When all the reported bearings have been 
entered into the bearing-input boxes and the 
variance switches are at their proper positions, 
the operator moves a vertical pencil on the end 
of a pantograph arm above the map. With one 
hand controlling the sensitivity of the summa¬ 
tion amplifier, the pencil is moved until a 
minimum is noted on a summation meter. A 
mark is made on the map at this point. Then 
the pencil is moved perpendicularly to the first 
straight line and a new motion described 
parallel to the first line and a mark made when 
a new minimum is found. Now on a line join¬ 
ing these two points a third minimum will be 
found. It will be very close to the most probable 
location of the transmitter. The pencil may be 
caused to describe short motions about this 
point to find an absolute minimum and this 
will locate the most probable location of the 
transmitter. 

Means are provided for rejecting “wild” 
bearings. In the contractor’s final report 1 are 


given a procedure for describing the boundary 
of search regions of any given probability, and 
statistical data resulting from field tests on 
east and west coasts; also the report gives 
consideration to further developments of the 
electrical-evaluator circuits, directions for 
making the cams, the use of servo mechanisms 
to eliminate the manual manipulation of the 
protractors, and to means of making the com¬ 
putations required automatic. 

1512 ACKNOWLEDGMENTS 

In the design and development of the elec¬ 
trical d-f evaluator, certain individuals and 
groups in the Armed Services of Great Britain 
and the United States rendered considerable 
assistance. 

The theoretical and practical requirements 
for an improvement in d-f evaluation were 
originally presented to Division 13 in great 
detail by Captain Stuart Martin, Office of Chief 
Signal Officer, U.S. Army. The results of his 
considerable research on statistical methods 
in d-f evaluation were generously provided by 
Commander D. H. Menzel, Op. 20G, U.S. Navy. 

The long-term statistical studies of d-f sta¬ 
tion errors and group d-f station experiments 
carried on in Great Britain by Crampton and 
Redgment were made available, together with 
valuable interpretive information, by Admiral¬ 
ty Signals Establishment. 

The U.S. Army Airways Communications 
System and the Air Sea Rescue Section of the 
U.S. Coast Guard cooperated continuously 
through their headquarters, their training 
centers, and their several d-f evaluation offices 
in providing equipment, operational data, 
special records, and experimental information 
at all stages of the development. 

Kenneth A. Norton and Ross Bateman, 
attached to the Office of Chief Signal Officer, 
U.S. Army, were, through the later stages of 
development, in such close cooperation with the 
designers that certain features of the evaluator 
are directly attributable to them. 

The final report was prepared jointly by 
personnel of Division 13, NDRC, by personnel 
of the Applied Mathematics Group, and by J. A. 
Maurer, Inc. 




PART III 


RADIO AND WEATHER 









































































































































































Chapter 16 


A STUDY OF SFERICS 


The work on this project a was divided into two parts. 
The first was a survey of existing literature on the 
subject of atmospherics and their relation to weather 
information; the second consisted in the operation of 
two radio stations in New Mexico in cooperation with 
the Signal Corps to gather visual, electrical, meteoro¬ 
logical, and photographic data on local thunderstorms. 
While the contractor submitted completion reports 1 - 2 
covering both phases of the project, the summary fol¬ 
lowing is condensed only from the one 4 covering the 
experimental operations. 


161 INTRODUCTION 

T he purpose of this project was to gather 
as much data as possible on thunderstorms 
and the types of sferics (atmospherics) they 
produced with the object of answering the fol¬ 
lowing questions. 

1. Can thunderstorms be located accurately? 

2. Given a distribution of thunderstorms, 
can the weather situation be analyzed ? 

3. Are there characteristics of sferic signals 
which can be associated with storms of definite 
type or energy which will supplement or clarify 
the information obtained from geographical 
distribution of storms? 

4. In any given region do thunderstorms oc¬ 
cur with such frequency that the sferic direc¬ 
tion-finding technique can be profitable? 

The project consisted of two parts, a survey 
of the pertinent literature available and an ex¬ 
ploratory experimental program. Only the ex¬ 
perimental program is described herein. 


16 2 EQUIPMENT UTILIZED 

Two observing stations were set up, one at 
the University of New Mexico in Albuquerque 
and one on top of the Sandia Mountains. The 


a Project 13-115, Contract OEMsr-1485, University 
of New Mexico. 


Signal Corps provided a mobile-unit-equipped 
sferic-waveform and d-f apparatus which was 
located at various distances from 80 to 1,500 
km from the University station. The observa¬ 
tional data on lightning flashes were synchro¬ 
nized with the sferic records in the mobile unit 
by means of radio signals. The Signal Corps 
also provided waveform and d-f apparatus for 
use at the University and a B-17 plane with 
equipment similar to that in the mobile unit. 
The plane was not continuously available 
during the time of the project. 

Each station was equipped with an electrical 
potential gradient change recorder consisting 
of an exposed insulated electrode connected to 
a quartz string electrometer and to ground 
through a high resistance. 5 The time constant 
of the system was chosen so that gradient 
changes due to lightning strokes occurring with¬ 
in a few hundredths or tenths of seconds 
produced large electrometer deflections but 
slow gradient changes of seconds’ duration 
produced no deflections. The gradient changes 
(electrometer deflections) were recorded on a 
16-mm film moving at constant speed past a 
slit 0.002 in. wide. The instruments were suf¬ 
ficiently sensitive to record gradient changes 
due to lightning strokes within a radius of 25 
miles and fast enough to resolve gradient 
changes due to repeated elements of lightning 
flashes. 

Each station also was provided with a tape 
recorder on which the time, type, and azimuth 
of lightning flashes and the time of the thunder 
were recorded. Frequent time signals and 
lightning stroke signals were keyed on the 
gradient change recorders and simultaneously 
transmitted by radio to the mobile unit to 
synchronize the several records. In addition, 
each station was equipped with an alidade to 
measure storm and lightning flash azimuth and 
cloud base and top elevation angles. 

Time lapse photographs of cloud development 
were taken from each station. 


197 



198 


A STUDY OF SFERICS 


Mobile Unit 

The sferic d-f equipment in the mobile unit 
consisted of AN/GRD-1 apparatus 4 made up of 
two square loops mounted at right angles for 
detecting perpendicular components of the in¬ 
coming signals. The separate amplifiers were 
properly phased and the component signals im¬ 
pressed on the horizontal and vertical plates 
of a cathode-ray tube. The sets were tuned to a 
frequency of approximately 10 kc. 

The sferic waveform equipment consisted of 
a vertical 36-ft antenna, an aperiodic antenna 
circuit, an amplifier with nearly constant am¬ 
plification up to about 200 kc, a cathode-ray 
tube, and a triggering circuit. The latter started 
the sweep after the sferic was received with a 
delay of about 5 /xsec. The amplified sferic 


nals was mounted between the scopes. The film 
moved continuously at a rate of approximately 
2 in. per second. 

163 OBSERVED WAVEFORMS AND 
STORM DISTANCE 

The waveforms observed can be divided into 
three principal types. 

1. A series of prominent, easily distinguished 
features (e.g., maxima, sharp breaks), usually 
with amplitudes decreasing in a fairly regular 
fashion forming a repeated pattern. The usual 
sweep with l,300-/xsec time base showed from 
two to five such features. The interval between 
the features characteristically increased from 
100 to 200 /xsec at the beginning of the trace to 
400 to 600 fisec at the end of the trace. 


APERIODIC 



SFERIC 
WAVE FORM 
C R O 


D-F 

CRO 


Figure 1. Block diagram of d-f and waveform equipment. 


was impressed on the vertical plates so that the 
cathode-ray tube trace represented the field 
variations of the sferic signal with time. The 
time base or sweep used varied between 1,500 
and 2,000 jx sec. The sweep was calibrated by 
impressing 10-kc or 20-kc sinusoidal signals of 
various amplitudes on the apparatus. A block 
diagram of the d-f and waveform equipment is 
given in Figure 1. 

Both the d-f and waveform scopes were 
photographed simultaneously by a 35-mm 
camera. A signal lamp for synchronizing sig- 


2. A series of prominent features with less 
regular intervals and greater amplitude varia¬ 
tion than in the first type. The waveform fre¬ 
quently suggested an interference pattern 
formed by two or more superimposed pulses or 
oscillations. 

3. Very complicated waveforms with varying 
amplitude and with intervals between maxima 
from 10 to 100 ^sec. 

Waveforms of Type 1 were analyzed accord¬ 
ing to the suggestions of Laby 7 and Schonland 8 
on the assumption that the pulses or oscillations 



































RESULTS OF ANALYSES 


199 


were due to multiple reflections between earth 
and ionosphere. According to this hypothesis 
the time of transit of an electromagnetic dis¬ 
turbance from a lightning stroke to the observ¬ 
ing station is 

tn=~ (4 nV + d*)' A 

where c is the velocity of propagation of the 
disturbance; 

h is the height of the ionosphere; 
d is the great circle distance between 
source and observer; 
n is the number of reflections at the 
ionosphere experienced by the pulse. 

The time between the arrival of a pulse 
which has been reflected at the ionosphere n 
times and one which has been reflected n — 1 
times is 

1 ( A r -\a\ 

Atn = — U4n 2 h* + d 2 ) - 4(n- 1 ) 2 h 2 + d 2 l- 

In the analysis of the sferic waveforms, the 
procedure was to choose distinguishable re¬ 
peated parts of the pattern (maxima, minima, 
sharp breaks, etc.), measure the time intervals 
between them, and calculate h and d by the 
above formula. 


for by multiple reflections from an ionosphere 
90 km in height suggesting a storm to the east 
where the path of the sferics would be in the 
dark. The largest concentration of directions 
lay between 80° and 90° azimuth with a maxi¬ 
mum at 85°. The calculated distance of the 
sources was 1,375 ± 100 km. The storms pro¬ 
ducing the sferics were thus located within 
120 km of the center of Arkansas. Weather 
data of the date showed that a number of 
thunderstorms occurred along a cold front ex¬ 
tending from Arkansas to Pennsylvania. At the 
time the records were made a storm was in 
progress at Little Rock, Arkansas. Thus the 
location of the storm at this site without previ¬ 
ous knowledge of its existence on the part of 
those analyzing the records offers convincing 
evidence of the validity of the multiple reflec¬ 
tion hypothesis. 

Lightning Flashes and Storm Character 

A study of the visual and electrical potential 
gradient change records of lightning strokes 
in storms near Albuquerque during August and 
September, 1945, yielded some interesting pre¬ 
liminary results. In this group of storms, the 
frontal storms were more intense, they had a 


Table 1 . Results of waveform analysis. 


Date 

Time of obs. of 
correlated 
flashes, 
M.S.T. 

Storm distance 
from mobile 
unit d 
in km 

Calculated 
ionosphere 
height h 
in km 

Path of sferic 

No. of corr. 
waveforms 
consistent with 
h and d 

No. of wave¬ 
forms corr. 
by time 
only 

No. of wave¬ 
forms corr. by 
time and 
direction 

Aug. 30 

1938-1946 

400 

90 

Dark 

3 

4 

4 

Aug. 31 

1437-1500 

580 

75 

Light 

21 

23 

21 


1513-1515 

580 

78 

Light 

2 

2 

2 


1543-1550 

600 

80 

Light 

5 

7 

7 


1626-1630 

600 

82 

Light 

10 

11 

11 

Sept. 6 

1853-1915 

840 

85 

Dark 

4 

5 

4 



Total. 



45 

52 

49 


16 4 RESULTS OF ANALYSES 

The results obtained by this means of analyz¬ 
ing the waveforms are given in Table 1. 

The record of the storm of November 2, 1945 
disclosed many simple patterns which, upon 
analysis, indicated that they could be accounted 


greater stroke frequency, a relatively larger 
number of cloud-ground strokes, and a larger 
number of repeated elements per stroke than 
intra-air-mass storms. If these observations 
are supported by further studies over entire 
thunderstorm seasons and in different climatic 
regions, there is a possibility of determining 

















200 


A STUDY OF SFERICS 


storm types as well as storm distance from 
sferics waveform records. 


165 GENERAL CONCLUSIONS 

1. Sferic signals from lightning flashes ex¬ 
perience multiple reflection between the iono¬ 
sphere and earth. The repeated pattern wave¬ 
form produced by sferic pulses which travel 
paths of different length due to different num¬ 
bers of ionospheric reflections may be used to 
calculate the height of the ionosphere and the 
distance of the flash from the observing station. 

2. The use of waveform equipment to deter¬ 
mine lightning flash distance in conjunction 


with equipment to measure direction and sense 
of the sferic signal makes possible location of 
thunderstorms from a single station. A 
thorough test of this technique should be made. 

3. Preliminary results on a small group of 
thunderstorms in one climatic region indicate 
that frontal and nonfrontal storms differ in 
lightning flash frequency, relative number of 
cloud-ground and cloud-cloud flashes, number 
of repeated elements in cloud-ground flashes, 
and the duration of cloud-ground flashes. 

4. The great advantage of determining storm 
type or intensity from sferics records indicates 
that the preliminary results should be checked 
and extended by observations on storms in 
several climatic regions. 





PART IV 


ANTENNA RESEARCH 
































































































































































































































Chapter 17 


ANTENNA PATTERNS FOR AIRCRAFT 


Studies and experimental investigations in connection 
with antenna patterns for aircraft and tanks as a func¬ 
tion of location of the antenna, frequencies employed, 
etc., also development of the “model” technique for 
studying aircraft antenna impedances and patterns. 
This contract was administered by Division 13 until 
April 1, 1944, when it was transferred to Division 15. 

171 INTRODUCTION 

P ROJECT C-ll a was initiated by NDRC at 
the request of Aircraft Radio Laboratory. 
Wright Field, to achieve the following princi¬ 
pal aims. 

1. To investigate methods for measuring 
antenna patterns on aircraft at various fre¬ 
quencies. 

2. To measure the patterns of various anten¬ 
nas on various types of aircraft at various 
frequencies. 

3. To obtain general statements on the 
effects of aircraft structure, antenna location, 
frequency, and other factors on the radiation 
patterns. 

4. To investigate the patterns of various 
special antennas and antenna arrays. 

5. To investigate methods for improving 
patterns of aircraft antennas for specific ap¬ 
plications. 

6. To investigate the construction of models 
to determine the accuracy of construction re¬ 
quired. 


172 RESULTS ACCOMPLISHED 

Although measurements of aircraft patterns 
using models had been made for several years 


a Project C-ll, Contract No. NDCrc-100, Ohio State 
University. 


prior to the start of this project, the measure¬ 
ments were limited to simple types of antennas 
and to an upper frequency of about 500 me. 
Under the project, techniques and equipment 
were developed to extend the model methods to 
a greater variety of structures and to cover 
greater frequency ranges. After the equipment 
and techniques had been developed to the point 
where routine measurements could be made, at 
frequencies as high as 10,000 me, patterns of 
various antennas were investigated to deter¬ 
mine the general factors which influence the 
patterns. It was found possible to predict the 
general features of patterns of simple types of 
aircraft antennas. 

Modeling techniques were applied to a 
variety of special problems and it is believed 
that these applications are new. Methods for 
measuring propeller modulation and for 
measuring ellipticity of polarization of aircraft 
antennas were developed. Modeling techniques 
were applied in the investigation of a tank an¬ 
tenna problem. The possibility of using models 
for measuring radar echoes from aircraft was 
considered and development of methods started. 
Methods using models for measuring the im¬ 
pedances of aircraft antennas were investi¬ 
gated. 

The research program outlined above was 
requested by Wright Field in order to develop 
the model technique for use as a tool in the de¬ 
sign of aircraft antennas to meet definite speci¬ 
fications. Models were used in the investiga¬ 
tion in preference to full-scale aircraft since 
they furnish more information with less labor, 
time and cost. The fact that the actual airplane 
is not always available for antenna tests also 
was an important factor. 

The information and techniques developed 
on this project were used in the design and de¬ 
velopment of aircraft antennas for a wide 
variety of applications. 


203 



204 


ANTENNA PATTERNS FOR AIRCRAFT 


17.3 PATTERNS of antennas on 
AIRCRAFT AT VARIOUS 
FREQUENCIES 

It is not easy to predict from theoretical con¬ 
siderations alone the approximate patterns to 
be expected from a proposed antenna installa¬ 
tion on an airplane. The relative importance 
of reflection and diffraction effects and the 
nature of the current distributions on the sur¬ 
faces of the aircraft are difficult to estimate. 
If sufficient antenna patterns measured under 
a wide range of conditions are available, it be¬ 
comes possible to make a better estimate of an 
antenna pattern. To provide such patterns, a 
group of patterns has been obtained over a 
wide frequency range for simple antennas 
mounted on various types of aircraft. 

Only the patterns for the principal planes 
have been measured. It has been found that 


e = o° 



Figure 1. Spherical coordinate system used in 
measurements. 

principal plane patterns are almost as useful 
as complete three-dimensional patterns, and 
much easier to obtain. The orientations of the 
coordinate planes with respect to the aircraft 
are shown in Figure 1. 

The following is a list of the patterns in¬ 


cluded with the contractor’s final report dated 
August 24, 1943. 1 

B-17F 

A 4-ft whip antenna on the lower frequen¬ 
cies and a A/4 stub at 15, 25, 35, 50, 75, 100, 
150, and 200 me, the antennas being located 
(1) directly ahead of the bomb bays, project¬ 
ing vertically downward, (2) directly behind 
the bomb bays, projecting vertically downward, 
(3) 4 ft ahead of the leading edge of the hori¬ 
zontal stabilizer, projecting vertically down¬ 
ward from the belly of the ship, and (4) cen¬ 
tered on wings on top of fuselage, projecting 
vertically upward. 

A-20-A 

A a/ 4 stub antenna on top of the fuselage, 
immediately above the trailing edge of the 
wing at 50, 100, and 200 me. 

P-38 

A 4-ft whip antenna projecting forward 
from the nose at 50, 100, 150, and 200 me. 

P-47 

A 4-ft whip antenna just behind the pilot’s 
cockpit at 50, 100, and 200 me. 

B-25 

Two types of antennas, a A/4 stub and a A/2 
*coaxial-type dipole at 100 and 200 me. The an¬ 
tennas were mounted in two locations, on top 
of the fuselage, first just above the leading 
edge of the wing, and then above the trailing 
edge. 

17.4 typical antenna patterns 

In making the measurements only half of the 
pattern was measured in those cases where 
symmetry could be assumed. The symmetry 
was checked in several of the patterns and 
found to be adequate. 

In Figures 2 and 3 the row of patterns on the 
left is for the plane 6 = 90°, the center row 
for the plane defined by = 0° and 180°, and 
the right-hand row for the plane </> = 90° and 










TYPICAL ANTENNA PATTERNS 


205 






vPrtY^i L± n Q \ e - nna 4-^. st ^ b on belly of B-17F directly in front of bomb bays. Dotted lines indicate 

vertical polarization, Ee ; full lines indicate horizontal polarization, E<t>. 


270 . In Figures 4 and 5 the patterns for The principal plane patterns in any hori- 
angles 10° below the horizon (0 = 100°) have zontal row in Figures 2 and 3 are plotted on 
been plotted also. the basis of a constant power input and there- 




















206 


ANTENNA PATTERNS FOR AIRCRAFT 



Figure 3. Antenna patterns of X/4 stub on belly of B-17F directly in front of bomb bays. Dotted lines indicate 
vertical polarization, Ee; full lines indicate horizontal polarization, E<t>. 


fore may be directly compared. It is not per¬ 
missible to make direct comparisions of rela¬ 
tive signal strengths between patterns in dif¬ 
ferent rows. 

The pattern of any simple antenna mounted 
on an airplane may be estimated with the aid 
of these sample patterns. The sample patterns 
which approximate the conditions of the an¬ 


tenna whose pattern is to be estimated are com¬ 
pared to determine the amount of diffraction 
and reflection to be expected. If the current 
distribution on the antenna is expected to dif¬ 
fer considerably from that obtaining on the 
stubs used in these measurements, due allow¬ 
ance for its effects on the pattern must be 
made. It will be found, however, that the 













METHOD OF MEASUREMENT EMPLOYED 


207 


(j) = 180° 


0 = 0 ° 


(j> = 270° 


) = 90° 




Figure 4. Patterns of X/4 stub at 100 me on top fuselage above leading edge of wing of B-25. Dotted lines indicate 
vertical polarization, Ee\ full lines indicate horizontal polarization, E<t>. 


sample patterns will be approximately correct 
for linear antennas of lengths from a small 
fraction of a wavelength up to roughly %A. 

As an additional aid in estimating antenna 
patterns, a number of patterns were measured 
on a A/4 stub mounted on a prolate spheroid, 
which approximates a fuselage. It is apparent 
from the patterns in Figures 6 and 7 that their 
shapes are determined more by the nature of 


the current distribution on the spheroid than 
by the current distribution on the antenna. 


175 METHOD OF MEASUREMENT 
EMPLOYED 

A fairly adequate description of the princi¬ 
pal methods employed in measuring antenna 














208 


ANTENNA PATTERNS FOR AIRCRAFT 


patterns with models is given in the final re¬ 
port dated August 31, 1942. 1 A few minor 
changes were made as a result of experience. 
The vibrator method described briefly below 
has certain advantages over other methods 
especially in certain applications. The fact 


<J> = I80° <t) = 0° 



the other hand, the vibrator method has certain 
disadvantages. 

1. The amount of modulation obtainable 
with a commercial vibrator is very low at fre¬ 
quencies about 2,000 me due to unavoidable 
stray reactances and losses in the vibrator. 


4> = 270° <j> = 90° 





Figure 5. Patterns of X/4 stub at 200 me directly above leading edge of wing of B-25. Dotted lines indicate ver¬ 
tical polarization, Ee; full lines indicate horizontal polarization, E<t>. 


that no connecting wires to the model are re¬ 
quired is of particular advantage in some 
measurements. The phasing adjustment offers 
possibilities for investigating the ellipticity of 
polarization of radiation from an antenna. On 


2. The need for phasing the system for each 
reading increases the time required to measure 
a pattern compared to other methods. It is 
possible, probably, to eliminate this phasing 
adjustment. 















METHOD OF MEASUREMENT EMPLOYED 


209 


3. The signal levels obtained are low, and 
the system is rather sensitive to changes in 
components. 



f \J VV) 





-— \ 


= 18.5 CM 


— \ = 16.5 CM 


Figure 6. Patterns obtained from X/4 stub pro¬ 
jecting from one end of prolate spheroid 15x60 cm 
in dimensions, vertical polarization. 


current which flows in the model antenna is 
modulated by connecting a periodically vary¬ 
ing impedance (tuned vibrator) to the termi¬ 
nals of the antenna. As a consequence of the 
variations in antenna current, a modulated 
wave is re-radiated. Some of the re-radiated 
energy re-enters the transmitting antenna 
system where it is picked up by a receiver sen¬ 
sitive to modulation only. Since there are two 
signals entering the receiver, the audio output 
of the receiver depends upon their relative 
phase. The phase may be varied by adjusting 
the separation between the model and the trans¬ 
mitting antenna. Variations in the adjustment 
for proper phasing (maximum audio output) 
yield information on phase variations in the 
field re-radiated from the model. 


The New Method 

The method employed for the majority of the 
pattern measurements uses a bolometer (Little- 
fuse) detector as a receiver in the model to de¬ 
tect modulated signals from a horn radiator. 
Small wires are used to connect the output of 


<t> = 45° 



= 90° 




F = 600 MC 

Figure 7. Patterns of X/4 stub on side of prolate spheroid, 15x60 cm in dimension, parallel to minor axis; 
vertical polarization. 



The Vibrator Method 

An unmodulated transmitter produces a rel¬ 
atively uniform field in the region occupied by 
the model exciting the model antenna. The 


the receiver to the observing position. Pro¬ 
vided suitable precautions are taken, the dis¬ 
tortion of the antenna pattern due to the 
presence of these wires in the field can be kept 
small. For antennas of low efficiency, the out- 
























210 


ANTENNA PATTERNS FOR AIRCRAFT 


put of a bolometer is rather low so that a sili¬ 
con crystal detector is usually substituted. 

The output of most detectors is essentially 
proportional to the square of the input voltage. 
Since antenna patterns are usually plotted on 
a voltage basis (to accommodate the large vari¬ 
ations in signals found in most patterns) it is 
necessary to take the square root of the volt¬ 
age output of the receiver. An amplifier which 
does this automatically has been constructed. 
It is essentially a logarithmic 50-kc amplifier 
whose components have been adjusted to give 
the desired square root characteristic. 

The model supporting structure described on 
page 34 of the final report dated August 31, 
1942, 1 is now used exclusively. Selsyn indicators 
give a remote indication of the rotational posi¬ 
tion of the horizontal member. The horn radia¬ 
tor is on rollers to allow complete freedom of 
rotation about its longitudinal axis. 


17.6 PATTERNS OF BALANCED 
ANTENNAS 

Patterns of antennas requiring a balanced 
feed cannot be measured as simply as those 
using a coaxial-feed system. Particular care 
must be taken to assure a balance in the cur¬ 
rents on the feed line otherwise stray currents 
appear on the outer shield, distorting the 
measured pattern. 

Since the measuring equipment was original¬ 
ly designed for use with coaxial lines, the first 
method used on balanced antennas employed a 


BOLOMETER 

DETECTOR 


RECTIFIED 

SIGNAL 


> 

" 4 X * 

A H- 

Viz_ U .- - "1 

1 -‘LINE TUNER 

MATCHING UNIT 



ANTENNA 




Figure 8. Coaxial skirt balancing unit. 


A/4 skirt or balancing section on the end of a 
coaxial line to obtain the phase reversal re¬ 
quired for a balanced antenna. (See Figure 8.) 
This method has several disadvantages, the 


most important being the necessity for chang¬ 
ing the length of the skirt with each frequency 
change. Also, the length of the skirt is quite 
critical if the antenna impedance is high. It is 
often difficult to find space in a model for the 
matching section. 

A modification of this method is shown in 
Figure 9. A sliding polystyrene plug inserted in 

MOVABLE POLYSTYRENE 
PLUG —j 


SHIELDED PAIR 


TO 

— a 

w MATCHING UNIT 

V _~ i x _ 

ANTENNA 

BOLOMETER 

DETECTOR 

0 

^ RECTIFIED 
- p. SIGNAL 


i 


LINE TUNER 


Figure 9. Tunable coaxial skirt balancing unit. 

the skirt unit allows some adjustment of the 
tuning of the skirt. The tuning range is rather 
restricted, however, and there is no good cri¬ 
terion for proper tuning. 

The next method tried used a balanced sys¬ 
tem throughout. Shielded-pair transmission 
lines and balanced detectors were constructed, 
as shown in Figure 10. Two coaxial tuners 



Figure 10. Balanced measuring system with sep¬ 
arate adjustments for balance. 


were used at the detector to allow adjustment 
of balance since the detectors were not quite 
symmetrical mechanically. This method was 
found to be satisfactory for a wider range of 
antenna impedances than the previous methods. 
There was still a lack of a criterion for proper 
tuning, however. 

A system which achieved greater mechanical 





























































THE SIMULATION OF DIELECTRICS IN MODELS 


211 


and electrical symmetry is shown in Figure 11. 
A twin-line tuner and dual detectors were used. 
The two bolometers were connected in series 
for the audio output. This equipment was rel¬ 
atively satisfactory. 



Figure 11. Balanced system using dual detector. 


A system which uses a resonator to couple 
an unbalanced detector to a balanced transmis¬ 
sion line is shown in Figure 12. This avoids 
the difficulties encountered in constructing bal¬ 
anced detectors. 


V SHIELDED PAIR 

^—ANTENNA 


SMALL LOOPS'* 


ADJUSTABLE TUNER ROD 


3_L n 


RESONATOR 


TO STANDARD 
UNBALANCED DETECTOR 

Figure 12. Use of resonator for coupling balanced 
line to coaxial line. 


propagation of the signal, it is possible to ob¬ 
serve the variation in signal when the propeller 
is oriented in various directions. From the 
maximum and minimum signals observed it is 
possible to determine the percentage of modula¬ 
tion due to the propeller. 

178 MEASUREMENTS ON ELLIPTICALLY 
POLARIZED FIELDS 

Radiation from even simple stub antennas 
mounted on aircraft is elliptically polarized at 
the higher frequencies. It is to be expected, 
therefore, that measurements of the ellipticity 
of the radiation would yield information of 
value in interpretations of patterns. 

The major and minor axes of the ellipse of 
polarizations at any given point in a field can 
be readily measured by rotating a linearly po¬ 
larized antenna to determine the maximum and 
minimum signals. If the field is linearly pola¬ 
rized the minimum signal will be zero. If the 
field is circularly polarized there will be neither 
a maximum nor a minimum. To determine the 
direction of rotation of the electric vector 
around the ellipse special measurements are re¬ 
quired. The phasing adjustment used in the 
vibrator method for measuring antenna pat¬ 
terns makes its determination possible. 

Measurements have been made of the ellip¬ 
ticity of the field radiated from a simple vertical 
stub antenna located to the rear of the cockpit 
of a P-40 at 150 me. The data obtained are 
tabulated in Table I of Appendix I (report 
dated August 31, 1942). 1 Table II 1 was ob¬ 
tained from measurements of the field radiated 
from a A/4 stub antenna located on the side of 
a prolate spheroid parallel to a minor axis of 
the spheroid. There is a considerable amount 
of elliptically polarized radiation in directions 
not in the planes of symmetry. The direction 
of rotation of the polarization was not measured 
in the pattern for the P-40. 


PROPELLER MODULATION 


179 THE SIMULATION OF DIELECTRICS 
IN MODELS 


Preliminary tests were made to determine 
the feasibility of using models to study pro¬ 
peller modulation. For a given direction of 


An accurate simulation of a dielectric in a 
model is obtained by using a material whose 
dielectric constant is the same and whose con- 























































212 


ANTENNA PATTERNS FOR AIRCRAFT 


ductivity has been increased by the factor by 
which the dimensions have been reduced. Since 
suitable materials were not readily available, 
an investigation was conducted to determine 


type aircraft, is in some cases important. Also, 
in certain special cases it is necessary to model 
plastics such as Plexiglas. The enclosures used 
on some antennas, such as loops and high-fre- 




Figure 13. Reflection from 2-cm layer of wood. A, horizontal polarization; B, vertical polarization, (e — 4; 
a = 50X 10~ 15 emu.) 


methods for constructing approximate models 
for pattern measurements. The modeling of 
plywood, such as is used for constructing cargo- 


quency radar antennas, sometimes affect the 
pattern of the antenna. 

The precise calculation of the effect of a 



















THE MODEL TECHNIQUE 


213 


curved dielectric surface, such as a plywood 
fuselage, on the propagation of waves radiated 
from an antenna is difficult and involves too 
much labor to be practical. Much useful infor¬ 
mation is obtainable, however, from the simpler 
calculation involving plane waves and plane 
surfaces. 1 

The reflection coefficient for a 2-cm layer of 
plywood was calculated for a number of angles 
of incidence and for both vertical and hori¬ 
zontal polarization. The values for dielectric 
constant e and the conductivity o- were obtained 
by averaging published values from a number 
of sources. At the time of the calculations the 
data of Roberts and Von HippeT was not avail¬ 
able. The results of the calculations are shown 
in Figure 13. An examination of these figures 
shows that there is negligible reflection for 
wavelengths longer than about 10 meters. As 
the wavelength is decreased below 10 meters 
the reflection increases to a maximum in the 
region around 15 cm. Beyond 15 cm the reflec¬ 
tion coefficient exhibits alternate maxima and 
minima, the plywood acting as a pure dielec¬ 
tric reflector. 

For antennas which operate at wavelengths 
longer than 10 meters the plywood may be ex¬ 
pected to have but small influence on the an¬ 
tenna pattern. Consequently it is not necessary 
to model the plywood at all. It will be necessary 
to model any conducting materials in the field of 
the antenna, such as the motors and gas tanks. 

For the region from 15 cm to shorter wave¬ 
lengths, a reasonably accurate model may be 
obtained using plywood of proper thickness in 
the model, since the conductivity becomes un¬ 
important. 

For the region between 15 cm and 10 meters, 
the situation is not so favorable. Reflections 
from the surfaces of the aircraft may have 
considerable influence on an antenna pattern. 
The model should be constructed of materials 
having the correct constants if accurate results 
are desired. An approximation to the pattern 
may be obtained by using a plywood model, and 
the results will usually be good enough to indi¬ 
cate the general performance of the antenna 
system. The errors in the pattern will depend 
on how much the waves reflected from the ply¬ 
wood surfaces contribute to the antenna pat¬ 
tern. 


1710 TANK ANTENNA PATTERNS 

The following investigation was undertaken 
to determine a method for measuring the 
patterns of certain high-frequency antennas 
mounted on a medium tank. It was considered 
necessary to include the effect on the patterns 
of the finitely conducting ground in the neigh¬ 
borhood of the tank. 


1711 THE MODEL TECHNIQUE 

The characteristics of the ground on which a 
tank is located may influence the pattern of an 
antenna on the tank in two ways. The most 
important effect at high frequencies is the 
change in the pattern due to reflections from 
the surface of the earth. Of lesser importance, 
generally, is the effect of the ground on the 
current distribution on the antenna and on the 
tank. 

An electromagnetic wave incident on a sur¬ 
face of finite conductivity and dielectric con¬ 
stant is ordinarily reflected with a change in 
magnitude and phase and possibly a change in 
polarization. The wave received at any point in 
space from an antenna near the earth’s surface 
will be the vector sum of a direct wave plus a 
wave reflected from the surface. Because of 
the change in phase on reflection and because 
of the difference in path traversed, the phase 
difference between the direct and reflected 
waves at a point in space will depend on the 
relation of the point to the antenna. 

An accurate simulation of the constants of 
the ground could be obtained for model mea¬ 
surements by using a model ground constructed 
of a material whose dielectric constant equals 
that of the earth and whose conductivity is in¬ 
creased by the factor by which the dimensions 
in the model are reduced. Suitable materials 
of these characteristics were not readily avail¬ 
able, although there was a possibility of ob¬ 
taining them by loading rubber with a large 
amount of carbon. Mechanical difficulties in 
the model equipment made it desirable to ob¬ 
tain the patterns by other means if possible. 

The antennas of principal interest operated 
at frequencies sufficiently high so that it could 
reasonably be assumed that the presence of 



214 


ANTENNA PATTERNS FOR AIRCRAFT 


the ground had only a negligible effect in de¬ 
termining the current distribution on the an¬ 
tenna and on the tank. If, therefore, the model 
tank is suspended in free space, it can be as- 



Figure 14. Top view of tank showing locations of 
antennas. 


sumed that the current distribution is un¬ 
changed. It thus becomes possible to measure 
the pattern of the current distribution in free 
space. From theoretical considerations, an es¬ 


timate of the pattern including the effect of 
the ground can then be made. 

Free-space pattern measurements were made 
on a stub-type antenna mounted on a medium 
tank in two positions (positions marked A and 
C in Figure 14). Photographs of three-dimen¬ 
sional models of the patterns appear in Fig¬ 
ures 15 and 16. The patterns in Figure 16 
show the influence of the position of the turret 
on the pattern, since the turret is not located 
symmetrically on the tank. The frequency used 
in these measurements was such as to make the 
height of the tank about 5A. 



Figure 15. Three-dimensional pattern of antenna 
mounted on small turret of tank. 


The free-space patterns can be modified to 
obtain an approximation to the true pattern 
including the effect of ground reflections. An 
examination of the pattern in Figure 15 shows 
that there is very little energy radiated at 
angles more than about 20° below the horizon, 
owing to the large surfaces of the tank body. 
Only those waves included in the region from 




















































PATTERNS FOR A CURVED ANTENNA ON A MEDIUM TANK 


215 


the horizon to 20° below the horizon can there¬ 
fore be expected to influence the pattern after 
reflection from the ground. Since the angle 


erage magnitude of the signal in the region 0° 
to 20° in the measured free-space pattern by 
the pattern for the corresponding region for 



Figure 16. Patterns for antenna mounted on large turret. A, guns pointing to rear; B, guns pointing to right; 
C, guns pointing left; D, guns pointing forward. 


of reflection is equal to the angle of incidence, 
only the pattern in the region up to 20° above 
the horizon will be distorted by the ground re- 



Figure 17. Pattern of vertical dipole approx¬ 
imately 5\ above average ground. 


flections. For angles greater than this, the 
measured patterns are probably about correct. 
For angles up to 20° above the horizon, the 



Figure 18. Pattern of vertical stub antenna on 
small turret. 

measured pattern must be modified to include 
the ground reflections. By multiplying the av- 


the dipole, an approximation to the true pat¬ 
tern is obtained. The calculated pattern for an 
infinitesimal dipole located 5 a above an average 
earth is shown in Figure 17. Figure 18 shows 
a pattern which has been modified in this way. 

1712 PATTERNS FOR A CURVED 
ANTENNA ON A MEDIUM TANK 

An examination of the patterns in Figures 
15 and 16 shows that there is a severe cone of 
silence along the axis of the antenna. As this 
cone of silence might be deleterious in the case 
of tank-to-plane communications, a curved an¬ 
tenna was investigated to see if the cone of 
silence could be eliminated. Measured free- 
space patterns indicated that a more uniform 
distribution of signal was obtained with this 
antenna. While there is some energy in the 
horizontally polarized component, it was not 
important enough to warrant measurement. 
The presence of the ground will have about the 
same effect on the pattern of this antenna as 
on the pattern for the stub discussed above. 
The exact shape of the antenna to produce the 
uniform distribution of signal is not too im¬ 
portant, provided only that an appreciable com¬ 
ponent of the axis of the antenna is horizontal. 













216 


ANTENNA PATTERNS FOR AIRCRAFT 


1713 IMPEDANCE MEASUREMENTS BY 
MEANS OF MODELS 

In designing an antenna for a specific appli¬ 
cation there are generally two electrical factors 
which have to be considered, the pattern and 
the impedance characteristics. For many appli¬ 
cations it is possible to design the antenna to 
produce the desired pattern and to accept what¬ 
ever impedance characteristics result. There 
are many applications, particularly broad-band 
antennas, where it is not permissible to choose 
the one characteristic independently. To be 
able to correlate the pattern with the impe¬ 
dance characteristics it was felt desirable to 
attempt measurement of antenna impedances 
by means of models. After a review of the 
principal methods described in the literature 
for measuring impedances at ultra-high fre¬ 
quencies, the standing - wave method and a 
modification of Chipman’s method offered most 
promise of being adaptable to the problem. 

Equipment was constructed for making 
standing-wave measurements in the frequency 
range 600 to 3,000 me. The first crude equip¬ 
ment revealed the necessity for very precise 
mechanical construction. Deviations of the 
center conductor from the axis of the outer 
conductor as small as 0.001 in. caused very 
noticeable distortions in the standing wave pat¬ 
tern. 



X =0 X-Z 

Figure 19. Basic circuit of Chipman’s method of 
measuring impedance. 


Most of the development of methods has 
centered on Chipman’s method, for which the 
basic circuit is shown in Figure 19. A sketch 
of the equipment used is shown in Figure 20. 
The antenna whose impedance is to be mea¬ 
sured is connected to one end of a coaxial trans¬ 
mission line, and excited by a remotely located 


transmitting antenna. Varying the length of 
the line to obtain a resonance curve supplies 
data from which the unknown impedance can 
be determined. An indication proportional to 
the current in the short-circuiting plunger of 
the transmission line is obtained by means of 
a small coupling loop and transmission line to 
a detector. The plunger is driven by a microm¬ 
eter drive. A silicon detector and galvanometer 
are used for the detector. 


SCREW FOR ADJUSTING AND 
•LOCKING ROTATIONAL POSITION 
OF COUPLING LOOP 
TO VOLTMETER 

-BOLOMETER DETECTOR 


IV 


J2 


O D COAXIAL LINE 


z 


PLUNGER DRIVE ROD 


n 


I 


LENGTH DEPENDS ON 


FREQUENCY APPROX V, 


<t 

z 

z 

ID 



micrometer drive not shown 



Figure 20. Measuring equipment used in region 
A = 10 to 50 cm. 


The calibration of the equipment was car¬ 
ried out as follows. Owing to the dissymmetry 
introduced by the plunger, there is some un¬ 
certainty as to the location of the origin from 
which the length of the line is measured. A 
calibration was obtained by short-circuiting the 
antenna end of the line with a disk, and deter¬ 
mining the resonance position for the fre¬ 
quency used by feeding energy into the de¬ 
tector line. The resistance introduced by the 
detector-coupling loop (it was assumed that 







































THE RECIPROCITY BETWEEN TRANSMITTING AND RECEIVING ANTENNAS 


217 


there is no reactance introduced since the line 
is tuned) was obtained from the resonance 
curve in the usual way with no antenna con¬ 
nected to the line. Energy was fed into the 
open end of the line by means of a probe intro¬ 
duced near the open end. 

Measurements have been made of the impe¬ 
dance of a vertical rod antenna mounted on a 
large plane conductor at 750 me. Measure¬ 
ments were made of the impedance as a func¬ 
tion of the length of the antenna. The pro¬ 
cedure was simply to determine the position 
of the plunger for resonance and the breadth 
of the resonance curve at the half-power points. 

The results of the measurements are shown 
in Figures 21 and 22. A curve calculated from 



Figure 21. Resistance of cylindrical rod antenna, 
ViQ-in. diameter, X = 40.02 cm. 


the formula given by King and Blake 3 has also 
been plotted for comparison. The large devia¬ 
tion from the calculated values has been found 
by other experimenters, and so is not consid¬ 
ered serious. 


1714 THE RECIPROCITY BETWEEN 
TRANSMITTING AND RECEIVING 
ANTENNAS 

Since some radio engineers have doubts con¬ 
cerning the validity of the reciprocity theorem 
for antenna patterns, 4 it seemed desirable to 
make a rough test of it. The basis for this 
doubt lies in the feeling that known differences 
in current distribution on an antenna when 
transmitting and when receiving should lead 



Figure 22. Reactance of rod antenna of Figure 21. 


to differences in pattern. The opinion has been 
expressed that the nature of the impedance at 
the terminals of a receiving antenna might af¬ 
fect the pattern. 

A rough test of the reciprocity theorem was 
made by measuring the patterns of a number 
of antennas when receiving and when trans¬ 
mitting. In no case was any attempt made to 
keep the impedances of the generator or the 
current indicator negligible, as required by the 
reciprocity theorem as usually stated. The 
agreement between the receiving and trans- 








218 


ANTENNA PATTERNS FOR AIRCRAFT 


mitting patterns was satisfactory in all cases. 
The antennas tested included A/2 linear sym¬ 
metrical antennas, and a flat antenna mounted 
on a disk. 

The method which has been described for 
measuring antenna impedances based on Chip- 
man’s work, measures the impedance of the 
antenna when receiving. It is, therefore, im¬ 
portant to show that this impedance is the 
same as when transmitting. The proof is es¬ 
sentially that given by Franz. 5 Consider any 
antenna with an impedance Z connected to its 
terminals and under the influence of an arbi¬ 
trary incoming wave. The wave sets up a cur¬ 
rent I r in the impedance Z. Suppose now that 
a generator is inserted in series with Z to re¬ 
duce the current flowing in it to zero. The 
generator sets up a current I t which is equal 
and opposite to 7 r . By the superposition the¬ 
orem, I r is unaffected by the presence of I t , 


and vice versa. Hence, 

Ir + It = 0 (1) 


where Z a is defined as the impedance of the 
antenna when transmitting. It follows imme¬ 
diately that the receiving antenna acts like a 
generator of open-circuit voltage E g and in¬ 
ternal impedance Z a . Hence the impedance of 
a receiving antenna is identically the same as 
that when transmitting. It is apparent that 
Thevenin’s theorem applies to a receiving an¬ 
tenna. 

Note that the equality of the currents ex¬ 
pressed in the equations applies only to the 
currents in the impedance Z. It does not imply 
that the currents are equal over the entire 
antenna, since in general they will not be equal 
except at the terminals. 




Chapter 18 

AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


Study of antenna design taking into account direc¬ 
tional properties, power-handling capacities, propaga¬ 
tion at u-h-f and v-h-f frequencies, horizontal and 
vertical polarization, broad-band antennas, surface an¬ 
tennas, effect of the structure on drag, precipitation 
static, etc. The summary which follows is condensed 
from the contractor’s final report, 1 the chief reduction 
consisting in the elimination of a great number of field 
patterns, drawings, and bibliographical references. 

181 INTRODUCTION 

T he object of this project 51 was to “study 
optimum radiation patterns for use on or 
in aircraft in the v-h-f and u-h-f ranges from a 
communication point of view, taking into ac¬ 
count the variety of attitudes in which an air¬ 
plane may be when communication may be nec¬ 
essary, to study designs of antennas required 
to realize such optimum radiation patterns, and 
to include consideration of aerodynamic aspects 
to the end of attaining antennas presenting 
minimum drag.” The work resulted in a report 1 
containing as complete information as could be 
obtained within the specified time. 

In view of the extremely general nature of 
these specifications and the short time allowed, 
it was not considered feasible to undertake 
special experimental or theoretical work. 
Therefore, this summary and the final report 1 
from which it is condensed are necessarily 
based upon work done for or by the various 
agencies of the Armed Services. The bulk of the 
work was performed by: 

1. The Antenna Section, Research Division, 
Aircraft Radio Laboratory, Wright Field, 
Dayton, Ohio. 

2. The Radio Test Department, U. S. Naval 
Air Station, Patuxent River, Maryland. 

3. The Robinson Laboratory, Ohio State 
University Research Foundation. 

4. The Radio Research Laboratory [RRL], 
Harvard University. 


a Project 13-105, Contract No. OEMsr-1396, Radio 
Corporation of America. 


5. The Radiation Laboratory [RL], Massa¬ 
chusetts Institute of Technology. 

6. The Bell Telephone Laboratories [BTL], 
New York City and Deal, New Jersey. 

7. RCA Laboratories [RCAL], Rocky Point, 
Long Island. 

182 GENERAL CONSIDERATIONS IN 
AIRBORNE ANTENNA DESIGN 

Since it is a rare antenna installation in 
which the transmitter or receiver can be lo¬ 
cated directly at the antenna terminals, the 
problem is usually complicated by the presence 
of a transmission line. In practice, with low- 
loss transmission lines of essentially real char¬ 
acteristic impedance, the power transfer prob¬ 
lem is solved by so designing the transmitter 
and the antenna that their respective input 
impedances are resistive and equal in value to 
the characteristic impedance of the line. Under 
these conditions the line is said to be “flat” or 
“matched,” the energy delivered to the line 
passing down the line in the form of a travel¬ 
ing wave, which, on reaching the antenna, is 
entirely absorbed and radiated into space. 

When the antenna is not matched to the 
transmission line, the incident voltage wave is 
reflected at the antenna terminals with a 
change in magnitude and a shift in phase de¬ 
termined by the input impedance of the an¬ 
tenna relative to the characteristic impedance 
of the line. The effect of such reflection is to 
set up a system of standing waves on the line, 
the characteristics of which are described, for 
engineering purposes, in terms of two equiva¬ 
lent quantities: the magnitude of the reflection 
coefficient \K\, and the standing wave ratio 
SJFR. These are defined in terms of the termi¬ 
nating impedance and the voltage distribution 
on the line by the following expressions: 


where Z t is the terminating (or antenna im- 



219 




220 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


pedance) and Z 0 is the characteristic impedance 
of the line, 

and SWR — Emax/ E m i n > (2) 

where E max and E min are the relative magni¬ 
tudes of the maximum and minimum voltages 
in the standing wave system. 

The two quantities \K\ and SWR are related 
by the expressions: 


1*1 


SWR — 1 
SWR + 1 


and 


1 + 1*1 

1 - | K\' 


(3) 


Resonant Lines 


It is evident from the general transmission 
line equation (for lines of real Z 0 and negligible 
attenuation) 


Zin 


7 jZp tan 6 + Z t 
0 Z 0 + jZ t tan 6 


(4) 


that while the input impedance Z in of a flat or 
matched line is independent of 6, the electrical 
length of the line, and always equal to the 
characteristic impedance of the line, the input 
impedance of a mismatched, or resonant, line 
is quite definitely a function of line length. If, 
at a given frequency, an antenna is mismatched 
to an extent described by a given SWR, the 
input impedance of the line is determined by 
the line length through equation (4), the re¬ 
sistive and reactive components R and X of 
the impedance assuming any values satisfying 
the equation 


(R 2 + X2 + Z 0 2) (SWR* + 1) 

RZ 0 SWR 

which is that of a circle (of constant SWi?) 
in the complex impedance ( R-X ) plane. 

The effect of this dependence of input impe¬ 
dance upon line length is that while a given 
transmitter may be adjusted to work directly 
into a given mismatched antenna over a range 
of frequencies, the same transmitter may re¬ 
fuse to work into a low-loss line terminated in 
the same antenna. It is necessary to minimize 
the SWR by matching the antenna, if it is 
desired that a transmitter deliver energy to 
the line over an appreciable range in frequen¬ 
cies without special adjustment, the range and 
rate of variation of input impedance being 


greater the greater the SWR and the longer 
the line. 


18 2,2 Effect of S WR on Line Voltage 

For a given power delivered to the antenna 
terminals, the maximum line voltage is greater 
the greater the SWR. The greater the voltage 
the greater the possibility of line failure due 
to arc-over, particularly at connectors and other 
discontinuities in the line. Dielectric losses, 
other things being equal, are proportional to 
the square of the line voltage; the presence of 
high SWR on a solid-dielectric line is often in¬ 
dicated by local heating effects in the vicinity 
of voltage maxima, particularly near the trans¬ 
mitter end of the line. Furthermore if the line 
voltage is high and a voltage maximum hap¬ 
pens to fall at a line discontinuity the effect of 
that discontinuity will be greater the higher 
the SWR; many an otherwise satisfactory an¬ 
tenna system has been impaired by the unfor¬ 
tunate location of a cable connector with re¬ 
spect to the standing wave system. Figure 1 



Figure 1. Effect of standing wave ratio (STF#) 
on power transfer into lossless line. 


shows the effect of SfFR on power transfer 
at the antenna terminals, for the mythical case 
of a lossless line. 


Transmission Line Losses 

Losses in the polythene-filled flexible coaxial 
cable ordinarily used in aircraft antenna instal¬ 
lations are of two general types: resistive or 
“skin-effect” losses in the cable conductors, and 
dielectric losses in the polythene. These losses 









CONSIDERATIONS IN AIRBORNE ANTENNA DESIGN 


221 


introduce attenuation according to the follow¬ 
ing expressions: 


and 


<xc = 13.6 


d_ 

X 



VT~ 
log' / 

a 


a D 


27.3 e tan 8 
X 


(6) 

( 7 ) 


where ac is the attenuation in db per cm due to 
conductor losses. 

an is the attenuation in db per cm due to 
dielectric losses. 
d is the skindepth in cm. 

X is the wavelength in air in cm. 
b is the radius of the dielectric in cm. 
a is the radius of the inner conductor in cm. 
e is the dielectric constant of the cable 
dielectric. 

tan 8 is the loss factor of the cable dielectric. 


18.2.4 Transmitting Antenna Characteristics 

Since the weight and power capacity of air¬ 
borne transmitters are severely limited, the 
necessity for a low &IFR on the transmission 
line imposes rather rigid restrictions on the 
characteristics of the transmitting antenna. 
For the antenna to be efficiently matched to the 
characteristic impedance of the line over a wide 
frequency band, its impedance must be char¬ 
acterized by a resistance of the order of the line 
impedance, and by low and not too rapidly vary¬ 
ing reactance. These conditions are most easily 
met in practice by antennas worked against 
the skin of the plane as ground, and operated in 
the vicinity of a/ 4 resonance. 5 Nonresonant 
antennas, much less than a/ 4 in length, do not 
make efficient transmitting antennas. 


The sum of these two attenuations is the total 
attenuation (a?.) of thejine. Since skin depth 
varies inversely as v77 conductor losses in¬ 
crease as v77 dielectric losses increase directly 
as the frequency. Therefore conductor losses 
are more important at low frequencies, dielec¬ 
tric losses becoming more serious at high 
frequencies; this effect is shown in the follow¬ 
ing table applying to RG-14/U coax, a medium- 
power cable in common use in aircraft radio. 


Attenuation in RG-14/U cable 


Frequency 




in me 

ac in db/100 ft 

aD in db/100 ft 

ccrin db/100 ft 

100 

1.04 

0.21 

1.35 

3,000 

5.66 

6.23 

11.9 


The effect of line attenuation on overall effi¬ 
ciency may be made evident by the fact that as 
little as 25 feet of RG-14/U cable has sufficient 
attenuation at 3,000 me to reduce the maximum 
efficiency to less than 50 per cent—even if the 
antenna and the transmitter are perfectly 
matched. For this reason the use of appreciable 
lengths of solid dielectric cable is avoided at 
frequencies in the upper u-h-f range, wave 
guides being used instead if efficiency is re¬ 
quired. 


i8.~.o Electrically Short Transmitting 
Antennas 

Vertical stub antennas worked against 
ground and less than a/ 4 long have low radia¬ 
tion resistance and large capacitative reactance, 
the latter rapidly varying with frequency. If it 
were desired to use an antenna only 0.04 x long 
for transmitting purposes, the stub, for a 
length/diameter ratio of 50:1, would have a re¬ 
sistance of about 1 ohm and a reactance of 
— 1,000 ohms. While such an impedance could 
be matched to a 50-ohm line by means of a two- 
element transmission-line matching section at 
very high frequencies (where there is no par¬ 
ticular point in using such a small antenna), at 
lower frequencies it could be matched in a prac¬ 
tical way only by a matching section of lumped 
impedances. An L section would perform the 
double function of tuning out the 1,000 ohms 
capacitative reactance by means of a loading 
coil and of stepping up the 1-ohm resistance to 
look like 50 ohms. While such a matching sec¬ 
tion matches the antenna to the line, at the spot 
frequency in question, the frequency band over 
which the SWR is less than some reasonable 
maximum, say 2:1, is extremely small. Further- 


b Throughout this report the symbol X is used for 
wavelength; thus a X/4 antenna indicates an antenna 
one-quarter wavelength long. 


















222 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


more, since the loading coil must necessarily 
have resistance (a Q of 250 has been assumed) 
only a fraction of the power entering the 
matching section will actually reach the an¬ 
tenna. In this example the transmitting sys¬ 
tem has a power efficiency that is at most 20 
per cent, and even that small figure neglects 
transmission line losses and losses due to the 
ohmic resistance of the antenna and its ground 
system. 

18 2 6 Characteristics of Good Transmitting 
Antennas 

Efficient transmitting antennas are realized 
on aircraft in the form of A/4 antennas worked 
against the metal skin of the ship, or antennas 
of a/2 dipole-type suspended in space from the 
structural members of the plane. Even surface 
antennas, i.e., antennas mounted inside the skin 
of the plane and radiating through the aper¬ 
tures of slots, horns, and cavities, have critical 
dimensions of the order of A/2 or more. Such 
antennas have high input resistance, of the 
order of the characteristic impedance of the 
feed line, and small reactance which is a rela¬ 
tively slowly varying function of frequency. 

18.2.7 Antenna Impedance Measurements 

Since the actual impedance characteristics of 
a practical v-h-f or u-h-f aircraft antenna 
rarely have more than a slight resemblance to 
theoretical impedance data, it is almost always 
necessary to determine these characteristics by 
actual measurement, if optimum performance 
is desired. This is particularly true with air¬ 
craft antennas for the lower v-h-f range and 
for any antenna located on surfaces of curva¬ 
ture comparable to the operating wavelength 
or near reflecting and resonating structures. 
In such cases the antenna impedance should be 
measured under conditions as nearly identical 
as possible with those under which the antenna 
is to be used in practice. The most satisfactory 
procedure, as far as results are concerned, is to 
conduct these measurements on the full-scale 
ship—in flight, if necessary—with the antenna 
complete in all details of its final mounting and 


feeding system. Where this is impractical, a 
poor second-best procedure is to measure the 
antenna impedance by means of models, a 1/n- 
scale model of the antenna being installed in 
proper location on a 1/w-scale model of the 
plane and its impedance measured at n times 
the actual full-scale frequencies. This method is 
capable of good results only if great care is 
, taken in scaling all details of the antenna and 
its mounting and feed system. 

At higher frequencies, or in general for any 
aircraft antenna mounted on or in an airplane 
surface which constitutes a good approxima¬ 
tion to a flat ground plane for at least A in all 
directions, satisfactory results may be obtained 
by means of impedance measurements made 
with the antenna worked against a ground 
plane in an ordinary laboratory setup. But in 
all cases an effort should be made to ensure that 
the antenna is studied under conditions which 
closely approximate actual flight conditions. 

18 2 8 Antenna Impedance Matching 

While satisfactory antennas for some pur¬ 
poses can be realized without knowledge of the 
antenna impedance, by trial-and-error adjust¬ 
ment of tuning stubs and simple matching sec¬ 
tions, modern methods of impedance matching 
presume a knowledge of the impedance char¬ 
acteristics of the antenna. These methods go far 
beyond the simple a/4 transformers and shunt 
tuners described in texts and other published 
literature. 

18.2.9 The R ece i v i n g Antenna Problem 

The receiving antenna problem is different 
from that of the transmitting antenna problem 
in that the former is not so much concerned 
with power transfer but with the attainment of 
a high signal-to-noise ratio. This end can be 
approached in a twofold manner, by increasing 
the received signal strength and by reducing 
noise. 

Noise may be picked up by the antenna along 
with the signal or may be generated in the re¬ 
ceiver itself. In the upper h-f and lower v-h-f 
ranges antenna noise may be large compared to 



CONSIDERATIONS IN AIRBORNE ANTENNA DESIGN 


223 


that developed in the receiving circuit, but as 
the frequency increases antenna noise de¬ 
creases, until at frequencies greater than 70 or 
80 me it is negligible compared to set noises. 
As far as the receiver proper is concerned, the 
attainment of high signal-to-noise ratio at 
ultra-high frequencies is largely a matter of 
reducing tube and circuit noise in a closer and 
closer approach to the limits set by thermal 
agitation. 

There are many other sources of noise in air¬ 
craft radio reception since not only the antenna 
but the skin of the ship itself are parts of the 
receiving system. If poor electrical contacts 
exist anywhere in this system, the vibration 
associated with normal flight is likely to result 
in relative motion of the adjacent conductors 
at such contacts, which motion will appear in 
the receiver as noise. Hence the necessity for 
“bonding.” Shielding is useful in reducing 
static of local origin. Antenna design features 
tending to minimize precipitation static are 
discussed elsewhere in this report. 



FREQUENCY IN MC 


Figure 2. Useful received power R per unit field 
strength as function of frequency for X/4 antennas. 

18.2.10 R ece i v i n g Antenna Efficiency as a 
Function of Frequency 

The useful power delivered to a receiver by a 
matched resonant antenna varies inversely as 


/ 2 , a fact which may severely limit the range of 
u-h-f communication. This relation is shown in 
Figure 2. While the received signal may be in¬ 
creased manyfold by me&ns of arrays, horns, 
or reflectors, increased gain implies increased 
directivity, and extreme directivity is not usu¬ 
ally a desirable feature in aircraft communica¬ 
tion antennas. Furthermore, since the gain of 
a directive system is roughly proportional to its 
aperture area in square wavelengths, it is evi¬ 
dent that, except at very high frequencies, a 
practical limit to the gain of an aircraft an¬ 
tenna is quickly reached. 

18.2.11 j m p e( j ance Matching of Receiving 

Antennas 

• 

The effect of antenna mismatch is much less 
serious in receiving than in transmitting sys¬ 
tems. Receivers may be designed to have input 
impedances equal to the characteristic impe¬ 
dance of the feed line over very wide frequency 
bands, and in such cases, even though the an¬ 
tenna may be very badly mismatched, there will 
be no standing waves on the line. The effect of 
mismatch is to reduce the signal reaching the 
receiver terminal, the loss in received signal 
voltage being a slowly increasing function of 
the degree of mismatch until the mismatch be¬ 
comes quite large. Thus an antenna system 
which would be quite impossible for efficient 
transmission may well be very satisfactory for 
reception. For this reason the design standards 
for receiving antennas are usually much lower 
than those for transmitting, a simple stub or 
whip much shorter than a A/4 often making a 
satisfactory antenna if the field strength is 
sufficiently high. 

18.2.12 Effect of Line Losses on Line 

Input Impedance 

The effect of line attenuation is to reduce 
the magnitude of the variation in input impe¬ 
dance of a mismatched line, with a resulting re¬ 
duction in the apparent reflection coefficient or 
SWR looking into the line from the receiver or 
transmitter terminals. Thus the longer the line, 
and the greater its attenuation per unit length, 





224 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


the flatter its input characteristic for a given 
degree of mismatch at the antenna. The effect 
of these losses is to make the loading of a trans¬ 
mitter or the tuning of a receiver a less critical 
function of frequency. 

18.2.13 R ece ption of Very Weak Signals 

In the u-h-f range, where the received signal 
is often not much greater than the receiver 
noise level, it may sometimes be found that a 
mismatch at the receiver will result in increased 
sensitivity. In such cases it is desirable that the 


directions in space. Unless certain fairly rigid 
conditions are met, there is usually not much 
resemblance between the actual field pattern of 
a given antenna on aircraft and that of the 
same antenna in free space or worked against 
a flat infinite ground plane. Except for special 
cases, which are most commonly met in prac¬ 
tice only at u-h-f frequencies, it is necessary to 
demonstrate, by actual measurement, that the 
field pattern of a given installation is satisfac¬ 
tory for the application at hand. 

A classic example of the absolute necessity 
for field pattern measurements is shown in 
Figure 3 where the diagrams represent the 


horizontal plane 

AHEAD 




1 

KM I 


ASTERN 


VERTICAL FORE-AND-AFT PLANE 



o 

< 

UJ 


Figure 3. Two wires strung from sides of fuselage to tips of horizontal stabilizer, fed out of phase, horizon¬ 
tal polarization, 100 me. 


receiving antenna be matched to the line, for 
otherwise standing waves can exist on the line, 
with multiple reflections resulting in multiple 
signals if line losses are low. 


18.2.14 Radiation Characteristics of 
Airborne Antennas 

In aircraft antenna design, particularly in 
the v-h-f range, it is necessary for the antenna 
to radiate or pick up energy in the desired 


measured horizontal plane and vertical fore- 
and-aft-plane patterns of a 100-mc V-antenna 
installed on a 4FU. The antenna consisted of 
two wires strung outward from opposite sides 
of the fuselage, aft near the tail, to the tips of 
the horizontal stabilizer. The antenna was used, 
with rather disastrous results, at the start of 
the war in connection with an application which 
requires a pattern having a maximum or lobe 
in a generally forward and downward direc¬ 
tion. This antenna actually had nulls in the im¬ 
portant directions. 















V-H-F AND U-H-F PROPAGATION 


225 


18 3 V-H-F AND U-H-F PROPAGATION 

Plane-to-plane and plane-to-ground commu¬ 
nication in the v-h-f and u-h-f bands is de¬ 
pendent almost entirely upon space-wave 
propagation, except for anomalies occurring 
under certain atmospheric conditions in which 
a sky-wave effect is introduced by “reflection” 
at the discontinuity between air masses of dif¬ 
ferent synoptic properties. 

1831 The Space Wave 

The space wave is not a single wave but 
rather the resultant of a direct or line-of-sight 
wave and a wave reflected by the ground. The 
direct wave is not direct in a strict geometri¬ 
cal sense, owing to refraction in the atmos¬ 
phere and to diffraction around the bulge of 
the earth and around other obstacles. Under 
ordinary conditions the direct-wave field in¬ 
tensity is subject to little more than the in¬ 
verse distance law attenuation of free space. 

The ground-reflected wave is subject to all 
the laws of optical reflection and is ordinarily 
subject to greater attenuation than the direct 
wave, since the former must necessarily travel 
a longer path. For this reason, and because 
the magnitude of the reflection coefficient is 
less than unity except at grazing incidence, the 
amplitude of the ground-reflected wave at the 
receiver is less than that of the direct wave. 
Since the space wave is the resultant of two 
waves of different amplitudes and different 
phases, it is evident that it is a complicated 
function of elevation, distance, frequency, and 
polarization. 


Ground-Reflection Coefficients 

The nature of the ground-reflected wave is 
determined by the ground-reflection coefficient, 
which is different for vertical and horizontal 
polarizations. For horizontal polarization (the 
electric vector normal to the plane of inci¬ 
dence) the magnitude of the reflection coeffi¬ 
cient drops steadily from unity to a smaller 
final value as the angle of incidence varies 
from grazing to normal incidence, while the 


phase of the reflection coefficient remains at 
substantially 180° for all angles from grazing 
incidence to normal incidence. For vertical 
polarization (the electric vector in the plane 
of incidence) the magnitude of the reflection 
coefficient drops rapidly from unity at grazing 
incidence to a small value at a small angle to 
the horizon, rising gradually with increasing 
angle to approximately equal to the value for 
horizontal polarization at normal incidence. 
Meanwhile the phase shift for vertical polariza¬ 
tion decreases rapidly from 180° at grazing 
incidence to 90° at the angle of minimum re¬ 
flection, finally becoming 0° for angles between 
the angle of minimum reflection and normal 
incidence. 

Therefore the received signal in aircraft 
communication will vary with distance, eleva¬ 
tion, and frequency. 

18 ' 3,3 Effect of Distance 

At distances small compared to the antenna 
heights the resultant space-wave amplitude os¬ 
cillates about its normal free-space value as 
transmission distance increases, since both the 
phase difference between the two component 
waves due to their different path lengths and 
the phase difference due to the fact that the 
ground-reflection coefficient is a function of 
angle of incidence depend upon distance. With 
increasing distance these oscillations develop 
larger amplitudes, since the amplitudes of the 
two component waves approach equality as 
their path lengths become more nearly equal 
and since the magnitude of the ground-reflec¬ 
tion coefficient approaches unity at grazing in¬ 
cidence. But while the amplitude of the oscilla¬ 
tions increase, their frequency decreases with 
distance, since the increment in distance for a 
given phase difference becomes greater with 
increasing distance. 

At distances large compared with the an¬ 
tenna heights, corresponding to ground reflec¬ 
tion at grazing incidence, but still above the 
line-of-sight horizon, the received signal is no 
longer oscillatory, but obeys almost exactly the 
inverse-square law. 

At still larger distances the component waves 
approach phase opposition and the resultant 



226 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


field drops off more rapidly than the normal 
free-space field as the distance increases be¬ 
yond the optical horizon. The phenomena de¬ 
scribed above are very similar for both hori¬ 
zontal and vertical polarization, except in the 
near region in which the distance is of the 
same order of magnitude as the elevation of 
the antennas; in this region the oscillatory re¬ 
sultant field is more complex for vertical polar¬ 
ization owing to the greater sensitivity of the 
vertical reflection coefficient to angle of inci¬ 
dence, when that angle is large. 

183,4 Effect of Elevation 

The effect of elevation is quite similar to 
that of distance in the near region in which 
the height is not negligible compared to the 
transmission distance; the same oscillations in 
received signal occur in the field strength ver¬ 
sus height curve as in the field strength versus 
distance curve, and for the same reasons. 

At greater distances, beyond the oscillatory 
region, the field strength is almost proportional 
to the product of the antenna heights. Below 
the optical horizon the field strength is very 
sensitive to antenna height. For low heights 
field strength is at first independent of height, 
then increases with altitude at an accelerated 
rate until it is directly proportional to height. 
At very great elevations (but still below the 
optical horizon) field strength increases more 
rapidly with height than as the product of the 
antenna heights. 

18 3 5 Effect of Frequency 

As far as the oscillatory region is concerned 
the effect of increasing frequency is to increase 
the number of oscillations per unit distance 
and to extend the distance over which oscilla¬ 
tions occur. This effect is particularly pro¬ 
nounced with horizontal polarization over 
ground, the extent of the oscillatory region 
increasing from approximately 5 to approxi¬ 
mately 100 miles as the frequency is increased 
from 30 to 600 me (for the case of communica¬ 
tion between a ground station and a plane at 
40,000 ft altitude). 


At greater distances, but still above the hori¬ 
zon, an increase in frequency results in an 
increase in field strength for horizontal polar¬ 
ization; for vertical polarization there is no 
significant change. Below the optical horizon 
(into which region the space wave extends by 
virtue of refraction and diffraction) the field 
strength is less the higher the frequency, par¬ 
ticularly for vertical polarization. 

For all of these reasons dependable v-h-f and 
u-h-f communication is restricted to stations 
above each other's optical horizon. Although 
this situation precludes the possibility of ex¬ 
tremely long-range communication, it is not 
nearly so severe a restriction as might be 
expected, since plane-to-plane and plane-to- 
ground communication can extend over quite 
respectable distances. 

Figures 4* and 5 summarize the effects of 
frequency, polarization, distance, elevation, and 
nature of the ground upon high-frequency 
propagation. While these curves are based on 
calculations for ideal short doublet antennas 
and take no account of the field pattern of an 
actual aircraft antenna, they are valuable in 
that they give a qualitative picture of how the 
controlling factors affect aircraft communica¬ 
tions. 



Figure 4. Range versus frequency, horizontal po¬ 
larization for 100 mv per meter field strength and 
10 w radiated power, one antenna 10 ft off ground, 
the other at altitudes indicated, short doublet an¬ 
tennas. (Data from Bell Telephone Laboratories.) 

is.3.6 Polarization and Propagation 

Figures 4 and 5, and the preceding discus¬ 
sion indicate little difference between vertical 
and horizontal polarization, as far as propa- 







227 


V-H-F AND U-H-F PROPAGATION 


gation over land is concerned. In the near 
region vertical polarization is preferable, since 
the field strength is greater and the magnitude 
of the oscillations less pronounced. Vertical 
polarization has better transmission over sea 
water and moist soil at the lower frequencies. 



Figure 5. Range versus frequency, vertical po¬ 
larization, for 100 mv per meter field strength and 
10 w radiated power, one antenna 10 ft off ground, 
the other at altitudes indicated, short doublet an¬ 
tennas. (Data from Bell Telephone Laboratories.) 

Anomalous Effects at High 
Frequencies 

Tropospheric Reflection 

Since the velocity of propagation of u-h-f 
radio waves in air depends upon the dielectric 
constant of the atmosphere, which in turn is 
a function of pressure, temperature, and hu¬ 
midity, it is natural that there be a correlation 
between anomalous propagation and the pas¬ 
sage of meteorological “fronts,” and with the 
existence in the upper atmosphere of any ab¬ 
normal distribution of temperature and hu¬ 
midity. Under such conditions, as the radio ray 
passes into the region of different electrical 
properties it will be refracted, perhaps suffi¬ 
ciently to return to earth, giving rise to what 
are known as tropospheric reflections, although 
they are not reflections in the strict optical 
sense, since the discontinuity is not sharply 
defined. 

Trapping or Waveguide Effect 

Such reflections may make communication 
possible over much greater distances than are 


ordinarily attained. If the discontinuity layer 
is sufficiently pronounced the radiation may 
be effectively trapped at the top of the in¬ 
version layer, the region between this layer 
and the surface of the earth acting somewhat 
like a waveguide having large attenuation. 
While this waveguide effect may be helpful in 
making extremely long distance communica¬ 
tion possible, it may also be a liability at 
smaller ranges, depending upon the height of 
the inversion in the stratified atmosphere, due 
to interference between the ordinary space 
wave and this pseudo sky wave. 

Fading at U-H-F 

This interference results in fading, which 
may vary with time as well as with altitude 
and distance, due to the relative motion of the 
two air masses resulting in shifting of the posi¬ 
tion of the discontinuity layer. 

Fading and trapping are generally more pro¬ 
nounced the higher the frequency, partly be¬ 
cause directive antennas are usually used for 
both transmission and reception at the higher 
frequencies; and since the sharper the radio 
beam the greater the fraction of energy re¬ 
flected by the discontinuities, and the more 
apparent their effects. 

At present there is no conclusive evidence 
that either polarization is less affected by at¬ 
mospheric disturbances. 

Static at V-H-F and U-H-F 

The higher frequencies are much less af¬ 
fected by static of natural origin than are low. 
Due to the absence of a sky wave under normal 
conditions, u-h-f communication is less suscep¬ 
tible to static from distant sources. Since the 
field strength of static is approximately pro¬ 
portional to wavelength, static of local origin 
is less effective the higher the frequency. 


Man-Made Interference 

Noise generated in rotating machinery and 
other sources of interference seems to be pre- 










228 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


dominately vertically polarized; consequently 
such interference is generally worse for verti¬ 
cal aircraft antennas than for horizontal. 


18 . 3.9 Multipath Interference 

Since the wavelengths corresponding to very- 
high and ultra-high frequencies are small com¬ 
pared to the dimensions of buildings, hills, etc., 
there may be multiple ground-reflected rays ; 
resulting in even more complex interference 
effects in space-wave communication than those 
discussed above. 

Furthermore, the structural members of the 
aircraft upon which the antenna is mounted 
are large enough relative to small wavelengths 
to cast shadows and cause reflection and dif¬ 
fraction effects which may interfere greatly 
with transmission and reception in certain di¬ 
rections. These effects are generally more pro¬ 
nounced the higher the frequency. 

18.3.10 Propeller Modulation 

Propeller modulation, at a frequency equal 
to the product of the number of propeller 
blades by the number of revolutions per sec¬ 
ond, will affect both transmission and recep¬ 
tion. Although such modulation can approach 
100 per cent in extreme cases, it can be mini¬ 
mized by removing the antenna from the im¬ 
mediate vicinity of the motors. 


184 FIELD PATTERNS OF ANTENNAS 
ON AIRCRAFT 

Since the radiating system formed by an 
aircraft antenna and the skin of the plane 
upon which it is mounted is generally quite 
complex, the field patterns of such a system 
are usually quite different from those of a 
similar antenna in free space or mounted on 
a flat infinite conducting plane. Except under 
certain special conditions, usually met in prac¬ 
tice only in the case of u-h-f antennas mounted 
on or in flat, unobstructed airplane surfaces 
of dimensions large in terms of wavelength, 
experience shows that the field pattern of a 


given antenna will be modified to a greater or 
less extent by the plane upon which it is used. 
To be certain that the field pattern of a given 
aircraft antenna installation satisfies the re¬ 
quirements of the problem, it is usually nec¬ 
essary to determine the field pattern experi¬ 
mentally. 

18.4.1 Flight Measurements of Field 
Patterns of Aircraft Antennas 

The most direct method for determining the 
radiational characteristics of a given antenna 
on aircraft is to install the antenna on the plane 
upon which it will be used, connect it to a trans¬ 
mitter covering the frequency range in ques¬ 
tion, and fly the plane in a definite course 
around a field-strength meter located on the 
ground. While direct, this method has dis¬ 
advantages: it is a difficult procedure, requir¬ 
ing that the plane be flown on a prescribed 
course maintaining constant speed, distance, 
and elevation; there are perhaps less than a 
dozen pilots in the country with sufficient skill 
and practice to make accurate pattern mea¬ 
surements possible. Furthermore flight mea¬ 
surements are expensive and time-consuming, 
often—in times of plane and personnel short¬ 
ages—an outright impossibility. 

The most serious objection to flight measure¬ 
ments is that at best they yield information 
about only a very small part of the total field 
pattern of an aircraft antenna. Because of the 
oscillatory nature of the space-wave field upon 
which aircraft communication depends it is 
difficult, if not impossible, to obtain meaning¬ 
ful field-strength measurements when the test 
airplane is at distances comparable with its 
elevation. The oscillatory region extends from 
5 to 100 miles from the ground station, depend¬ 
ing upon elevation, frequency, polarization, and 
ground conditions; accordingly the angular 
spread of the space pattern that can be mea¬ 
sured in flight with any pretense at accuracy is 
extremely limited—0 to 10° below the horizon 
being an optimistic range. While this range of 
elevation angles could be extended by banking 
the plane, the pattern pilot usually has enough 
to do without having to maintain his plane at 
a constant angle of tilt. The use of a second 



FIELD PATTERNS OF ANTENNAS ON AIRCRAFT 


229 


plane to replace the ground station would 
merely multiply the difficulties already present 
and would introduce the further complication 
of the directivity of the receiving antenna sys¬ 
tem on that second plane. 

For these reasons flight measurements are 
usually restricted to the determination of field 
pattern in the horizontal plane of the ship. 

18 ' 4 ' 2 Pattern Measurements by Means of 
Models 

Model measurements, based upon the prin¬ 
ciples of similitude and reciprocity, form the 
most satisfactory method for determining the 
radiation patterns of aircraft antenna systems. 
In this method an accurate 1/w-scale model of 
the plane is mounted on a nonmetallic tower 
in such a way that the model has two degrees 
of rotational freedom and is located in the 
uniform field of a pyramidal horn radiating 
energy of frequency n times that used on the 
full-scale plane. The model plane is remote 
from the horn (in terms of wavelength), the 
directivity of which is such that there is no 
danger of interference effect due to reflection 
from the ground or from nearby obstacles. A 
1/ft-scale model of the antenna is mounted 
upon the metallic surface of the plane in its 
proper position and is connected, through ap¬ 
propriate feed and matching systems, to a 
thermocouple, bolometer, crystal, or other de¬ 
tector, located inside the model, the d-c output 
of the detector being fed from the model to a 
remotely located microammeter or other indi¬ 
cating or recording device. The reading of the 
d-c instrument bears some simple relationship 
to the r-f signal received by the antenna. By 
properly orienting the model plane, whose po¬ 
sition with respect to the horn is usually re¬ 
motely controlled, it is possible to determine 
the relative field pattern of the antenna in any 
or all directions in space. 

The validity of model pattern measurements 
depends largely upon the accuracy with which 
the model and the antenna are constructed and 
scaled. It depends also upon the measuring 
equipment, particularly in regard to frequency 
stability and constancy of output of the oscil¬ 
lator and upon uniformity of the field pattern 


of the horn over the entire region in space in 
which the antenna can be situated as a result 
of the motion of the model. Aside from the 
care required in scaling the length of the an¬ 
tenna and in locating it in its proper position 
on the ship, no further precautions need be 
taken with the antenna and its associated feed 
system, in ordinary work. That is, as far as 
field patterns are concerned it is not necessary 
to scale every detail of the feed system nor to 
be sure that the antenna is matched to the 
detecting system. An exception to this general 
rule is met in the case of multiple-antenna sys¬ 
tems, in which two or more individual antennas 
are fed with currents of definite relative mag¬ 
nitude in definite phase relationship. In such 
cases it may be necessary to scale every detail 
of the antenna system exactly and to be sure 
that impedances are matched throughout the 
system. 

Although the conditions upon which the 
principle of similitude is based are not com¬ 
pletely fulfilled in that no attempt is made to 
scale either conductivity or dielectric constant, 
there is ample experimental evidence that this 
defect introduces negligible error. 

18,4,3 Calculation of Aircraft Antenna 
Patterns 

In the v-h-f range particularly, the radiating 
system formed by the antenna and the skin of 
the ship upon which it is mounted may be ex¬ 
ceedingly complex. Here the dimensions of 
structural parts of the plane are of the same 
order of magnitude as, or large compared to, 
the operating A; such structural members tend 
to become increasingly effective in casting 
shadows, in causing reflection effects, and in 
generally disturbing the resultant field pattern 
as the frequency increases. Because of the simi¬ 
larity in size between such parts of the ship’s 
structure and A it is possible that resonance 
effects will occur in tail fins, stabilizers, guns, 
and in other antennas. It is further possible 
that resonance effects may occur in the smooth 
unobstructed skin of the fuselage itself; such 
respnant surface currents may have radiation 
characteristics that entirely mask that of the 
antenna proper, the antenna functioning some- 




230 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


what like a coupling loop by means of which the 
skin of the ship is loaded. Because of all these 
possibilities and because of the geometrical 
complexity of the surfaces of modern aircraft, 
it is usually impossible to calculate the field 
pattern of a v-h-f aircraft antenna with any 
degree of accuracy. 

At very high frequencies and under certain 
conditions it is sometimes possible to make pat¬ 
tern calculations that are qualitatively useful. 
These possibilities are limited to cases in which 
the antenna is located remote from reflecting 
and resonating objects, on or near smooth clean 
surfaces of extent large compared to the operat¬ 
ing A. As an example of a pattern calculation 
under such conditions, consider Figure 6. Here 


the fin constitute semi-infinite conducting 
planes intersecting at right angles. At 450 me 
the surfaces involved are large compared to A, 
and when the antenna is close in to the side of 
the fin the angle subtended at the antenna by 
the lower edge of that surface is large, ap¬ 
proaching the 90° that would be subtended by 
the “edge” of a semi-infinite plane. It will be 
seen that measured and calculated patterns are 
in good agreement for the small spacing of A/2 
(corresponding to a subtended angle of 77°). 
At the larger spacing of 2a (subtended angle 
45°) the agreement is only qualitative in that 
both patterns have the same number of lobes in 
about the same position. At 4a (subtended 
angle only 27°) the agreement is poor. 



Figure 6. Discrepancy between calculated and measured field patterns of antenna under tail stabilizer of 
PBY. Dotted lines are calculated data; full lines as measured. 


the solid lines represent the measured vertical- 
athwart-ship-plane patterns of a horizontal a/2 
dipole suspended with its axis in the line of 
flight a/4 below the undersurface of the hori¬ 
zontal stabilizer of a PBY, at various distances 
out from the side of the vertical tail fin. The 
dotted lines represent the corresponding pat¬ 
terns calculated from simple image and antenna 
array theory, upon the assumption that the 
undersurface of the stabilizer and the side of 


18 . 4.4 Definition of Polarization 

Unlike the clear definitions of polarization 
used in optics, these concepts are used in air¬ 
craft radio engineering with considerable con¬ 
fusion. In the horizontal plane there is little 
difficulty, the electric vector of horizontally 
polarized radiation lying in the “horizontal” 
plane of the ship in normal flight; that of ver¬ 
tically polarized radiation lying normal to this 
















FIELD PATTERNS OF ANTENNAS ON AIRCRAFT 


231 


plane. In the vertical fore-and-aft plane the 
electric vector for horizontal polarization is 
always “horizontal” in that it lies parallel to the 
horizontal surfaces of the ship and is always 
in the same sense, that is, in the athwart-ship 
direction. In this plane the electric vector for 
vertical polarization is truly vertical only in 
two directions, dead ahead and dead aft. At all 
other angles of elevation in the vertical plane 
of flight “vertical” polarization has a horizontal 
component, proportional to the sine of the angle 
of elevation as measured from the horizontal 
plane. Directly above and directly below the 
ship “vertical” polarization is entirely hori¬ 
zontal in the usual geometrical sense, the direc¬ 
tion of the vector being along the line of flight. 
In the vertical athwart-ship plane, “horizon¬ 
tal” polarization is always horizontal, the elec¬ 
tric vector lying parallel to the horizontal sur¬ 
faces of the ship, its direction along the line of 
flight; but “vertical” polarization is truly verti¬ 
cal only off the port and starboard wing tips, 
the electric vector being 100 per cent horizontal 
and athwart ship, directly above and directly 
below the ship. 

There have been attempts in the past to avoid 
this confusion by the use of symbols, polariza¬ 
tion being expressed in terms of its components 
along the two angular coordinates of a spherical 
reference system. This is correct and quite un¬ 
ambiguous to people having the spherical co¬ 
ordinate system perfectly in mind at all times, 
but such persons seem to be few and far be¬ 
tween. Aircraft radio engineers continue to use 
the phrases “vertical polarization” and “hori¬ 
zontal polarization” with their customary 
promiscuity. 

18 . 4.5 Presentation of Pattern Data 

Many different methods of presenting pat¬ 
tern data in graphical form have been used in 
the past. 

The most usual procedure is to give only 
three complete polar patterns for relative field 
strength or relative power in the horizontal, the 
vertical fore-and-aft, and the vertical athwart- 
ship planes. In most cases the distribution of 
radiation in these three planes suffices to give 
a pretty fair idea of the directivity of the an¬ 
tenna system. 


Where complete information is desired, or 
where an unusual installation makes a peculiar 
field distribution probable, the complete spheri¬ 
cal pattern, covering a solid angle of 4tt, may be 
taken. Complete spherical data may be pre¬ 
sented by means of three-dimensional models, 
by means of a series of plane polar diagrams, or 
by means of stereographic projection diagrams. 
An example of the latter form of presentation 
is shown in Figure 7. It has the big advantage 
over other methods of showing at a glance just 
where the radiation is going, with all lobes and 
nulls clearly evident, rapidly changing parts of 
the field being indicated by the crowding to¬ 
gether of the constant-field-strength or con¬ 
stant-power contour lines. In the general case 
two such stereographic projections—one for 
each hemisphere—are necessary (for each 
polarization) for complete presentation of the 
data. But in the case of symmetrically located 
antennas a single diagram is sufficient for each 
polarization. 


18.4.6 Absolute versus Relative Field 
Strength Patterns 

It has been the practice in the past to present 
pattern data on a relative basis, simply in terms 
of arbitrary field strength or power units plot¬ 
ted against azimuthal or elevation angle. There 
has been some interest in the presentation of 
aircraft antenna pattern data on an absolute 
basis, in terms of millivolts per meter watt 
input power per mile, or in terms of the direc¬ 
tivity of the system with respect to an isotropic 
radiator, a Hertz doublet, or a A/2 dipole. 0 

Model measurements cun be made to yield 
absolute patterns, but the procedure involved is 
extremely tedious. 


0 If D is the power directivity referred to an isotropic 
radiator, then: 

2/3 D is power directivity with respect to a Hertz 

doublet. 

V2/3 D is field-strength directivity with respect to a 
Hertz doublet. 

0.61 D is power directivity with respect to a A/2 
dipole. 

0.78 VD is a field-strength directivity with respect to 
a A/2 dipole. 

3.40 VD is absolute field strength in millivolts per 
meter per watt per mile. 





232 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 



Figure 7. Stereographic projection field pattern, B-17 94.5-mc V antenna on tail; power distribution in star¬ 
board hemisphere. 

































































FIELD PATTERNS OF ANTENNAS ON AIRCRAFT 


233 


Similarities between Aircraft 
Antenna and Ideal Antenna Patterns 

The radiation maximum in the vertical pat¬ 
tern of a A/4 (or less than A/4) antenna worked 
against flat infinite ground is along the horizon, 
the entire field pattern of such a system being 
simply the upper half of the pattern of a verti¬ 
cal dipole in free space. This is not the case 
with ground planes of finite size. 

Carter 2 has measured the vertical plane pat¬ 
terns of a/4 antennas mounted on circular and 
rectangular ground planes of various size, and 
has found an approximately linear relation be¬ 
tween the angle of throw-up of maximum radia¬ 
tion and the logarithm of the distance in A from 
the base of the antenna to the edge of the sur¬ 
face upon which the antenna is mounted, the 


This same effect occurs in the case of stub 
and whip antennas mounted vertically on air¬ 
craft. Figure 8 shows a plot of throw-up angle 
as a function of relative distance from the an¬ 
tenna was mounted on a smooth unobstructed 
logarithmic scale, the linear relation obtained 
by Carter for simple flat ground screens being 
indicated by the straight line. The twelve points 
shown in this figure were obtained from meas¬ 
ured vertical plane patterns of stub antennas 
on aircraft in installations such that the an¬ 
tenna was mounted on a smooth unobstructed 
surface of curvature small compared to the 
operating A in the direction in which the pat¬ 
terns were run. It is evident from the close 
agreement between these two sets of data that 
many antenna installations are found in prac¬ 
tice under conditions such that the surface on 



0.4 0.5 0.6 0.0 1.0 2 3 

DISTANCE IN A TO EDGE OF GROUND SURFACE 


Figure 8. Angle of maximum radiation as function of distance from antenna to edge of surface on which an¬ 
tenna is mounted. Solid line represents experimental data for A/4 antennas on flat ground planes. 


angle of throw-up being less the larger that 
distance, the theoretical value of zero elevation 
for an infinite plane being approached very 
slowly as the size of the surface becomes large 
in terms of A. 


which the antenna is mounted is a close enough 
approach to a flat ground plane so that Carter’s 
data may be used to advantage in predicting 
the general nature of the distribution of radia¬ 
tion in the vertical planes. This simple relation 










234 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


will be quite invalid for antennas mounted on 
surfaces having pronounced curvature near the 
antenna, or for installations such that obstruc¬ 
tions like fins, motors, turrets, and other an¬ 
tennas are likely to effect the field pattern in 
the plane in question. 

A similar correspondence between aircraft 
antenna patterns and the patterns of antennas 
in ideal locations is found in the case of v-h-f 
and u-h-f horizontal antennas mounted near flat, 
horizontal surfaces on aircraft. Patterns quite 
similar to theoretical patterns are actually ob¬ 
served in the case of horizontal antennas 
mounted near a wing or fuselage surface that 
is large in terms of wavelength. In the case of 
aircraft installations, the field strength will not 
drop off to zero along the horizon because of 
the finite size of the aircraft surfaces. 

18.4.8 Patterns of Vertical Antennas 

on Aircraft 

Pattern versus Frequency 

The field patterns of a given type of antenna 
in a given location on a given plane are mark¬ 
edly sensitive to frequency. Patterns of a given 
antenna on a given plane (see Chapter 17) re¬ 
veal the following frequency-dependent phe¬ 
nomena : 

In the Horizontal Plane. At the lower fre¬ 
quencies the horizontal plane patterns are 
fairly symmetrical, the vertical members of the 
ship’s structure (for example the vertical fin of 
a B-17) being too small relative to the operat¬ 
ing a to be capable of casting sharp shadows or 
of causing pronounced reflection effects. At the 
higher frequencies these disturbing structures 
become large relative to A, and the horizontal 
patterns are then more complex. Definite 
shadow regions appear, in which the field in¬ 
tensity is small compared to the average, al¬ 
though rarely zero owing to diffraction around 
the edges of the obstacles. Other minima are 
doubtless due to destructive interference be¬ 
tween the direct ray in certain directions and 
the ray reflected from the tail-fin surfaces. 
Similarly maxima also appear in the pattern, 
resulting from constructive interference be¬ 
tween direct and reflected rays. 


In the Vertical Fore-and-Aft Plane. Even 
though the antennas (of a B-17 for example) 
may be mounted atop the fuselage over the 
wings, a great deal of radiation is found below 
the horizon at the low frequencies, at which the 
wing and fuselage surfaces are too small in 
terms of A to act as efficient screens. At the 
higher frequencies little radiation is found 
below the horizontal plane. 

At the lower frequencies the angle of maxi¬ 
mum radiation is higher than at higher fre¬ 
quencies, agreeing with measurement and 
theory in the case of stub antennas worked 
against ground planes of finite size, showing 
that the smaller the extent of the ground plane 
in terms of A the greater the angle of throw-up 
of maximum radiation. 

Another effect evident in vertical fore-and- 
aft patterns is the increasing number of lobes 
and nulls as the frequency is increased. These 
are partially explained on the basis of structure 
effects, which would naturally be more pro¬ 
nounced the higher the frequency, but would 
also appear if the surface of the plane were 
absolutely flat and unobstructed, due to the 
presence of standing surface waves on ground 
planes of finite size. 

In the Vertical Athwart-Ship Plane. The field 
patterns in this plane may be marked by similar 
effects, such as decreasing radiation on the 
opposite side of the ship from that on which 
the antenna is located, decreasing angle of 
throw-up of maximum radiation, and decreas¬ 
ing symmetry of pattern, as the frequency is 
increased. 

Pattern versus Location of Antenna 

The field pattern of a given type of antenna 
for a given frequency on a given plane, is 
greatly dependent upon the location of the an¬ 
tenna on the plane. This dependence is best 
demonstrated by reference to experimental 
results. 

Figures 9 and 10 show the field patterns of a 
100-mc a/4 stub antenna in four widely differ¬ 
ent locations on a PBY. The great effect of loca¬ 
tion is obvious in this series. The patterns for 
the installation atop the vertical stabilizer are 
of particular interest in that the effectiveness 



FIELD PATTERNS OF ANTENNAS ON AIRCRAFT 


235 


HOR I ZONTAL PLANE VERTICAL FORE-AND-AFT PLANE 



HORIZONTAL PLANE VERTICAL FORE-AND-AFT PLANE 



of a large surface (the wings) as a reflector as a screen (note the relatively large amount of 
(notice the large forward lobe in the upper downward radiation in the same pattern) are 
vertical fore-and-aft pattern) and the ineffec- both demonstrated. The horizontal plane pat- 
tiveness of a small surface (the tail stabilizer) terns of this series are interesting in that they 

f 






























236 


AIRBORNE ANTENNA DESIGN AT U-H-F AND Y-H-F 


HORIZONTAL PLANE VERTICAL FORE 'AND* AFT PLANE 



HORIZONTAL PLANE 


AHEAD 



VERTICAL FORE-AND-AFT PLANE 



Figure 10. Antenna pattern versus location. X/4 stub at 100 me located as follows: A, underside hull below 
tail, 135 cm forward of tip of tail; B, over center of cockpit. 


show the effect of slight deviations of antenna 
location from the axis of symmetry of the plane 
upon the symmetry of the horizontal pattern. 

Another unusual example of the effect of 
location on field pattern is shown in Figures 11 
and 12. The project in connection with which 


these patterns were taken involved a series of 
antennas for a B-24, the antennas to yield “uni¬ 
form” distribution of vertically polarized radia¬ 
tion in the horizontal plane and in the upper 
hemisphere for 30° to 40° above the horizon. 
At 20 to 40 me, suitable patterns could be real- 



























FIELD PATTERNS OF ANTENNAS ON AIRCRAFT 


237 


ized by means of antennas installed atop the 
fuselage, on the line-of-flight centerline, ap¬ 
proximately 12 ft forward of the leading edge 
of the horizontal stabilizer. To simplify installa¬ 
tion problems it was desired to mount the high- 
frequency antennas (40 to 60 me) in this same 
location. While the horizontal and vertical fore- 
and-aft plane patterns are satisfactory, Figure 
11 shows that the vertical athwart-ship pattern 
indicates that most of the radiation in that 


elsewhere in this report. In the case of this 
particular problem, such loading effects with 
consequent strong radiation from surface cur¬ 
rents in the skin of the fuselage were inhibited 
by moving the antenna forward to where the 
wing surfaces could interfere. 

VERTICAL ATHWART-SHIP PLANE 
UP 


VERTICAL ATHWART-SHIP PLANE 
UP 


o 

oc 

< 

o 

CD 

cr 

cn cn 
o 

CL 


h- 

cc 

o 

CL 


DOWN 


Figure 12. Moving 48-mc antenna of Figure 11 
forward along fuselage centerline to trailing edge 
of wing produced desired upward radiation instead 
of undesired downward radiation as in Figure 11. 




Figure 11. Effect of locating 40- to 60-mc antenna 
in same location as an antenna for 20 to 40 me. 
Sleeve antenna, overall length 0.26X, sleeve length 
0.125X; 48 me; vertical polarization; location verti¬ 
cal atop fuselage on centerline 11.5 ft forward of 
leading edge of horizontal stabilizer on B-24. 

plane is downward, and that there are sharp 
nulls about 20° above the horizon to either side 
of the ship. For these reasons this location for 
the h-f antennas had to be abandoned, and the 
48-mc antenna was moved forward along the 
fuselage centerline to the trailing edge of the 
wing. Now the patterns in all three planes were 
satisfactory, the radiation in the vertical 
athwart-ship plane (Figure 12) now being uni¬ 
form and upward as desired. This effect, be¬ 
lieved due to the loading of the surface of the 
cylindrical fuselage when its circumference is 
resonant, is discussed in greater detail in the 
treatment of the broad-band whip antennas 


Pattern versus Airplane 

The pattern of a given type of antenna for a 
given frequency is greatly dependent upon the 
type of aircraft upon which it is mounted. This 
is particularly true in the middle and upper 
v-h-f range. The effect of the nature of the 
plane is most pronounced when the antenna is 
worked at frequencies such that sizes of struc¬ 
tural members of the ship are of the same order 
of magnitude as the A, and when the antenna is 
located on surfaces of pronounced curvature. 
About all one can say as to the patterns of simi¬ 
lar antennas in similar locations on different 
planes, is that if the planes are much alike, dif¬ 
fering mainly in size, then the pattern char¬ 
acteristics found on one plane will be found on 
the other plane, at a higher or lower frequency, 
depending upon the relative sizes. In the gen- 














238 


AIRBORNE ANTENNA DESIGN AT U-H-F AND Y-H-F 


eral case, where v-h-f and u-h-f antennas are 
mounted on planes of widely different dimen¬ 
sions and shapes, the patterns will be greatly 
dependent upon the nature of the plane and 
upon the relation of the antenna to the predomi¬ 
nate structural features. 

1849 Effect of Nearby Structures on 
Patterns 

One of the hazards of aircraft antenna design 
is the possibility that an antenna designed to 
have certain pattern characteristics and in¬ 
stalled on the ship in question will later be 
ruined by the installation of another antenna, 
a new turret, or an auxiliary gas tank in the 
immediate vicinity of the first antenna. Unless 
the effects of such disturbing structures are 
allowed for in the design of the original an¬ 
tenna, it is well to locate any metallic object of 
size or length comparable with the antenna 
dimensions at least one wavelength from the 
antenna. 

18.4.10 (] rogg Polarization from Aircraft 
Antennas 

There is usually some of the opposite polar¬ 
ization present in the field pattern of a simple 
vertical or horizontal antenna on aircraft, par¬ 
ticularly in the lower v-h-f range. While the 
actual percentage of energy in cross polariza¬ 
tion for a given installation depends in a com¬ 
plicated manner upon the frequency, the size 
and shape of the ship, and the location and 
orientation of the antenna, it is fairly safe to 
say that under ordinary circumstances it is 
small compared to the normally polarized radia¬ 
tion. 

Under certain conditions, when the currents 
in the skin of the plane are properly disposed, 
the amount of cross polarization may be much 
greater. Also, the amount of cross polarization 
may be greater in other planes than the hori¬ 
zontal. While the presence of relatively large 
amounts of horizontal polarization is found 
with vertical antennas in particular installa¬ 
tions, the phenomenon is not one upon which 
one may count in general; that is to say, it is 


ill-advised to expect to receive or transmit hori¬ 
zontally polarized radiation efficiently with a 
vertical antenna. 

Cross polarization is usually more pro¬ 
nounced with horizontal antennas than with 
vertical, particularly in the h-f and lower v-h-f 
ranges. Here two factors are at work: (1) part 
of the vertical radiation may be due to loading 
of the skin of the ship or to resonance effects in 
vertical surfaces of the structure of the plane; 
and (2) at very low frequencies a horizontal 
antenna may have an appreciable vertical com¬ 
ponent, which, although small compared to the 
total antenna length, may be a relatively much 
more efficient radiator than the horizontal com¬ 
ponent, especially since the horizontal antenna 
current tends to be inhibited in its radiational 
effects by the presence of an opposite image 
current in the nearby surface of the ship. 


18411 Conclusion 

Most of the pattern problems discussed above 
are serious only in the v-h-f range. At higher 
frequencies the pattern of a given antenna 
worked against a large clean surface on air¬ 
craft, will be quite similar to the pattern on an 
infinite ground plane, except, of course, in cases 
where obstructions exist in the near field of the 
antenna. The effect of disturbing factors is then 
more pronounced the higher the frequency. 


18 5 ANTENNAS FOR VERTICAL 
POLARIZATION 

Compared to the problem of obtaining good 
aircraft antennas for horizontal polarization 
the design of vertical antennas for the v-h-f and 
u-h-f ranges is relatively easy. Not only is it 
easier to secure good input-impedance char¬ 
acteristics consistent with satisfactory me¬ 
chanical and aerodynamical features, but, be¬ 
cause of the symmetry inherent in the field of 
simple vertical antennas and because of the 
wide variety of locations in which they may be 
mounted on a plane, the problem of obtaining 
satisfactory radiational characteristics on air¬ 
craft is also much less difficult. 



ANTENNAS FOR VERTICAL POLARIZATION 


239 


Broad-Band Antennas 

The design of antennas having flat impedance 
characteristics is easier the higher the fre¬ 
quency. There are two reason for this; (1) the 
higher the frequency the smaller the physical 
dimensions of an antenna structure large in 
terms of A and the less difficult the problems of 
mechanical and aerodynamical design, and (2) 
the higher the frequency, the smaller the an¬ 
tenna, and the less difficult the problem of feed¬ 
ing the antenna structure in a manner which 
will not impair its inherent broad-band char¬ 
acteristics. For these reasons antennas having 
band widths of the order of several octaves are 
practical in the u-h-f range, while in the lower 
v-h-f range it is a triumph of design to obtain 
a flyable antenna having a band width of only 
20 to 30 per cent. d Bearing these facts in mind 
we proceed to a discussion of several successful 
broad-band antenna designs. 

Wide-Band U-H-F Cone Antennas 

The flat impedance characteristics of biconi- 
cal antennas fed by balanced two-wire trans¬ 
mission lines were demonstrated experi¬ 
mentally and theoretically by Carter, by 
Schelkunoff, and by others, years ago. 3 More 
recently the Radio Research Laboratory has 
developed single unbalanced cone antennas for 
broad-band use on aircraft at ultra-high fre¬ 
quencies. 

The cone antenna consists of a sheet-metal 
circular cone tapering from a small diameter 
at its base, where it is attached to the inner 
conductor of the coaxial feed line or to the 
inner conductor of a coaxial taper leading into 
the feed line, outward to a diameter of the same 
order of magnitude as its height. The apex 
angle varies for different applications, a typical 
value being 60°. The top of the cone is capped, 
either with another section of a cone of greater 
apex angle or with a segment of a spherical 
surface. The impedance characteristics of the 


d Percentage band widths (F raa x/F miI1 — 1) X 100% 
where F max and F mi „ are the upper and lower limits to 
the frequency range over which the antenna is matched 
to some specified standard, usually to a better than 2:1 
»SWR on a 50-ohm line. 


cone antenna are remarkably flat over a range 
of frequencies corresponding to a range of an- 
tenna-height-to-wavelength ratios extending 
from about 0.2 to 2 or more. For many appli¬ 
cations the cone antennas are sufficiently broad 
band in themselves to permit their being fed 
directly from the feed line without need for 
conventional matching sections. Figure 13 
shows a sketch of a cone capable of covering the 
entire u-h-f range, 300 to 3,000 me, with less 
than 2.5:1 SWR on a 50-ohm feed line. 




Figure 13. Wide-band cone antenna covering 
range 300 to 3,000 me. (From RRL Report 411- 
TM-79.) 

Cone antennas may be supported by insulat¬ 
ing brackets attached to their peaks, or may be 
mounted within lucite radomes or blisters. Be¬ 
cause of the large cross-sectional dimensions 
of the cone, or its surrounding blister, these 
antennas are not suitable for use on aircraft at 
frequencies much lower than 300 me. 

The patterns of cone antennas are in general 
similar to those of simple cylindrical radiators 
of equal electrical length. If desired, the cone 
antennas may be mounted at an angle with the 
















240 


AIRBORNE ANTENNA DESIGN AT U-H-F AND Y-H-F 


side of the ship or in a horizontal position, in 
order to secure various amounts of horizontally 
polarized radiation. 

The Sleeve Antenna (for Upper V-H-F 
and Lower U-H-F) 

The sleeve antenna, developed by RCA 
Laboratories, is essentially a A/4 stub sur¬ 
rounded for about half its length by a coaxial 
sleeve which may be simply an extension to the 
feed line. The sleeve is grounded to the skin of 
the ship at its base, the antenna being fed at 
the mouth of the sleeve. Since this feed point 
is approximately half-way up from the base, in 
a low-current region, the apparent input re¬ 
sistance is high, of the order of 100 ohms. This 
constitutes a big advantage in broad-banding, 
since a high-impedance antenna can be matched 
down to a 50-ohm line over a much wider band 
than that over which a low-impedance antenna 
can be matched up, other things being equal. 
The sleeve antenna differs from simpler broad¬ 
band antennas in that the attainment of flat 
input impedance characteristics is not of pri¬ 
mary importance. The sleeve antenna involves 
four adjustable parameters—the ratios of sleeve 
length/total length and sleeve diameter/stub 
diameter in addition to the basic parameters 
(i.e., length/wavelength and diameter/length) 
of the simple stub—which make possible a high 
degree of control over the impedance character¬ 
istics of the antenna. By properly manipulat¬ 
ing these four variables one can attain char¬ 
acteristics which are not necessarily flat and 
which may vary rapidly with frequency but 
which vary in the right way to “track” with a 
preselected type of matching section. Because 
of this flexibility, the sleeve antenna can be 
used to take better advantage of the properties 
of a simple matching section than can be ob¬ 
tained with antennas affording less control over 
input impedance. The effectiveness of this ap¬ 
proach to broad-banding, i.e., that of distorting 
the antenna impedance to fit the characteristics 
of a given matching section, can be striking, a 
tenfold increase in useful band width resulting 
from the application of a properly designed 
matching transformer to a properly designed 
sleeve antenna being not unusual. A sketch of 


a simple sleeve antenna, and a typical SWR- 
frequency curve are shown in Figure 14. 

Since the current distribution on the outer 
surfaces of the sleeve antenna is very much the 
same as that existing on the surface of a simple 
stub, the field patterns of sleeve antennas on 
aircraft will be similar to those of whip or stub 
antennas in corresponding locations. 




Figure 14. Schematic of sleeve antenna and STFR 
for given dimensions. L — 25.3 cm, S — 12.7 cm, 

D = 2.5 cm, d — 0.95 cm; matching section 100 
ohms solid polyethelene dielectric, small type N 
connector. 

The sleeve antenna is most useful on aircraft 
in the general range 100 to 1,000 me. At higher 
frequencies a cone antenna may be used to bet¬ 
ter advantage, while at lower frequencies the 
cross section of wide-band sleeves becomes pro¬ 
hibitively large, so that broad-band whip an¬ 
tennas are more satisfactory. 

Broad-Band Whip Antennas 

Broad-band whip antennas developed by the 
Antenna Section Research Division, Aircraft 
Radio Laboratory, Wright Field, constitute 
some of the most successful aircraft antennas. 
In addition to their electrical features these an¬ 
tennas have very low wind drag, are tactically 
inconspicuous, are easily mass-produced, and 
are relatively easy to install on aircraft. 

The antennas are tough steel whips, the 
diameter tapering from about in. at the base 
















241 


ANTENNAS FOR VERTICAL POLARIZATION 


to about y 8 in. at the tip, the height (roughly 
A/4) ranging from 30 in. to 6 ft, depending on 
the frequency band. When used in conjunction 
with a simple two-element matching section 
consisting of lengths of standard coaxial cable 
compactly coiled in a metal can attached to the 
base of the whip, these antennas are capable of 
approximately 40 per cent band width in the 
lower v-h-f range. Several such antennas, each 
with its associated matching section, have been 
designed to be easily interchangeable in a single 
mounting fixture, four of them together cover¬ 
ing the 36- to 110-mc band with less than 2:1 
SWR on 50-ohm cable. 

The whip antennas depend for their broad¬ 
band characteristics partly upon the fact that 
the impedance level of a stub antenna worked 
against a cylindrical ground surface having a 
circumference of the order of magnitude of the 
operating a is higher than that of the same 
antenna worked against a flat ground plane. 
This fact is shown by the curves of Figure 15, 
which represent the variation with frequency 
of the resonant resistance of stub antennas 
identical except for length mounted on curved 
surfaces. Both curves indicate that impedance 
levels much higher than the normal 36 ohms of 
a stub antenna worked against flat ground can 
be obtained with stub or whip antennas 
mounted on the roughly cylindrical fuselages of 
large planes in the lower v-h-f range. 

Evidence indicates the necessity for basing 
the design of antennas for the lower v-h-f 
range upon impedance measurements made 
with the antenna installed in the location in 
which it is to be used. While such measure¬ 
ments are preferably made on the actual ship, 
or on a partial full-scale mock-up it is possible, 
if great care is taken in scaling both the plane 
and the antenna and its feed system, to obtain 
useful results with measurements made on 
models. While the effect of the fuselage upon 
the input characteristics of broad-band whip an¬ 
tennas is favorable, both as regards high-im¬ 
pedance level and flat reactance characteristics, 
if a whip antenna incorporating a matching 
section based upon flat-ground-plane impedance 
measurements were used in the same location 
as these whips, the results would not be satis¬ 
factory. The characteristics of the antenna are 
too greatly affected by currents in the curved 


fuselage for ground-plane measurements to be 
valid. This effect is not limited to the lower 
v-h-f range, but occurs in any aircraft antenna 
installation where the radius of curvature of 
the skin of the ship at the antenna location is 
small compared with the operating A. 


FUSELAGE HEIGHT/ WAVELENGTH 
•34 .42 .51 .60 



FUSELAGE HEIGHT/ WAVELENGTH 
• 3I .42 .52 .63 



Figure 15. Characteristics of A, similar whips 
atop fuselage of B-24, 15 ft aft of trailing edge of 
wing, full-scale ship in flight (data from ARL) ; 

B, stub antennas below fuselage of B-24 on center- 
line of wing and fuselage; % -scale model (data 
from RCA Laboratories). 

Figure 16 shows a typical £TFK-frequency 
curve for a low-frequency broad-band whip 
mounted on a B-24. The band width shown, 
obtained by means of a simple two-element 
transmission line matching section, is more 
than twice as great as that obtainable with the 






242 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 




42 44 46 

FREQUENCY IN MC 


Figure 16. Broad-band whip antenna for B-24. 
(Data from ARL Report 360.) 


same antenna mounted on a large flat ground 
plane and used in conjunction with a much 
more complicated matching section. 

Figure 17 shows the principal plane patterns 
of whip antennas mounted atop the fuselage 
of a B-24. The typical butterfly distribution 
evident in the vertical athwart-ship planes is in 
fair agreement with that calculated by Carter 
for stub antennas worked against cylindrical 
surfaces of corresponding relative size. The 
effect of currents in the skin of the fuselage can 
be seen by comparing the vertical athwart-ship 
plane patterns with those for the vertical fore- 
and-aft planes; the former show much more 
radiation in directions below the horizon than 


do the latter, as would be expected since the 
top of the fuselage represents a much closer 
approach to large flat ground plane along the 
line of flight than it does athwart ship. 


Broad-Band Fan Antennas 

A fan-shaped array of three or more wires, 
strung from a lead-in on the side or top of the 
fuselage to some supporting structure, such as 
a guy wire or part of the ship itself, forms a 
satisfactory broad-band antenna at the ex¬ 
tremely low-frequency end of the v-h-f band. 
These antennas, developed by the Antenna Sec¬ 
tion, Research Division, Aircraft Radio Labo¬ 
ratory, Wright Field, have numerous advan¬ 
tages. 

1. They have impedance characteristics such 
that they are easily matched to 50-ohm cable 
over frequency bands of the order of 25 to 35 
per cent wide, an unusual band width in view 
of the low frequencies involved. Impedance 
matching of fan antennas is usually effected 
by means of a two-element matching section 
consisting of lengths of commercially available 
coaxial cable compactly coiled in a metal con¬ 
tainer mounted just after the lead-in, inside 
the ship. 

2. Since the fans are made of ordinary air¬ 
craft-antenna wire they offer much less wind 
resistance than do conventional large-surface- 
area antennas of comparable band width at 
comparable frequencies. 

3. Because of the fineness of these wires a 


HORIZONTAL PLANE 


VERTICAL FORE-AND-AFT PLANE VERTICAL ATHWART-SHIP PLANE 



ASTERN 



DOWN 


UP 



Figure 17. Pattern of whip antenna installed on B-24, top centerline 55 in. behind trailing edge of wing, 40 me. 

























ANTENNAS FOR VERTICAL POLARIZATION 


243 




Figure 18. Broad-band fan antenna for B-24. 

(Data from ARL Report 357.) 

fan antenna is practically invisible at distances 
of the order of 20 ft or more, an obvious tacti¬ 
cal advantage. 

Among the disadvantages inherent in fan 
antennas, of negligible importance compared 
to their advantages for most applications, are 
the following: 

1. Fans must be tailored especially for each 
installation. Because of their spread in area, 
the characteristics of fans are sensitive to 


minor differences in the structure of the plane 
in their immediate vicinity and to the presence 
of near-by antennas. 

2. Installation of a fan is not a very con¬ 
venient procedure. 

3. The field patterns of fan antennas are 
not notably symmetrical, since to get sufficient 
height for a large percentage of vertically 
polarized radiation the antennas must usually 
be worked against one side or other of the 
ship. In some cases the asymmetry is such 
that a tactical course must be flown. 

4. There is usually a considerable percentage 
of horizontally polarized radiation in the field 
of a fan antenna. 

Figure 18 shows a sketch of a typical fan 
antenna installation and its /SIUR-frequency 
characteristic as measured in flight at Wright 
Field. Figures 19 and 20 show field patterns 
at the center of the band of this antenna, for 
vertical and for horizontal polarization, re¬ 
spectively, as measured by means of models 
by the Ohio State University Research Foun¬ 
dation. 


Broad-Band Inverted-L Antenna 
(Low V-H-F) 

The broad-band inverted-L antenna, devel¬ 
oped by RCA Laboratories, is an adaptation 
of the simple inverted-L, or flat-top, antenna 
used on aircraft at frequencies so low that the 


HORIZONTAL PLANE 


VERTICAL FORE-AND-AFT PLANE 


VERTICAL ATHWART-SHIP PLANE 



ASTERN DOWN OOWN 


Figure 19. Vertical polarization patterns for 3-wire fan 31-mc antenna for B-24, wires strung from lead-in 
near top starboard fuselage 47 in. forward of leading edge of starboard stabilizer to point near bottom side of 
fuselage 105 in. forward of leading edge of vertical stabilizer. (From ARL Report 357.) 






















244 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 



ASTERN 



DOWN 



DOWN 


Figure 20. Horizontal polarization field strength pattern of 3-wire fan 31-mc antenna for B-24; wires strung 
from lead-in near top starboard fuselage, 47 in. forward of leading edge of starboard stabilizer, to point near 
bottom side of fuselage, 105 in. forward of leading edge of vertical stabilizer. (From ARL Report 357.) 


height of a conventional A/4 whip or stub 
would be prohibitively great. A sketch of a 
simple inverted L having a height equal to half 
its total length is shown in the upper portion 
of Figure 21, where the measured input im- 




ANTENNA HE IGHT/WAVELENGTH 

Figure 21. Impedance characteristics of A, simple 
inverted L antenna, and B, a broad-band inverted 
L antenna. 


pedance of such an antenna is also shown. It 
is evident from these impedance characteristics 
that while the antenna has a much smaller 
physical height at resonance than the corre¬ 
sponding whip antenna, its impedance level is 
low and its reactance characteristic is steep, 
limiting its usefulness to spot-frequency or 
narrow-band applications. 

It has been found possible, by means of a 
sleeve (that is, by extending the coaxial feed 
line beyond the ground plane up to the bend 
in the antenna) and by properly proportioning 
the relative cross sections of the vertical and 
horizontal members of the antenna, to retain 
most of the reduction in vertical height gained 
in the simple inverted L and at the same time 
secure an antenna of broad-band characteris¬ 
tics. The modified inverted L is sketched in 
the lower part of Figure 21. The modified ver¬ 
sion has a much higher impedance level and 
an appreciably flatter reactance-frequency 
curve than the simpler antenna. Consequently 
it can be matched to a low-impedance line over 
much wider frequency bands. The actual band 
width obtainable with an inverted L depends 
upon the complexity of the matching section 
used; band width varies from about 12 per 
cent in the case of an L fed directly from 
50-ohm line to about 58 per cent when the L 
is used in conjunction with a three-element 
matching section consisting of lengths of com¬ 
mercially available coaxial cable. 

























ANTENNAS FOR VERTICAL POLARIZATION 


245 


18,3,2 Quarter-Wave Antennas for Vertical 
Polarization 

Among the more simple antennas suitable 
for use on aircraft in applications calling for 
limited band width in the u-h-f and upper 
v-h-f ranges are the following. 


ANTENNA ANTENNA BASE 




Figure 22. Thick stub antenna (AN-155). (Data 
from RRL Report 411-TM-92.) 


Thick Stub Antenna 

A a/4 stub of fairly large cross section is 
known to have broad-band input characteris¬ 
tics. However, there are two difficulties inher¬ 
ent in these antennas: (1) The fact that they 
must be base-insulated requires the use of low- 
loss solid-dielectric mounting fixtures of great 


mechanical strength, and (2) the large base 
must be connected to the small inner conductor 
of a coaxial feed line in a manner which will 
not destroy the intrinsic broad-band charac¬ 
teristics of the antenna. This last is a difficult 
problem which has not been satisfactorily 
solved to date. 

The AN-155 antenna, developed by the Radio 
Research Laboratory, is an example of the 
thick stub as used on aircraft. This antenna, 
sketched in Figure 22, consists of a phenolic- 
impregnated maple mast, covered, except at its 
base, by a metallic sheath. The base is held by 
an insulating bracket and the sheath is fed by 
a tapered metal strip, or “dog ear,” connecting 
the lower edge of the sheath to the inner con¬ 
ductor of a standard coaxial cable connector. 

The measured input impedance of this an¬ 
tenna is also shown in Figure 22. It is to be 
remarked that these characteristics depend to 
a rather large extent upon the shape and posi¬ 
tion of the “dog ear.” Such a mast antenna, 
30 in. high, and of 2%xl*4 in. streamline cross 
section, is capable of covering the 90-to-110-mc 
band without need for external matching sec¬ 
tions. By cutting down the length of the an¬ 
tenna, higher frequency bands, of increasingly 
greater width, may be covered. 


Brown-Epstein Antennas 

A simple u-h-f antenna, combining a strong 
mechanical mounting with provision for mod¬ 
erate band width, is shown in Figure 23A. In 
its simple form the antenna consists of a A/2 
rod mounted coaxially in a A/4 deep cylindrical 
well set into the ground plane against which 
the antenna is worked. The portion of the sys¬ 
tem contained in the well serves two purposes: 
(1) It acts as a shorted A/4 line, which, by 
presenting a high impedance to ground at the 
central feed point, effectively insulates the base 
of the protruding A/4 radiator while at the 
same time it provides a strong metallic mount¬ 
ing, and (2) this shorted a/4 line acts as a 
parallel-resonant circuit in shunt with the an¬ 
tenna, a circuit well known to be effective in 
flattening the reactance characteristic and in 
raising the impedance level of a series-resonant 
antenna. 



















246 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


A version more suitable for use on aircraft 
is sketched in Figure 23B. Here the shorted- 
line support has been placed inside the radi¬ 
ator, where it functions exactly as before, with 



vertical fin of a B-17, where surface currents 
in the immediate neighborhood of the feed 
point of a conventional stub would greatly 
modify the stub impedance and radiation 
characteristics. 

Half-Folded-Dipole Antenna 

One-half a Carter folded dipole worked 
against ground (see Figure 23D) forms an 
aircraft antenna with two advantages over a 
simple stub: (1) It includes its own matching 
section, for, by proportioning the relative di¬ 
ameters of the two conductors properly, it is 
possible to secure a perfect match to the feed 



QUARTER-WAVE 
HIGH IMPEDANCE 
CHOKE. 


FEED LINE 



STRONG 

METALLIC 

CONNECTION 


FEED LINE 


Figure 23. Antennas for vertical polarization: A, 
Brown-Epstein antenna; B, modified Brown-Ep- 
stein antenna; C, skin-back antenna; D, half-folded 
dipole. 



the additional advantage that the radiating 
surface is now larger and consequently will 
have flatter impedance characteristics. This 
surface may be streamlined to reduce drag. 

Skin-Back Antenna 

The skin-back antenna, sketched in Figure 
23C, consists of a A/4 radiator which may be 
considered as the continuation of the inner con¬ 
ductor of the coaxial feed line, the outer con¬ 
ductor of which is folded back upon itself to 
form the lower half of a vertical A/2 dipole. 
The shorted A/4 line acts as a high-impedance 
choke in series with the lower half of the an¬ 
tenna, effectively isolating it from the re¬ 
mainder of the outer surface of the feed line. 

This antenna has been used on aircraft in 
installations such that it is desirable to isolate 
the antenna from the adjacent surfaces of the 
ship. For example, it has been used atop the 



Figure 24. Impedance characteristics of: A, half- 
folded dipole, and B, bent half-folded dipole. 


line, and (2) since one side of the antenna is 
metallically grounded the antenna is inherently 
strong. Measured impedance characteristics of 
a typical half-folded dipole are shown in the 
upper part of Figure 24. 
































ANTENNAS FOR VERTICAL POLARIZATION 


247 


18.5.3 Less-than-Quarter-Wavelength 
Vertical Antennas 

An obvious means of minimizing aerodyna¬ 
mic and mechanical difficulties while at the 
same time retaining some of the desirable 
features of the A/4 antenna is to use resonant 
antennas which are short compared to the op¬ 
erating A. There have been many attempts at 
antenna design along this line, four of which 
are shown in Figure 25. 


STREAMLINED 


B 




HELICAL COIL 

DIELECTRIC 
CYLINDER 


COAX FEED 



•COAX FEED 
METALLIC CONNECTION 

Figure 25. Antennas less than X/4 long for verti¬ 
cal polarization: A, dielectric; B, helical; C, in¬ 
verted L; D, bent half-folded dipole. 

Dielectric Antenna 

A simple stub radiator, surrounded by di¬ 
electric material, will be resonant at a much 
lower frequency than the same antenna in air, 
the actual reduction in physical length depend¬ 
ing upon the inductive capacity and relative 
volume of the dielectric. The impedance char¬ 
acteristics of such an antenna are plotted in 
Figure 26. It will be noted that the impedance 
level is too low to permit successful broad 
banding. 

Helical Antenna 


Inverted-L Antenna 

The inverted L has short* physical height 
and, if modified, can be broad banded, as is 
described elsewhere in this report. 




Figure 26. Impedance characteristics of A, heli¬ 
cal antenna of Figure 25, and B, stub antenna in 
dielectric cylinder. 


Bent Half-Folded-Dipole Antenna 

This antenna has impedance characteristics 
which are matchable to 50 ohms over moderate 
frequency bands. 


The helical antenna contains its own loading 
coil, and while it can be made to be resonant at 
a height equal to a small fraction of A/4, its 
characteristics are marked by a very low-im¬ 
pedance level and by a very steep reactance 
characteristic so that its usefulness, if any, is 
limited to spot-frequency applications. 


is.5.4 Half-Wave Grounded-Loop Antenna 

A semi-circular A/2 loop antenna, mounted 
in a vertical plane, with one end grounded to 
the skin of the ship and the other end attached 
to the inner conductor of a coaxial feed line, 
has several advantages over the simple A/4 



























248 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


stub for vertical polarization in the u-h-f range. 

1. Its vertical height is less than one-sixth 
of the resonant A, so that if mounted with the 
plane of the loop in the line of flight this an¬ 
tenna will have less wind drag than the corre¬ 
sponding stub. 

2. Since one side of the loop is grounded it 
can be made to have great mechanical strength. 

3. Its impedance characteristics are such that 
it can be easily matched over quite wide fre¬ 
quency bands. 

The field pattern, for vertical polarization, 
of this antenna worked against a flat surface 
on aircraft will be similar to that of a vertical 
stub antenna in the same location. The patterns 
of full-wave loops have been investigated theo¬ 
retically by Carter, and are discussed in Sec¬ 
tion 18.7. 

186 MULTIPLE-RESONANT ANTENNAS 

It may sometimes be desirable in aircraft an¬ 
tenna work to use a single antenna for trans¬ 
mission or reception in two or more separate 
frequency bands, the frequencies being so 
widely spaced as to make the use of a broad¬ 
band antenna covering the entire range in¬ 
cluding these bands impractical. One scheme 
for realizing such an antenna system is shown 
in Figure 27. The antenna consists of a cylin¬ 
drical stub made in two pieces, the upper 
portion being supported by the inner conductor 
of a shorted coaxial line which is recessed into 
the lower part of the stub. The lower part of 
the antenna is of such length as to be resonant 
at the center of the high-frequency band, the 
A/4 line built into this section effectively isolat¬ 
ing the lower stub from the upper portion of 
the antenna. The total length of the antenna 
is made such that, allowing for the reactance 
introduced by the line section, the complete 
antenna will be resonant in the center of the 
low-frequency band. It will be seen from the 
impedance curves of Figure 27 that the partic¬ 
ular antenna shown is useful from 320 to 372 
me and from 468 to 520 me, representing an 
extreme range of frequencies which could not 
be covered by a conventional antenna of like 
cross section. By varying the dimensions of the 
various parts of the antenna it is possible to 


secure pass bands of greater or less spacing and 
of greater or less band width. The principle 
can, of course, be extended to three or more 
pass bands. This antenna has one advantage 




Figure 27. Stub antenna having two resonant 
bands, 333 and 480 me. 


over very wide-band antennas, in that it avoids 
the pattern difficulties which may appear at 
the high-frequency end of the band of such an¬ 
tennas, where the antenna may be several 
multiples of A/4 long. 


18 ^ ANTENNAS FOR HORIZONTAL 
POLARIZATION 

The design of aircraft antennas for hori¬ 
zontal polarization in the v-h-f and u-h-f ranges 
generally involves much greater difficulties 
than the design of corresponding vertical an¬ 
tennas. Not only are there grave mechanical 
problems, particularly in the lower v-h-f band 
where the structure of the plane offers few 
alternative methods of supporting the antenna 




















ANTENNAS FOR HORIZONTAL POLARIZATION 


249 


and consequently permits only a restricted 
choice of antenna locations, but there are elec¬ 
trical problems as well. The pattern and im¬ 
pedance requirements for horizontal antennas 
at these frequencies are usually such as to 
demand that the two components of the an¬ 
tenna be fed in some definite phase relation¬ 
ship, usually 180° out of phase, with the result 
that balance transformers as well as conven¬ 
tional matching sections must be included in 
the feed system if band width is desired. 

18,71 Broad-Band Balance Transformers 

Since most antenna installations for hori¬ 
zontal polarization on aircraft are of the bal¬ 
anced type, and since most aircraft transmit¬ 
ters are designed to work into low-impedance 



Figure 28. Broad-band balance transformer 
known as a “bazooka” or “balun.” At bottom, re¬ 
flection introduced by balun between matched coax 
and matched twinax of impedance Z 0 . (From RRL 
Report 411-TM-22.) 

unbalanced cable, it is usually necessary to 
insert between the antenna and the line a 
device for maintaining balance even though the 
frequency departs considerably from reso¬ 
nance. One of many such transformers is the 


“bazooka” or “balun” developed by the Radio 
Research Laboratory, sketched in Figure 28. 
When properly constructed this transformer 
maintains both sides of the two-wire line at 
equal and opposite potentials with respect to 
ground over large frequency ranges to either 
side of that for which the twinax line inside 
the balun is A/4 long. Furthermore, if the 
characteristic impedance of the twinax line in 
the balun is large compared to those of the 
coaxial and balanced cables between which it is 
inserted, the presence of the balun will cause 
little reflection over very wide frequency bands, 
as may be seen from the reflection-frequency 
curves of Figure 28. These curves apply to a 
rather artificial case, since the impedance of 
balanced cable or of balanced antennas is usual¬ 
ly higher than that of coaxial cable. In practice 
a transforming section must usually be in¬ 
serted on one side or other of the balun. 

18-7,2 Antennas for “Uniform” Horizontal 
Plane Pattern 

Bent-Sleeve Dipole Antenna 

The sleeve dipole antennas developed by the 
Radio Research Laboratory give pear-shaped 
horizontal plane patterns for horizontal polar¬ 
ization and have broad-band characteristics at 
frequencies less than 600 me. The antenna 
consists of a a/2 dipole bent into a V having 
an included angle of about 100°; each arm of 
the V is surrounded for about half its length 
by a coaxial sleeve; the arms tie in to the bal¬ 
anced side of a broad-band balance transformer 
which in turn is fed by 50-ohm coaxial cable. 
The antenna and its attached balun form a unit 
which plugs into a streamlined cylindrical 
mount which is permanently attached to the 
skin of the plane, the mount holding the plane 
of the V horizontal, in proper relation to the 
skin of the ship and to the direction of flight. 
A series of such antennas may be used inter¬ 
changeably in the same mount, in order to 
cover a very wide total frequency range. 

Figure 29 shows plan and elevation sketches 
of this antenna, and includes a sketch of the 
feed system. The £PFR-frequency curve of that 
figure gives some idea of the band width at- 




















250 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 



GROUND PLANE OF SKIN OF SHIP 

K\\\\x\\\\\ / 


BALUN 


J 


K 


SLEEVE 


V 


COAX FEED 


J 


ELEVATION 



FREQUENCY IN MC 


Figure 29. Plan and elevation views, feed system, 
and SWR of bent-sleeve dipole. (From RRL Re¬ 
port 411-TM-122.) 


tamable in the upper v-h-f and lower u-h-f 
ranges. While these antennas have very satis¬ 
factory characteristics in the 200- to 600-mc 
range, their impedance characteristics are 
marred at higher frequencies by the adverse 
effects of the feed-system discontinuities on 
band width, and at lower frequencies they be¬ 
come physically large, introducing mechanical 
and wind-drag difficulties. 

Figure 30 shows the pattern yielded by the 
bent-sleeve dipole mounted approximately a/4 


below the undersurface of the fuselage of a 
large plane. The large energy throw-down in 
the vertical plane could be remedied by moving 
the antenna farther out (closer to A/2) from 
the skin of the ship. 

The Coaxial-Fed V-Dipole Antenna 

In some respects this antenna is similar to 
the bent-sleeve dipole antenna. It consists of a 
coaxial-fed (unbalanced) dipole with a/4 arms 
attached to the inner and the outer conductors 
of the supporting feed line, the arms lying in a 
plane perpendicular to the feed line and 
forming a V with an included angle of 95 to 
100°. The supporting line extends A/4 beyond 
the ground plane on which the antenna is 
mounted, its outer conductor being grounded 
at the base. 

The horizontal plane pattern of the antenna 
is peanut-shaped with side minima in field 
strength about 4 db down from the maxima. 
The vertical plane patterns show the large 
amount of throw-down to be expected from a 
horizontal antenna mounted A/4 from a con¬ 
ducting sheet. 

The antenna yields little vertically polarized 
radiation at resonance, but as the length of the 
supporting line departs from A/4, the currents 
in its outer surface result in increasingly larger 
percentages of vertical radiation. It has been 
found experimentally that the antenna can be 
used over frequency bands approximately 35 
per cent wide before the maximum average per- 



—- 440 MC 




400 MC 


380 MC 


Figure 30. Patterns of bent-sleeve dipole for horizontal polarization. (From RRL Report 411-TM-122.) 































ANTENNAS FOR HORIZONTAL POLARIZATION 


251 




Figure 31. Coaxial-fed V dipole for horizontal 
polarization. 


centage of vertical polarization becomes greater 
than 20 per cent of the total radiation in the 
plane containing the most vertically polarized 


energy. Since no effort is made to maintain 
balance between the two sides of the V, the 
patterns tend to become asymmetrical at fre¬ 
quencies far from resonance, another factor 
limiting useful band width to about 35 per cent. 

Figure 31 shows a sketch of a simplified 
version of the antenna, a typical set of input 
impedance characteristics, and a £IFR-fre- 
quency curve for an antenna having a built-in 
low-impedance series transformer. Because of 
the simplicity of the feed system these antennas 
have wide-band u-h-f impedance characteristics, 
a set of four interchangeable antennas covering 
the 500- to 1500-mc range with less than 2:1 
SWR on 50-ohm line. 

Figure 32 shows the measured horizontal 
plane patterns at three frequencies distributed 
over the range of a model intended for use in 
the 1175- to 1500-mc band. 

Split-Can Antenna 

Figure 33 shows a sketch of a u-h-f split-can 
antenna developed by the Radio Research Labo¬ 
ratory for horizontal polarization. The antenna 
consists of a cylinder of streamlined cross sec¬ 
tion, split longitudinally along the trailing side, 
and mounted normal to a ground surface from 
which its base is insulated. The antenna is fed 
by a balanced line, which ties on at the two 
opposing edges of the split. In a tentative 
theory the edges of the split are regarded as a 
continuation of the two-wire feed line, the 
surface of the antenna acting as a shunt loop 


horizontal plane 

AT 1185 MC 


HORIZONTAL PLANE 
AT 1280 MC 


HORIZONTAL PLANE 
AT 1510 MC 



Figure 32. Measured field patterns for coaxial-fed V dipole, horizontal polarization. 





























252 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


across this line. Since the currents in this sur¬ 
face loop are horizontal, the resulting radiation 
is horizontally polarized, the horizontal plane 
pattern being substantially uniform with the 
minimum only about 3 db down from the maxi¬ 
mum. When the surface of the antenna is less 
than a/2 around, its effect is that of an induc- 




Figure 33. Split-can antenna, horizontal polariza¬ 
tion. (From RRL Report 411-TM-34.) 

tive shunt across the feed line, resulting in an 
increase in the physical length of the antenna 
at resonance. For example, the 30-cm high an¬ 
tenna of Figure 33 is resonant at approximately 
375 me, corresponding to an electrical length of 
0.375 A compared to 0.24 or less for a conven¬ 
tional resonant stub of similar cross section. 

Loop Antennas 

The possibilities of circular loop antennas of 
dimensions large compared with the operating 
A have been explored, independently, by 
Carter, 4 Foster, 5 and Sherman. 6 Large loop an¬ 
tennas have characteristics quite different from 


those of the small loops used for direction find¬ 
ing at low frequencies, characteristics which 
make these antennas attractive in certain ap¬ 
plications where horizontal polarization is 
desired in the u-h-f range. 

In the absence of experimental data on the 
impedance and pattern characteristics of full 
circular loops on aircraft this discussion will be 
limited to the presentation of theoretical data 
taken from a report by Carter. 4 While the im¬ 
pedance level of loops less than A/2 in circum¬ 
ference is low, loops of the order of A in cir¬ 
cumference have respectable input resistances. 
A loop a/2 in circumference mounted with its 
plane horizontal would yield a very uniform 
pattern for horizontal polarization in the hori¬ 
zontal plane; unfortunately its radiation resist¬ 
ance would be only about 13 ohms, a value 
which would have to be stepped up consider¬ 
ably, possibly by means of a sleeve, before the 
antenna would be useful for any but narrow 
band applications. 

18 7 3 Antennas of theX/2 Dipole Type 

Antennas of the center-fed A/2 dipole type 
are commonly used for horizontal polarization 
in applications such that the nulls in the field 
pattern along the direction of the antenna 
axis are not objectionable. 

Brown’s Antenna 

An interesting u-h-f antenna for horizontal 
polarization on aircraft is that developed by 
G. H. Brown. The A/4 arms of the dipole 
consist of strong tubing held in line by an axial 
insulating rod, the dipole being supported in a 
horizontal position A/4 out from the skin of the 
ship by means of two vertical metal cylinders, 
closely spaced and connected to the two arms 
of the dipole at either side of the central feed 
point. Since the vertical supports are base- 
grounded, a mechanically strong mounting is 
secured, while the fact that these supports are 
A/4 long insures that the antenna is electrically 
insulated from ground. A coaxial feed line, 
which may include a series transformer match¬ 
ing section, runs up through one of the sup- 
















ANTENNAS FOR HORIZONTAL POLARIZATION 


253 


porting cylinders, tying on to the two sides of 
the dipole at the feed point. 

In another version of this antenna the axial 
rod aligning the two halves of the dipole is 
metallic and connected to the radiators only 
through metal plugs at each end of the dipole. 
This system is fed from a balanced line, one 
side of the line tying on to each radiator at the 
feed point, the coaxial lines included inside the 
radiators acting as a shunt matching section, 
which results in flat input impedance charac¬ 
teristics over a fairly wide frequency band. 

Wire Dipoles 

Because of mechanical and wind drag con¬ 
siderations none of the antennas previously con¬ 
sidered are suitable for use on aircraft at fre¬ 
quencies much below 200 me. For frequencies 
in the lower v-h-f range it becomes necessary 
to use wire antennas, an example being a A/2 
dipole of ordinary aircraft antenna wire strung 
parallel to the line of flight, supported at its 
two ends either by masts or by convenient 
points of the ship’s structure, and center-fed 
from a twisted pair or twinax cable. Such an¬ 
tennas are, of course, extremely narrow band, 
and are therefore useful only for spot-frequen¬ 
cy applications, or for applications in which 
manual or mechanical antenna tuning is per¬ 
missible. 

The band width of wire dipoles can be con¬ 
siderably improved by making each radiator 
in the form of a cylindrical or conical cage of 
wires, thus simulating large-surface conductors 
while at the same time retaining the low-drag 
features of wire antennas. The average char¬ 
acteristic impedance of multi-wire cage dipoles 
is discussed at length in a report prepared by 
Division 15. 7 


Polyphase Antennas 

Another example of antenna system which 
has u-h-f possibilities, and perhaps even for 
much lower frequencies, is the turnstile an¬ 
tenna developed by Brown and by Lindenblad 
for f-m and television transmitting purposes. 
This antenna, which may be regarded either as 
two crossed a/2 dipoles fed in phase-quadrature 


or as four A/4 antennas arranged along the 
diagonals of a square and fed in 90° phase 
rotation, yields an unusually symmetrical pat¬ 
tern for horizontal polarization in the hori¬ 
zontal plane, and has the further advantage of 
naturally broad-band characteristics in that the 
reflection coefficient on the main feed line is 
equal to the square of that existing on the 
branch lines leading to the individual antennas. 
The desirable features of the turnstile antenna 
are accentuated in the three-phase Y antenna, 
which, since it has only three radiators, instead 
of four, is perhaps more attractive for low-fre¬ 
quency use on aircraft. This antenna consists 
of three A/4 radiators arranged symmetrically 
in the horizontal plane, the radiators being fed 
with equal currents in three-phase relationship. 
In this system the reflection coefficient on the 
main feeder is equal to the cube of that existing 
on the individual branch lines, resulting in still 
greater broad banding due to feed than is ob¬ 
tained with the turnstile antenna. 

Despite their advantages there are very good 
reasons why polyphase antenna systems have 
not been exploited thus far. In the first place 
these systems require high-impedance com¬ 
ponent antennas, of 100 ohms input impedance 
in the case of the turnstile, and 150 ohms in 
the case of the Y, presuming a 50-ohm main 
line. While high-input-impedance-A/4 antennas 
may be obtained by means of sleeves there re¬ 
mains the problem of obtaining high-impedance 
branch lines. Furthermore the patterns of these 
antennas depend upon proper phasing of the 
currents in the component antennas, which in 
turn depends upon a proper impedance match 
throughout the system. For this reason it is 
difficult to measure the field patterns of these 
antennas on aircraft by means of models, since 
not only must the antenna dimensions be prop¬ 
erly scaled, but the individual antennas must 
be accurately matched to their feed lines. The 
latter condition is difficult to realize at the s-h-f 
range used in model work. Whether the large 
amount of experimental work required in the 
development of these antennas is justified 
depends upon the need for uniform pattern and 
broad-band impedance characteristics. The fact 
remains that these are among the very few an¬ 
tennas that have even a chance of satisfying 



254 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


such requirements on aircraft in the lower 
v-h-f range. 

18.7.5 ]\J u ltiple Antennas for Horizontal 
Polarization 

Any of the antennas suitable for vertical 
polarization may be used for horizontal polari¬ 
zation if properly mounted. However, such an¬ 
tennas, mounted on one side of the fuselage 
in a horizontal position or at an angle with the 
side of the fuselage, necessarily give asymmet¬ 
rical patterns. If symmetry is desired it is 
necessary to use two similar antennas mounted 
in corresponding positions on opposite sides of 
the plane and fed in proper phase relationship. 
Such antenna combinations have been devel¬ 
oped by the Radio Research Laboratory in col¬ 
laboration with the Ohio State University Re¬ 
search Foundation. When antennas made up 
of a/4 stubs and cones are fed in phase, the 
horizontal plane pattern contains lobes ahead 
and astern for vertical polarization, with nulls 
in the corresponding positions for horizontal 
polarization; when the antennas are fed 180° 
out of phase the situation is reversed. As far 
as symmetry is concerned, these antenna sys¬ 
tems become less satisfactory the higher the 
frequency. 

Since no compensation is gained in in-phase 
or in out-of-phase feeding, individually broad¬ 
band antennas must be used if band width is 
desired. 

j. 8.7. 6 Surface or Interior Antennas 

Flush-mounted antennas such as slots, horns, 
and wedges are suitable for horizontal polari¬ 
zation at upper v-h-f and u-h-f frequencies. In 
many cases, such antennas, mounted singly on 
the underside or wing or fuselage or in pairs 
on either side of the ship, have pattern and 
impedance characteristics satisfactory for cer¬ 
tain applications. These antennas are described 
elsewhere in this report. 

188 ANTENNAS FOR BOTH VERTICAL 
AND HORIZONTAL POLARIZATION 

While any of the conventional linear anten¬ 
nas can be mounted at odd angles with the skin 


of the ship in order to secure varying amounts 
of both horizontal and vertical polarization, 
and while other antennas, such as fans and 
loops, incidentally give radiation of both types, 
there are special antennas for this purpose. 

The Fish-Hook Antenna for Circular 
Polarization 

The M2201 and M2202 antennas developed 
by the Radio Research Laboratory consist of 
two thick dipoles, crossed at right angles, with 
the individual radiating elements bent down¬ 
ward at an angle of approximately 30° with 
the horizontal. Each radiator is supported by 
one conductor of a four-conductor open line A/4 
long. This line leads into the interior of the 
ship to a phasing and matching unit which 
feeds the two dipoles in phase quadrature and 
which matches the combined input impedance 
to the main transmission line. A sketch of the 
antenna, together with its SWR-frequency 
characteristic and measured field patterns, is 
shown in Figure 34. 




500 540 580 620 660 700 740 780 

FREQUENCY IN MC 



Figure 34. Fish-hook antenna for circular polar¬ 
ization. (Data from RRL Report 411-TM-53.) 

The antenna is intended to be mounted on 
the underside of the fuselage of a plane, the 
maximum radiation being downward with ap- 






































ANTENNAS FOR BOTH VERTICAL AND HORIZONTAL POLARIZATION 


255 


proximately equal vertically and horizontally 
polarized components. The antenna feed sys¬ 
tem is so arranged that it can be used with 
either one or two transmitters. It is designed 
to be mounted inside a plastic radome. 


Trailing-Wire Antennas (H-F and 
Low V-H-F) 

The simple trailing-wire antenna used on 
aircraft for long-range communication in the 
m-f and h-f ranges is an example of an antenna 
with which greater or less amounts of either 
vertical or horizontal polarization may be ob¬ 
tained. It consists simply of a length of copper- 
clad steel antenna wire wound on a reel and 
passed out through a fair-lead installed in the 
bottom or side of the fuselage. The wire ter¬ 
minates in a wind sock, for horizontal polariza¬ 
tion, or in a streamlined weight if vertical po¬ 
larization is desired. The antenna is fed from a 
coaxial cable through a contact located where 
the wire enters or leaves the fair-lead, and may 
be considered as working against the skin of 
the ship as ground. 

When trailing wires are operated at a fixed 
length less than A/4 their impedance character¬ 
istics are marked by low resistance and by 
large capacitative reactance, varying rapidly 
with frequency. If transmitting efficiency is 
desired, the antenna must be fed through a 
matching section or tuning unit containing 
manually adjusted or motor-controlled variable 
lumped elements. When trailing wires are op¬ 
erated at some resonant length, such as A/4 
or 3a/4, they are, of course, nonreactive and 
have a reasonably large input resistance which 
varies with frequency in a complicated manner 
depending upon the size of the plane, the length 
of the wire, and the relative positions of plane 
and wire. Under many conditions this resis¬ 
tance is close enough to 50 ohms so that the an¬ 
tenna may be fed directly from the transmis¬ 
sion line without recourse to matching sections. 
If extreme transmitting efficiency is required 
the antenna resistance may be matched to that 
of the feed line, by means of simple circuits 
of coils and capacitors, as is shown by the ex¬ 
ample of Figure 35. 


At the low frequencies at which trailing- 
wire antennas are ordinarily used, matching 
sections consisting of lengths of transmission 
line are too bulky to be practical. 

The patterns of simple trailing antennas in 
the lower v-h-f range are generally messy as 
compared with those of the fixed aircraft an¬ 
tennas recently developed for low-frequency 
use. 



6.0 7.0 8.0 9.0 10.0 11.0 

FREQUENCY IN MC 



L g = 2.2 JJ.h 
C 2 = 29.5 JU-JU. f 

Figure 35. Characteristics of resonant trailing- 
wire antenna operated at 3X/4 resonance. (Data 
from ARL.) 

Stingeree Antenna 

The stingeree antenna, developed by the Bell 
Telephone Laboratories, is intended for broad¬ 
band use, for either vertical or horizontal po¬ 
larization, in the lower v-h-f range. 

The antenna consists of a A/2 dipole of the 
skin-back type, trailed from the plane at the 
end of 50 to 100 ft of standard coaxial cable. 
The antenna, sketched in Figure 36, contains a 
two-element transmission line matching sec¬ 
tion which is built into one side of the dipole. 
The radiating surfaces of the dipole consist of 
cylindrical metal-braid sheathing, quite similar 
to the armor used on RG-35/U coaxial cable. 
The far end of the antenna terminates in a 
streamlined weight. The combination feed and 
tow cable is coiled just before entering the 
dipole proper, the coil acting as a high-im¬ 
pedance choke in series with the antenna and 

























256 


AIRBORNE ANTENNA DESIGN AT U-H-F AND Y-H-F 


therefore tending to keep radiating currents 
from the outer surface of the feed line. 

This antenna is said to have band widths of 
the order of 25 to 35 per cent at the extreme 
low end of the v-h-f band, the SWR at the in¬ 
put of the feed line, some 50 to 100 ft from the 
antenna, being less than 2:1 over such ranges. 



Figure 36. Stingeree antenna of Bell Telephone 
Laboratories. 


The pattern of the stingeree is said to closely 
resemble that of a A/2 dipole in free space. The 
antenna has the further advantage in that it 
is towed A or more behind the ship and there¬ 
fore its radiational characteristics may be ex¬ 
pected to be much less dependent upon the 
nature and size of the plane than are those of 
ordinary fixed aircraft antennas. 

189 SURFACE ANTENNAS 


conspicuity, are solved at once, simply by 
elimination. These spectacular advantages have 
aroused great interest in surface antennas, an 
interest which has extended to the development 
of planes especially designed to accommodate 
such antennas, an example being the Bell D-6, 
a plywood plane upon whose nonconducting 
surfaces antennas were simply to be painted. 
Relatively little, however, had been accom¬ 
plished in the field at the time the present re¬ 
port was prepared. In a rather complete file of 
the reports issued by the various laboratories 
engaged in aircraft antenna research there was 
not a single one dealing with surface antennas 
at frequencies lower than 3,000 me. 

The following material constitutes what little 
was learned about surface antennas at this 
laboratory (RCAL). It represents work done 
here largely at the request of the Radio Test 
Department, U. S. Naval Air Station, Patuxent 
River, Maryland. 




MODIFIED VERSION OF THE ABOVE 



END VIEW 


ys sy sssrrry / ; s /; ^ 


¥ 


Figure 37. Simple-slot antennas and modified 
version where X/4 cavity has been bent back paral¬ 
lel to ground plane. 


By mounting an aircraft antenna inside the 
plane, with its radiating surfaces flush with the 
skin of the ship, many of the problems of an¬ 
tenna design, including wind-drag, mechanical 
strength, icing, precipitation static, and tactical 


18,91 The Single-Slot Antenna 

Figure 37 shows a sketch of a simple slot, 
approximately 7A/8 long by A/30 wide, cut out 
of the skin of the ship. The slot is backed by a 












































SURFACE ANTENNAS 


257 


rectangular resonating cavity, of the same 
cross section, and A/4 deep. The system is fed 
by a short cylindrical radiator, running across 
the slot, and introduced along the center line of 
the wide side of the cavity as an extension to 
the inner conductor of the coaxial feed cable. 

Figure 38 shows that the system behaves as 
an antiresonant circuit of fairly high Q. 



750 850 950 1050 1150 



r = 0.87 CM 


-*-W 



Figure 38. Impedance characteristics of simple- 
slot antenna. 


This figure also shows the effect of the posi¬ 
tion of the feed, relative to the bottom of the 
cavity, upon the impedance level of the system. 
Since the input resistance at antiresonance de¬ 
creases from high values to zero as the feed^ 
radiator approaches the bottom of the cavity, 
it is evident that the simple slot can be matched 


to the feed line, at one frequency, simply by 
adjusting the position of the feed. 

Because of the steep characteristics of the 
antenna input impedance it is possible to ob¬ 
tain only four percent band width by means 
of a conventional a/4 transformer, a band 
width which may be approximately doubled if 
a two-element transmission line matching sec¬ 
tion is used. In view of the size of the antenna 



Figure 39. Double-slot antenna; two slots fed in 
phase. 


in wavelengths, it is evident that a simple-slot 
antenna has impedance characteristics very 
much less suitable to broad-banding than those 
of conventional cylindrical antennas. 6 

e It may be objected that the above data on band 
width apply to an extremely unfavorable case, in that 
the impedance level of this slot is much higher than 
that of the line to which it is to be matched. It might 
appear that band width could be increased by changing 
the position of the feed point so that the resonant 
resistance more nearly approaches the characteristic 
impedance of the line. This is not the case; while the 
resistance match can be improved by lowering the feed 
point, the reactance is not affected in the same propor¬ 
tion; the result is that low-impedance antennas have 
higher Q’s and less band width. 








































































258 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


The radiation from slot antennas is confined 
to the same side of the ship as that upon which 
the antenna is located. 

18 9 2 Double-Slot Antenna 

Figure 39 shows a sketch of a system of two 
parallel slots, each 3a/4 long, spaced A/4 apart, 
fed in phase. Although no impedance measure¬ 
ments have been made for this antenna, it is 
not likely that the band width of this system 
will be greater than that of a single slot, since 
there is no compensatory effect in the in-phase 
feeding of identical antennas, and since the 
effect of the mutual impedance at such small 
spacing is likely to be adverse. 




HIGH IMPEDANCE TRAPS 
ISOLATING RADIATING SURFACE 
FROM REST OF GROUND-PLANE 


Figure 40. Designs of double-slot antennas by 
Lindenblad. 

The field patterns of a double slot mounted 
under the wing of a PBY-5A are more sym¬ 
metrical than those of the single slot, and the 
downward beam in the vertical plane trans¬ 
verse to the slots is sharper. 


189 3 Lindenblad’s Double-Slot Antenna 

Figure 40 shows alternative arrangements of 
two slots spaced a/2 apart, each slot being fed 
through a A/4-deep resonant cavity which is 
folded back parallel to the skin of the ship. In 
this system there is evidence that surface cur¬ 
rents in the A/2-wide strip are responsible for 
most of the radiation, the strip behaving much 
like an array of thin a/2 dipoles lined up side 
by side. The A/4 feed cavities serve to isolate 
the radiating surface from the rest of the sur¬ 
face of the ground plane, performing the 
double function of placing a high impedance in 
series with the immediately adjacent outer sur¬ 
faces and of insuring that what current does 
exist in these surfaces will be in phase with 
that in the strip. 

SWR measurements indicate that band 
widths of the order of 10 to 15 per cent may be 
obtained without recourse to matching sections. 

18-9,4 Lindenblad’s Broad-Band Slot 
Antenna 

A very interesting slot system, which in¬ 
cludes a novel broad-band feed, is sketched in 
Figure 41. 8 From the outside of the ship the 
antenna appears as two thin slots, approxima¬ 
tely 0.65a long, spaced 0.15a apart. From the 
interior of the ship the antenna appears as a 
thin square box, approximately 0.55a on a side 
and 0.07A thick, so oriented that the two outer 
slots lie parallel to one diagonal. This box is 
divided into two layers of approximately equal 
thickness by means of an inner sheet of metal, 
which contains an inner slot 0.06a wide lying 
under the strip separating the two outer slots. 
A septum attached to this strip passes down 
through the inner slot to the bottom of the box. 
A feed strip, shaped as an equilateral triangle, 
leads from one edge of the inner slot to the 
bottom corner of the lower layer of the box, 
where it ties on to the inner conductor of a 
standard coaxial cable connector. By sys¬ 
tematically varying the width, length, and 
spacing of the outer slots, the spacing of the 
inner slot, and the shape of the feed triangle, 
it has been possible to attain band widths of 
20 per cent without need for external matching 













































SURFACE ANTENNAS 


259 


sections. Much wider band widths are possible 
if the standard of matching were to be slightly 
relaxed, say to 2.5:1 SWR on a 50-ohm line. 

The SIFR-frequency characteristic of a 
broad-band slot system designed for altimeter 
use is included in Figure 41. 



Figure 41. Plan and elevation views of broad¬ 
band slot antenna of Lindenblad. 


18 ' 9 ’ 5 Louvre, or Wedge, Antenna 

The louvre antenna developed by P. S. Carter 
for an application quite remote from communi¬ 
cations is an interesting example of a flush- 
mounted antenna. The system, Figure 42, con¬ 
sists of three very thin wedges arranged to 
overlap so that their open bases are spaced 
approximately A/4 apart. The system is in¬ 
tended to be mounted upon the side or under¬ 
surface of the plane, depending upon the polar¬ 
ization and pattern desired, the open ends of 
the wedges appearing as long thin slots covered 
with low-loss dielectric, the antenna being fed 


directly from a coaxial line entering the middle 
wedge. The impedance characteristics of the 
antenna are such as to make it very sharply 
resonant. The field pattern, except in the plane 
normal to both the surface on which it is 
mounted and the length of the louvre openings, 
consists of fairly sharp single lobes, the posi¬ 
tions of which in space may be adjusted simply 
by manipulating the tuning condensers in the 
two outer wedges. 




Figure 42. Wedge, or louvre, antenna useful in 
drift indicating, tail warning, and other applica¬ 
tions. 

While the louvre antenna has few features 
attractive for communication purposes, it does 
have possibilities for other uses such as drift 
indicating, tail warning, and applications 
where easily managed lobe switching is desir¬ 
able. 

1896 The Waveguide Antenna 

The waveguide antenna sketched in Figure 
43 is a special type of horn antenna, i.e., a horn 
of zero flare. It is excited in the H 01 mode by 






























260 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 



Figure 43. Waveguide antenna, special form of 
horn of zero flare. 

means of a stub antenna mounted parallel to 
the short side of the guide, located approxi¬ 
mately a/ 4 from the closed end and fed directly 
from the coaxial feed line entering at the cen¬ 
ter of the long side. 9 

The open, or radiating, end of the guide can 
be covered with a sheet of low-loss dielectric 


material mounted flush with the skin of the 
ship. The patterns of a waveguide antenna hav¬ 
ing the dimensions shown in Figure 43 and 
mounted in the tail of an F6F are shown in 
Figure 44. This antenna was intended for hori¬ 
zontal polarization at 400 me, the guide being 
oriented so that its long side is vertical. The 
patterns are quite similar to those predicted by 
the theory and experiment of Barrow and 
Greene. 


is.9.7 Antennas in Semicylindrical Cavities 

Figure 45 is a conventional cylindrical anten¬ 
na mounted axially in a semicylindrical recess 
in the skin of the ship. The recess or cavity 
has an aperture approximately 0.4 a square 
which can be covered with a dielectric sheet 
mounted flush with the surface of the plane. 
The STFR-frequency characteristics show that 
while no band width is attainable with a simple 
stub radiator in the cavity (the resistance is 
too low and the reactance variation too steep 
in this case), the use of sleeve antennas results 
in quite appreciable band widths. 


horizontal plane vertical fore-and-aft plane 




q 

< 

Ll) 

X 

< 


Figure 44. Antenna power pattern of waveguide antenna having dimensions of Figure 43, operating on 400 
me, located in F6F tail with mouth of guide facing directly aft. 

























POWER CAPACITY OF AIRCRAFT ANTENNAS 


261 


1898 Conclusions 

The material presented in the preceding sec¬ 
tions summarizes what is known about surface 
antennas at this laboratory at the present time. 



Figure 45. Sleeve antenna in semicylindrical 
cavity. 


From this data we draw the following conclu¬ 
sions : 

1. Surface antennas, whether they be slots, 
horns, or cavities, are much larger relative to 
the operating A than are conventional exterior 
antennas. The maximum dimension is usually 
of the order of a A/2 or more. While slots or 
horns of such aperture are feasible at frequen¬ 
cies down to 100 me (assuming their use on 
large aircraft), a 30-mc slot antenna would 
require quite a little mechanical engineering. 
Surface antennas also have more or less bulk 
inside the skin of the ship, a fact which means 
that installation of even a small h-f antenna 
will be something of a major operation. 

2. Surface antennas have much less intrinsic 


band width than ordinary antennas. For most 
communication purposes this is not much of an 
objection, particularly in the light of recent 
developments resulting in band widths of the 
order of 20 per cent or more. 

3. Surface antennas have field patterns 
characterized by more directivity than is usu¬ 
ally desirable in communication work. They do 
not transmit or receive energy in directions op¬ 
posite to that in which they face, a situation 
which can probably be remedied by mounting 
two antennas on opposite sides of the ship. 

4. Surface antennas, while having reached 
a stage of development permitting their im¬ 
mediate application to many aircraft antenna 
problems, constitute a rich and virgin field of 
research, particularly along the lines of in¬ 
creasing band width (by continued develop¬ 
ment of broad-band methods of feeding them), 
reducing bulk (possibly by means of filling 
them with low-loss dielectrics of high induc¬ 
tive capacity), and improving patterns by 
means of multiple-antenna systems. 

1810 POWER CAPACITY OF AIRCRAFT 
ANTENNAS 

The maximum power that can be handled 
by aircraft antennas depends upon the nature 
of the antenna and upon atmospheric con¬ 
ditions. 

Power capacity varies approximately as the 
square of the conductor diameter, and conse¬ 
quently will be greater for thick cylindrical 
and conical antennas than for antennas con¬ 
sisting of one or more small wires, such as 
fixed- or trailing-wire antennas or fans. 

Since breakdown due to corona or arc-over 
depends upon field strength rather than volt¬ 
age, maximum power will depend upon the 
orientation of the antenna with respect to the 
ground plane against which it is worked, being 
greater for simple vertical antennas than for 
antennas having components parallel to the 
skin of the ship. Furthermore, since antenna 
voltage for a given power input is a function 
of the current distribution along the antenna, 
it is evident that an antenna with top-loading 
will have a different power limit from that of a 
simple A/4 stub. 



















262 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


18.10.1 Antenna Length and Resistance 

The antenna voltage for a given power input 
is proportional to the square root of the input 
resistance, implying that the maximum power 
for a given corona voltage will be proportional 
to the radiation resistance of the antenna. 
Hence a A/4 or longer antenna will handle more 
power than a short antenna. Since an electri¬ 
cally short antenna requires inductive loading 
to be fed at all, and since the ohmic resistance 
of the coil may be of the same order of magni¬ 
tude as the radiation resistance of the antenna, 
an appreciable fraction of the input power will 
not reach a short antenna at all. The loading 
coil must be designed to dissipate that frac¬ 
tion safely. 


18102 Antenna Surface 

The surface of a high-power antenna should 
be smooth and of relatively larger radius of 
curvature, since corona sets in at lower volt¬ 
ages the rougher the surface. 

The effect of dirt on the antenna surface is 
to start local discharges and may cause the on¬ 
set of general corona at lower voltages. 


Atmospheric Conditions 


static will set in, regardless of the voltage on 
the antenna. 


1810 - 4 Summary 

The problem of power limits of aircraft an¬ 
tennas is too complex to permit solution by 
simple, unqualified rules or formulas. It is, 
however, possible to make simplifying assump¬ 
tions which may be useful in a qualitative way 
in showing the effect of a few of the factors 
entering into the problem. The curves of Fig¬ 
ure 46 represent such approximations. While 
in particular cases they may not even be cor- 



SMALLEST DIAMETER IN INCHES 


The breakdown voltage of air is a compli¬ 
cated function of its density, and so depends 
upon pressure and temperature. The dielectric 
strength of air increases with density, density 
decreases with decreasing pressure and with 
increasing temperature. The power capacity of 
aircraft antennas is therefore less at high alti¬ 
tudes than at low, the effect of decreasing pres¬ 
sure much more than compensating the effect 
of decreasing temperature as the altitude in¬ 
creases. 

While moisture present in the air has little 
effect upon the starting point of corona, once 
corona is started rain and humidity reduce the 
spark-over voltage greatly. 

Ionization pre-existing in the air surround¬ 
ing the antenna has little effect on the onset of 
corona. However if the plane picks up sufficient 
charge, corona in the form of precipitation 


Figure 46. Power-handling limit of A/4 antenna 
at elevation of 40,000 ft. 

rect as to order of magnitude, they show, in a 
general way, the relation between power limit 
and antenna diameter for two types of anten¬ 
nas important in aircraft radio. 


1811 PRECIPITATION STATIC 

Precipitation static interferes with aircraft 
communication when the receiving plane passes 
through rain, snow, or through clouds of dust 
or ice particles. When first observed the static 
appears as a series of popping noises in the 
receiver, which noises finally develop, as flight 
continues, into a continuous roar completely 
obscuring the signal. The effect seems to be 





















AIRCRAFT ANTENNAS AND AIR DRAG 


263 


worse with higher speed planes, and in a given 
case static can usually be reduced by reducing 
speed. 

1811,1 Remedies 

The elimination of precipitation static is 
achieved, by a twofold attack on the problem. 

1. Sharp points on the surface of the plane 
are removed. As far as antenna design is con¬ 
cerned, this demands that the use of fine wire 
or of fittings involving surfaces of small radius 
of curvature be avoided. 

2. Provision is made for dissipation of the 
charge accumulated on the plane in a noise-free 
manner at a point remote from the antennas. 
Such discharge can be effected by means of a 
very thin wire, ending in a sharper point than 
any on the surface of the plane, trailing from 
the rear of the plane. A large-value resistor in 
series with this wire tends to damp the oscilla¬ 
tions ordinarily associated with the discharge. 


1812 AIRCRAFT ANTENNAS AND AIR 
DRAG 

All antennas projecting beyond the surface 
of the airplane are aerodynamic liabilities in 
that they are sources of parasitic drag. At ordi¬ 
nary subsonic velocities parasitic drag may be 
considered as consisting of two distinct types; 
frictional drag and form drag, which although 
interrelated in their effects will be considered 
separately. 

18,12,1 Frictional Drag 

Frictional drag is the resistance experienced 
by a moving body due to the viscosity of the air 
through which it moves. It is always propor¬ 
tional to the total surface area exposed to the 
airstream. Any moving surface is surrounded 
by a transition layer in which the air velocity 
relative to the surface increases from zero at 
the surface (neglecting the phenomenon of 
slip) to the full value of the stream velocity at 
the outer edge of the boundary layer. For low 
Reynolds numbers (the product of the air 


density, the stream velocity, the maximum 
linear dimension of the body normal to the 
stream, and the reciprocal of the coefficient of 
viscosity of the air) the flow in this boundary 
layer is laminar, consisting of layers in which 
all or almost all the fluid motion is parallel to 
that of the stream. Under this condition the 
coefficient of frictional drag is almost inde¬ 
pendent of the nature of the surface of the 
body, depending only upon the Reynolds num¬ 
ber and the shape of the body. At higher Reyn¬ 
olds numbers, above a certain critical velocity 
which depends upon the shape of the body, the 
flow in the boundary layer becomes turbulent 
and there is greater frictional drag. For turbu¬ 
lent flow, frictional drag is greater the rougher 
the surface. 

18,12,2 Form Drag 

Form drag is due to the disturbance created 
in the airstream by passage of the moving body 
and depends largely upon the shape of that 
body. For objects with sharp edges the form 
drag is virtually independent of Reynolds num¬ 
ber, being almost entirely due to the differ¬ 
ence in pressure upon the leading and trailing 
surfaces. For rounded bodies the form drag 
coefficient depends upon the Reynolds number, 
the surface roughness, and the degree of turbu¬ 
lence in the airstream. Such rounded bodies as 
spheres and cylinders may have smaller drag 
coefficients at high velocities than at low, the 
reason being that at low Reynolds numbers the 
boundary-layer flow is laminar, the flow sepa¬ 
rating on the leading side of the body, resulting 
in a wide wake and a large form drag, while at 
higher Reynolds numbers the boundary flow is 
turbulent and does not separate until it reaches 
the trailing side of the body, resulting in a nar¬ 
row wake and a correspondingly smaller drag. 
The magnitude of this effect can be startling. 
In the case of a sphere the drag coefficient sud¬ 
denly decreases sixfold when the velocity 
reaches the critical value at which turbulence 
sets in. Turbulence pre-existing in the air¬ 
stream reduces the critical velocity at which 
this decrease in form drag occurs. At still 
higher velocities, beyond the critical velocity, 
the drag coefficient rises slowly with increasing 



264 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


velocity until sonic speeds (75 per cent or more 
of that of sound) are reached. 

At sonic velocities the entire character of the 
airflow around the moving body changes, the 
leading surfaces setting up shock or compres¬ 
sion waves resulting in a type of drag known as 
wave drag. The wave-drag coefficient rises 
rapidly as the velocity of the body approaches 
that of sound, the total drag becoming much 
greater than that ascribable to form or friction. 

18.12.3 Calculation of Antenna Drag 

Tables of values of frictional- and form-drag 
coefficients are available in the literature of 
aerodynamics which also includes formulas by 
means of which the drag on a given antenna 
may be computed with reasonable accuracy for 
either laminar or turbulent flow. Wave drag is 
a relatively new phenomenon, encountered since 
the start of the war with the attainment of 
sonic velocities in dives with super-fast planes. 
Very little quantitative data information on 
wave drag was available in published litera¬ 
ture at the time this report was prepared. 

It should be remarked here that the total 
drag of a body moving at ordinary speeds may 
consist of frictional drag and form drag in 
almost any proportion, ranging from 100 per 
cent frictional drag for a properly designed 
streamline form to almost 100 per cent form 
drag for a smooth sharp-edged plate. 

An approximate semi-empirical formula for 
the total drag of a smooth circular cylinder—a 
shape common among conventional aircraft an¬ 
tennas—is 

D = 0.0030F 2 

where D is the drag in pounds per square foot 
of projected area and F is the velocity in mph. 
This formula results in good agreement with 
experiment for wire and rod antennas moving 
at moderate speeds. 

18.12.4 Measurement of Antenna Drag 

A more direct procedure, giving more satis¬ 
factory results when the antenna is of compli¬ 
cated shape or located in such a position on the 


plane that the assumptions underlying the drag 
formulas are not fulfilled, is to measure the 
drag of a model of the antenna mounted on a 
scale model of the plane by means of a wind 
tunnel. An alternative method is to mount the 
antenna on the actual plane and put it through 
all the maneuvers likely to be met in ordinary 
flight. 

18.12.5 gj m pi e Means of Reducing Drag 

• 

Frictional drag may be reduced by smooth¬ 
ing the antenna surface. While at low speeds 
the nature of the surface is more or less im¬ 
material, at high speeds (turbulent flow) a 
smooth surface is essential to low drag. 

Form drag may be greatly reduced by 
streamlining, that is, by so shaping the an¬ 
tenna that it produces little eddy-current dis¬ 
turbance as it passes through the air. The shape 
of the streamline form resulting in minimum 
drag depends upon the velocity, a ratio of 
major-to-minor axis of 2 or 3 being satisfactory 
for moderate speeds, of the order of 200 mph; 
larger ratios, i.e., thinner forms, are required 
at higher speeds. 

The effectiveness of streamlining in reducing 
drag is evident in Table 1, in which the drag in 
pounds per projected foot for standard circular 
aircraft cable is compared with that for stream¬ 
lined wire of similar nominal diameters. 


Table 1 . Effect of streamlining on antenna drag. 


Nominal diameter 
in inches 

Drag in pounds per projected foot at 
100 mph 

Circular cable 

Streamlined cable* 

0.25 

0.64 

0.056 

0.3125 

0.80 

0.067 

0.375 

0.96 

0.077 

0.50 

1.21 

0.092 


* Axis ratio 4:1. 


While antenna wires are rarely streamlined 
in practice because of the difficulty of maintain¬ 
ing the wire orientation in flight and because of 
the fact that the drag of the antenna fittings 
are usually much greater than that of the wire 
itself, it is worthwhile to streamline antennas 
of thick cylindrical form. 









AIRCRAFT ANTENNAS AND AIR DRAG 


265 


18.12.6 Qf] ier F ac tors Affecting Antenna 
Drag 

Obviously the length, cross section, and 
orientation of the antenna are important fac¬ 
tors in determining its total wind resistance. 
Unfortunately these parameters are determined 
by electrical considerations, which—since they 
are inextricably connected with the reason for 
having the antenna on the airplane in the first 
place—must be regarded as being at least as 
important as the matter of air drag. 

Antenna length is controlled by frequency. 
A conventional transmitting antenna, to be rea¬ 
sonably efficient and capable of even modest 
band width, must be of the order of A/4 long. 
Since wavelength is inversely proportional to 
frequency, the drag of a smooth-surfaced cylin¬ 
drical antenna will also be inversely propor¬ 
tional to frequency, other factors remaining 
constant. This situation is emphasized in Table 
2 in which the approximate drag of smooth 
cylindrical antennas 1 in. in diameter, moving 
at 200 mph, are compared for various resonant 
frequencies. 


Table 2. Antenna drag as a function of fre¬ 
quency. Drag on vertical quarter-wave antennas 
1 in. in diameter at 200 mph. 


Frequency 

(me) 

Antenna length 
(in.) 

Drag 

(lb) 

Wasted power 
(hp) 

3,000 

1 

0.831 

0.44 

300 

10 

8.3 

4.4 

30 

100 

83.0 

44.0 


These figures are only approximate, and in 
certain antenna installations may not even be 
of the right order of magnitude. They are in¬ 
tended only to support the following rule: If 
drag must be minimized , avoid low frequen¬ 
cies , particularly if the impedance and pattern 
requirements are such as to demand the use of 
a conventional exterior antenna. 

The cross-sectional dimensions of conven¬ 
tional antennas are controlled by the band 
width desired. Neglecting special cases in 
which the antenna impedance is markedly 
affected by its location on the particular plane 
involved, the fatter the antenna the greater its 
intrinsic band width, assuming that it can be 


fed in a manner which will not detract from 
that intrinsic band width. Other things being 
equal, if the impedance characteristics of a 
given h-f antenna of cylindrical form are to be 
duplicated at a lower frequency, the cross-sec¬ 
tional dimensions of the antenna must be 
scaled up in the same proportion as its length. 
The consequences of this fact on air drag as a 
function of frequency, for constant band width, 
is shown in Table 3. 


Table 3. Antenna drag as a function of fre¬ 
quency. Drag on vertical quarter-wave antennas 
of comparable band width at 200 mph. 


Frequency 

(me) 

Antenna 

length 

(in.) 

Antenna 

diameter 

(in.) 

Drag 

(lb) 

Wasted 

power 

(hp) 

3,000 

1 

0 25 

0.21 

0.11 

300 

10 

2.5 

21 

11 

30 

100 

25.0 

2,100 

1,100 


Again the purpose of the table is purely illus¬ 
trative, to show that, for conventional antennas 
at least, the desires for band width at low fre¬ 
quencies and for low drag are incompatible. 

Antenna orientation is controlled by the 
nature of the pattern and by the type of polar¬ 
ization required. For vertical polarization in 
the v-h-f and u-h-f bands, orientation is usually 
not a factor, since a vertical A/4 antenna is 
necessarily perpendicular to the airstream. For 
vertical polarization in the lower v-h-f and 
higher h-f ranges, where flat-top antennas must 
be used, the horizontal portion of the antenna 
should be strung back parallel to the line of 
flight. For horizontal polarization the antenna 
should be oriented so that it presents a mini¬ 
mum projected area to the wind stream, provid¬ 
ing such orientation is consistent with the 
nature of the field pattern desired. 

Special Low-Drag Antennas 

In many instances it is possible to satisfy the 
polarization, pattern, and impedance require¬ 
ments of a given problem by means of antennas 
having much less air drag than the simple 
wires, stubs, and whips to which the preceding 
discussion applies. A classic example of a satis- 































266 


AIRBORNE ANTENNA DESIGN AT U-H-F AND V-H-F 


factory low-drag antenna is the wire fan de¬ 
veloped by the Antenna Section, Research Divi¬ 
sion, Aircraft Radio Laboratory, and described 
elsewhere in this report. Other antennas hav¬ 
ing reduced drag to the extent that they are 
shorter than conventional radiators, are treated 
in Section 18.5.3. 

The most effective way to minimize drag is to 
remove the antenna from the airstream, placing 


it inside the skin of the ship. Several such an¬ 
tennas are described in Section 18.9. 

In many cases low-drag antennas of the types 
mentioned above will have impedance, pattern, 
or mechanical features making them unsuitable 
for a particular application, in which cases 
about all that can be done is to reduce friction 
by smoothing the antenna surface and to reduce 
form drag by streamlining. 




Chapter 19 

DEVELOPMENT OF FAIRED-IN ANTENNAS 


Development of a suitable device for exciting the sur¬ 
face of an all-metal plane to serve as the radiator. 
Slots, bars, etc., are looked upon as exciting devices and 
not as the primary radiators. Field pattern, surface 
current, and impedance measurements were made on 
scale-down models at a wavelength of 10 cm and on 
full-scale plane models using v-h-f frequencies. 

191 INTRODUCTION 

I F antennas in the v-h-f band are to be used 
on all-metal high-speed aircraft it is neces¬ 
sary that the antennas be streamlined into the 
contour of the airplane. This means that the 


exciting device that would not protrude, that 
would be compact, that would have suitable im¬ 
pedance characteristics, and that would give 
the required field pattern. 

Although the work was only well under way 
at the close of the war and the project termi¬ 
nated, much work was accomplished that will 
serve as background for its continuance under 
the Office of Naval Research. 

192 DEVICES INVESTIGATED 

The following current-exciting devices were 
investigated. 




JC 




n 

T 

Figure 1 . Schematic drawing of W-slot antenna. 


surface of the plane becomes an important 
component of the radiating system. In fact, 
the approach in this project a was to consider 
the current on the surface of the plane as the 
principal source of radiation. The plan was 
to investigate the possibility of designing an 

a Project 13-110, Problem No. 5, Contract OEMsr- 
1441, Harvard University. 


19,21 On Sheets at 10 Cm 

Slot Antennas 

Slot antennas are recognized by the presence 
of these features: (1) the surface to be ex¬ 
cited, (2) a cavity, (3) a slot to couple the 
cavity to the surface, and (4) a dipole or other 
exciting device to set up a field in the cavity. 


267 


































268 


DEVELOPMENT OF FAIRED-IN ANTENNAS 


W Slot. The advantage of this type of slot 
over most others is that the skin of the air¬ 
plane need not be cut except for the coaxial 
line. Two W slots were investigated having 
the following dimensions (see Figure 1) : 

W-l (short slot) 

l = 0.10a d = 0.29a w = 0.03a. 

W-2 (long slot) 

l = 0.60a d = 0.45a w = 0.03A. 

A Slot. The essential dimensions of this 
type of slot are shown in Figure 2. Two models 
were examined having these dimensions: 

A-l (narrow slot) w = 0.02a l = 0.64a. 

A-2 (broad slot) w = 0.315a l = 0.64a. 


reversing sleeves, one sleeve located at each 
end of the bar beneath the surface. The pur¬ 
pose of the phase-reversing sections is to in¬ 
crease the radiation resistance by spreading 
the current on the surface. If these sleeves 
are not present, the surface current is merely 
the image current of the bar. This current falls 
off very rapidly as one moves away from be¬ 
neath the bar. The radiation resistance is low 
without the sleeve because the image current 
and the bar current are in opposite directions 
and for this reason the field of the bar tends 
to be cancelled by the field of the image cur¬ 
rent. 

The following bar antennas were tested. 



Figure 2. Schematic drawing of A-slot antenna. 


H Slot. This type of slot (Figure 3) was 
used in an investigation of the proper location 
of an AN/APN-1 altimeter in a P4M bomber. 
(See Chapter 20.) 

C Slot. This slot (Figure 4) was devised to 
see what effect bending the slot back on itself 
would have on the surface current and field 
pattern. 


Bar Antennas 

A simple bar antenna consists of a metal rod 
parallel and very close to the surface to be ex¬ 
cited, together with two coaxial type phase- 


1. B-l end-fed bar. This was adapted from 
the B-2 center-fed bar by using an unbroken 
bar and by covering up the center section. The 
coaxial cable from the transmitter was con¬ 
nected to one of the coaxial fittings, the other 
fitting was connected to a variable-length line. 
This latter connection permitted locating the 
maximum current at the center of the bar. The 
bar length was about 0.5a. 

2. B-2 center-fed bar. The coaxial cable from 
the transmitter was connected to the center as 
shown in Figure 5. The two side fittings may 
be connected together by an adjustable length 
of coaxial line or each fitting may be attached 
separately to an adjustable length of line. In 






























DEVICES INVESTIGATED 


269 


either case the adjustment is for optimum lo¬ 
cation of the current maximum. 

3. 2B-1 parallel-bar antenna. Two B-l bars 
separated by about 0.5A (adjustable), driven 



phase difference adjusted for circular polariza¬ 
tion. 

5. 1 / 2 B-I or half-bar. For proper adjustment 
of the B-l antennas a voltage node appears in 



Figure 3. Schematic drawing of H-slot antenna. 




Figure 4. Drawing of C-slot antenna. 


either in phase or 180° out of phase. The 
length of the bars was about 0.5 a. 

4. 2BX-1 crossed-bar antennas. Two perpen¬ 
dicular B-l antennas of standard size with 


the middle of the bar. Therefore it can be con¬ 
nected directly to the surface at this point. 
This makes possible an antenna of half the 
length of the B-l bar. 

















































































270 


DEVELOPMENT OF FAIRED-IN ANTENNAS 


19 2 2 On Sheets in the V-H-F Band 

B-l and B-2 antennas were tested for their 
proper impedance characteristics. Various 
sizes and shapes of bars and phase-reversing 
sleeves were tested. 


Position 3 front right 0.47a from the nose 
of plane. 

Position 4 rear right 2.75a from the nose 
of plane. 

Total length of plane was 5.3 a. Combina- 



19,2,3 On a Scaled-Down Model of a P47N. 

Slot Antennas 

A W slot was adapted to the vertical stabi¬ 
lizer and tested for 360° coverage with good 
results. 

Bar Antennas 

The B-l end-fed bar was mounted vertically 
on the sides of the fuselage in the following 
positions: 

Position 1 front left 0.23a from the nose 
of plane. 

Position 2 front left 0.67a from the nose 
of plane. 


tions of the above positions using two B-l 
antennas were also used. 

Stub Antenna 

A vertical stub A/4 in length was mounted 
on top the fuselage at 2.87a from the nose. 
This antenna was useful for comparison pur¬ 
poses. 


i9.3 METHODS OF MEASUREMENT 

Measurements undertaken under the project 
were (1) the field patterns of the antennas as 
well as the effect of different conditions of ad¬ 
justment, (2) the distribution of surface cur- 






























































































METHODS OF MEASUREMENT 


271 


rent on the sheet of metal or the model plane 
on which the antenna was mounted. The re¬ 
sults of such measurements were to be prop¬ 
erly correlated. 


19.3.1 Field Pattern Measurements 

The antenna under test was excited through 
a 50-ohm coaxial cable from a modulated klys¬ 
tron oscillator. A double stub tuner matched 


modulation could be varied in frequency from 
500 to 5,000 cycles. 

The antenna system was mounted on a re¬ 
motely controlled turntable driven by geared- 
down synchronous motors and equipped with 
selsyns for remote indication of the azimuth 
position of the antenna. 

In addition the antenna could be rotated 
about a horizontal axis to give an “elevation” 
rotation. 

The receiving antenna consisted of a half- 



the line to the oscillator. Part of the oscillator 
output was fed to a crystal detector circuit 
which was part of the modulation monitor unit. 
The relative output of the oscillator could be 
checked by the detector current as read on 
a meter or by the deflection on the screen of 
a cathode-ray oscilloscope. 

The operating wavelength of the klystron 
oscillator was 10 cm while the square wave 


wave dipole at the focus of a 120-cm parabolic 
reflector and placed at a distance of about 130 a 
from the transmitter to insure far-zone condi¬ 
tions. The received signal was detected by a 
bolometer consisting of a 10-ma Littlefuse in¬ 
serted in the coaxial fittings of the paraboloid. 
The output of the bolometer was amplified 
(Figure 6) and the amplifier output was read 
on a d-c milliammeter or was recorded on an 

















































272 


DEVELOPMENT OF FAIRED-IN ANTENNAS 


external automatic recorder. The readings 
were directly proportional to the square of the 
electric field. 

Surface-Current Measurements 

In securing surface current measurements it 
is of primary importance that the indicating 
instrument response be determined by the cur¬ 
rents flowing in the small area of the surface 
surrounding the point investigated. A small 
loop placed in a plane perpendicular to the 
surface being excited satisfied this require¬ 
ment. At S-band frequencies, a loop area of 
from 1 to 2 sq cm gave good localization of 
response with sufficient sensitivity to permit 
measurement of very weak currents. 

The detecting element was a crystal. The 
square-wave modulation of the klystron (type 
410R at 80-100 watts input) made it possible 
to amplify the detector output. 

The final report gives the results of a great 
many tests and includes surface current dia¬ 
grams. 

1932 Impedance Measurements 

Delays in procuring a P47 plane limited the 
impedance measurements planned to actual 
v-h-f band data on B-l and B-2 antennas on 
flat sheets. Since measurements on a scaled- 
down model of the antennas at S-band fre¬ 
quencies were seen to involve many difficulties, 
investigations along these lines were under¬ 
taken only as a sideline. The results of the 
measurements made are contained in the final 
report 1 which also contains the theoretical an¬ 
alyses necessary for proper evaluation of the 
work accomplished. 

194 CONCLUSIONS 

Although the planned work was not com¬ 
pleted, due to the termination of the contract 
at the end of the war, certain conclusions were 
reached. One of the most striking results of 
the surface current investigation is that, at 
v-h-f frequencies, skin antennas usually excite 
the plane as a whole. Thus the behavior of 


such antennas is dependent on the general 
shape of the aircraft. A single antenna located 
properly on the side of the fuselage may be 
able to so excite the plane itself as to give 
360° coverage with no serious nulls. Also there 
may be a large horizontally polarized field from 
an antenna that, judging from the result on a 
flat sheet, should give only vertical polarization. 

On the other hand the interpretation of the 
surface-current patterns given in the final re¬ 
port 1 is seriously limited by a lack of knowl¬ 
edge of phase relationships. Further research 
on surface currents should include the mea¬ 
surement of relative phase along with mea¬ 
surement of direction and magnitude of cur¬ 
rent. 

Slot-antenna studies were very largely con¬ 
ventional. The bar antennas, however, repre¬ 
sent a significant departure from other skin 
antennas. Their importance lies not only in 
their merits as possible antennas but also in 
the fact that they emphasize the significance of 
current on the plane itself. With this view¬ 
point in mind, many other novel ways of ex¬ 
citing a plane at v-h-f should result from fur¬ 
ther research. Such research should lead to 
antennas capable of satisfying a great variety 
of requirements. 

The B-l and B-2 bar antennas differ de¬ 
cidedly in their behavior from that of a dipole 
mounted at the same distance (0.02a) from a 
flat sheet. The difference expresses itself in a 
greater extent of surface current excitation, a 
narrower beam width, and greater efficiency. 

Impedance measurements of the bar anten¬ 
nas were made only in the v-h-f band and at 
full scale where the phase-reversing sleeves 
were very much smaller in diameter relative 
to the wavelength and the distance of the bar 
from the sheet than they were in the 10-cm 
models. Here the band width for a 2:1 stand¬ 
ing wave ratio was about 0.8 per cent. Thus 
the bar with phase reversing sleeves of very 
small diameter is serviceable as it stands at 
one frequency only. If the reversing stubs are 
tuned by remote control, a very broad band is 
possible. 

The fact that a plane may be excited as a 
whole by a bar antenna or other device makes 
it seem likely that its impedance is changed 



CONCLUSIONS 


273 


considerably from that as measured on a flat 
sheet. To investigate this effect a full-scale 
plane mounted on a platform so that it is suf¬ 
ficiently decoupled from its ground image, or 
a careful scaling down of the plane, the exciting 
antenna, and the antenna feed systems, is neces¬ 
sary. 

One of the conclusions that was drawn from 
the measurement of surface currents on the 
model plane was not only that currents may be 
large over the entire plane, but that currents 
remote from the antenna may be of primary 
importance in determining the field. Hence 


the shape of the plane may, at v-h-f frequen¬ 
cies, materially affect the impedance at the ter¬ 
minals of the exciting antenna. 

In the appendices of the final report 1 will 
be found certain analytical studies useful to 
any work in this field. These subjects include 
“Surface current distributions that produce 
circular horizontal polarization’'; “Broad-band 
characteristics of a dipole using a series trans¬ 
former as a matching section”; “On the proper 
spacing of insulating beads”; and “A conver¬ 
sion chart for impedance measurement using 
transmission line.” 





Chapter 20 


MISCELLANEOUS ANTENNA RESEARCH 


20..1 LOCATION OF ANTENNA FOR 

AN/APN-1 ALTIMETER ON 
NAVAL AIRCRAFT 

T his project was set up to study the problem 
of locating H-type slot antennas for mini¬ 
mum feed-through on Navy type P4M aircrafts 
The study indicated that the slots should be 
located on the opposite sides of the horizontal 
stabilizer undersurface and that for absolute 
minimum feed-through the slots would have to 
be approximately perpendicular to each other, 
thus seriously affecting the operation of the 
altimeter. 


2011 Introduction 

Since the AN/APN-1 altimeter operates in 
the region 420 to 460 me, it was decided to use 
a 1/7 scale-down in constructing a model P4M 
jet-assist bomber tail section and to employ a 
frequency of 3,080 me. The plan was to mea¬ 
sure the surface current distribution on the hori¬ 
zontal stabilizer surface and on the surface of 
the fuselage in the vicinity of the stabilizer for 
a wide variety of positions of the transmitter 
H slot. These measurements were to determine 
lines of flow and contours of constant surface- 
current amplitude. Then for each position of 
the transmitting slot, positions for the receiv¬ 
ing slot would be chosen: 

1. Only in regions permitted by the internal 
structure of the plane. 

2. So that the angle formed by the lines of 
orientation of the transmitting and receiving 
slots would not exceed 45°. 

3. At positions of minimum surface-current 
amplitude. 

4. So that the receiving slot would be 
oriented parallel to the lines of surface-current 
flow in its vicinity. 

a Project 13-110, Problem No. 7, Contract OEMsr- 
1441, Harvard University. Originally Project 13-108. 


Conditions (1) and (2) constitute limitations 
imposed by the practical location of the slots 
and acceptable altimeter performance. Condi¬ 
tions (3) and (4) constitute limitations on the 
location of slots for minimum feed-through 
caused by surface-current coupling. 

2012 Laboratory Technique 

To obtain accurate measurements of feed¬ 
through it was necessary to minimize direct re¬ 
flection of energy from one antenna to the 
other. This made it desirable to simulate flight 
conditions by mounting the model on a platform 
far from ground and upside down so that the 
H slots would be directed skyward. Difficulties 
in getting the platform delayed this part of the 
study until near the time the experimental 
work was terminated. 

Concurrent with surface-current measure¬ 
ments, attempts were made to determine a 
satisfactory method of measuring absolute 
feed-through. The problems of establishing a 
proper reference level, of matching, and of cable 
losses all had to be solved before the actual 
feed-through data could be collected. 

As a first step in this direction a flat metal 
sheet was constructed so that it could be driven 
by a waveguide slot. Holes were cut in the 
sheet at positions of different current ampli¬ 
tude. A rotatable H-slot mounting disk was de¬ 
signed and constructed so that when located in 
any one of the holes it could be rotated con¬ 
tinuously through 360° and clamped in any 
desired position. With this setup an investiga¬ 
tion was made as to the correlation between 
feed-through and (1) surface-current ampli¬ 
tude at the receiving H-slot position and (2) 
the angle between H-slot direction and lines of 
surface-current flow in the vicinity of the slot. 
The results obtained are of value in estimating 
the extent to which the feed-through is mini¬ 
mized by locating a receiving H slot according 
to conditions (3) and (4) above. 


274 



PROBLEMS ARISING FROM CLOSELY GROUPED ANTENNAS 


275 


Conclusions 

The final report 1 gives a description of the 
experimental equipment employed and the re¬ 
sults of the measurements to the end of the 
contract. Work of a somewhat more general 
nature is continuing under a contract with the 
Office of Naval Research. 

The conclusions cited below are tentative. 

Surface-current coupling between transmit¬ 
ting and receiving antennas can be minimized 
by choosing the slot positions in such a way 
that the receiving slot is located in a region of 
minimum surface-current amplitude and orient¬ 
ing the receiving slot so that it is parallel to the 
lines of surface-current flow in its vicinity. 

Measurements made indicate that minimum 
feed-through values of between 70 and 80 db 
down may be reasonable and that values be¬ 
tween 90 and 100 db down may not be beyond 
the realm of possibility. 

It may be necessary to orient the slots at an 
angle with respect to the line of flight. Surface 
currents are smaller on the stabilizer surface 
opposite to the side on which the transmitting 
slot is located and less than on the bottom sur¬ 
face of the fuselage section adjacent to the hori¬ 
zontal stabilizer. 

To achieve best final results in locating the 
slots, the measurement technique may have to 
be carried out on a full-scale mock-up of the 
tail assembly mounted at a sufficient height 
above ground so that the presence of the ob¬ 
server and observing equipment may be less¬ 
ened in its power to affect the measurements. 

202 STUDY OF PROBLEMS ARISING 

FROM CLOSELY GROUPED ANTENNAS 

Introduction 

Experience in the theaters has indicated that 
the first practical step in minimizing the sever¬ 
ity of local radio interference in a headquarters 
area is to establish separate sites for groups of 
transmitting and receiving antennas. The prin¬ 
cipal purpose of the survey conducted under 
Project 13-103 b was to determine the minimum 

b Project 13-103, Contract No. OEMsr-1412, Western 
Electric Co. 


required separation between such transmitting 
and antenna “parks, 1 ” and between individual 
antennas within an antenna park. This sepa¬ 
ration is largely a function of certain spurious 
interference-producing properties of existing 
military radio sets, and on the coupling between 
various antenna types over different types of 
soil. Some data on spurious radiations and 
responses of radio sets were obtained in earlier 
work under Project C-79 3 (Contract OEMsr- 
1018), and these were supplemented by addi¬ 
tional measurements on a number of sets. Be¬ 
cause of the theater needs, this information on 
set characteristics was incorporated as part of 
War Department publication TM 11-486 4 pre¬ 
pared by the contractor prior to publication of 
the final report on Project 13-103. 

Results of the Survey 

The final report 2 on the project contains 
information for estimating the required sepa¬ 
ration between transmitting and receiving an¬ 
tenna parks for both h-f and v-h-f tactical radio 
circuits, separations which should exist between 
individual antennas in an antenna park, and 
the relative advantages of the several methods 
of connecting several receivers to a single 
antenna. 

Considerable information is given on trans- 
mitter-to-receiver interference as a result of 
spurious radiation at harmonics of the master 
oscillator, spurious outputs caused by interfer¬ 
ence between transmitters, effect of radiation 
from receivers, and spurious responses of 
superheterodyne types of receivers, with curves 
and charts enabling one to predict where such 
undesired receiver responses will occur in fre¬ 
quency. 

Separation between transmitting and receiv¬ 
ing antennas is considered from several angles 
and data given in tables and charts taking into 
account the types of antennas employed, the 
ground characteristics, the weakest usable sig¬ 
nals, and the tolerable r-f interference-to-signal 
ratios. 

Suggested layouts of h-f sky-wave transmit¬ 
ting or receiving antenna parks are given based 
on (1) assigned frequencies being divided into 
groups in such a way that the frequencies of 





276 


MISCELLANEOUS ANTENNA RESEARCH 


any pair within the group are not less than 10 
per cent apart in frequency, (2) half-wave an¬ 
tennas for the frequencies within a group being 
placed parallel to each other and about 5 ft 
apart, and (3) antenna groups being located 
about 250 ft apart. 

Similar layouts are given in the final report 
when other types of transmission are utilized, 
ground wave signals, for example, for other 
types of antennas, etc. 

Considerable data are given on the mutual 
impedance between coupled antennas over an 
imperfectly conducting earth, and on possible 
methods of connecting several receiving an¬ 
tennas to a common antenna. 

20 3 STUDY OF IMPROVISED V-H-F 
ANTENNAS 

203 1 The Problem 

Reports from combat areas indicated that 
dipole antennas made from ordinary field wire 
and using paired field wire for feed lines might 
be used when standard antennas and feed lines 
were not available. The purpose of this project 0 
was to evaluate the losses in such systems and 
to suggest effective arrangements which could 
readily be improvised from available materials. 

20 - 3 - 2 The Solution 

The type of wire in common theater use con¬ 
sisted largely of ordinary field wire (W-110-B), 
long-range tactical wire (W-143) and spiral- 
four cable (WC-548). Measurements indicated 
that the losses in these wires would attain 
values as high as 10 to 25 db or more per 100 ft 
at 100 me. At 30 me, ordinary field wire when 
wet had losses as high as 15 db per 100 ft. 

The high losses of ordinary wire used as a 
transmission line indicated the use of spaced 
leads for the feed line. An improvised antenna 
consisting of a half-wave dipole and a spaced 
line with a quarter-wave matching section at 
each end operated satisfactorily. The line con¬ 


c Project 13-102, Contract OEMsr-1411, Western 
Electric Co. 


sisted of two conductors formed by field wire 
spaced about 2 in. apart while the quarter-wave 
matching sections at each end of the line con¬ 
sisted of suitable lengths of paired W-110-B 
wire. With such a line 100 ft long, the power 
radiated from a transmitter was only a few db 
less than it would have been with a flexible 
coaxial cable. The actual losses were 2 to 4 db 
greater at 30 to 40 me and 3 to 6 db greater at 
70 to 100 me. With the line wet these values 
were increased an additional 1 to 3 db. These 
losses were relatively unimportant when receiv¬ 
ing unless the signals were marginal. 

Loss characteristics, figures illustrating an¬ 
tennas hung from trees, etc., will be found in 
the contractor’s final report. 5 

20 4 DISGUISED ANTENNAS 

2041 Introduction 

The problem 01 was to design an antenna that 
would not project into the air, revealing the 
presence of the radio set to which it was con¬ 
nected. The research was confined to the por¬ 
table radio set SCR-300 which is ordinarily 
used with one of two antennas, one being 10 ft 
8 in. long, the other being 33 in. long with a 
parallel loading circuit grounded to the case of 
the set. 

The tests were made largely in the field, one 
pack set using the improvised antenna and the 
other the standard 33-in. antenna. The testing 
procedure consisted in comparing the impro¬ 
vised antenna with the standard collector under 
the same conditions. 


20.4.2 Results of Field Tests 

The most promising disguised antenna tested 
was that employing helmet and counterpoise. 
A short length of wire connected the helmet to 
the parallel matching section. Another wire, 
connected to the ground terminal of the pack 
set and serving as counterpoise, extended 
almost to the ground. The latter could probably 


d Project 13-110, Problem No. 10, Contract OEMsr- 
1441, Harvard University. 





DISGUISED ANTENNAS 


277 


be sewed into the trouser leg. Good signal 
strength and intelligibility was possible over 
ranges of 1 to 3 miles. If the operator’s head 
was less than 15 in. above the ground the maxi¬ 
mum range was about one mile. 

Another promising arrangement was to use 
the pack set itself as antenna and to drive it 


against the ground or the operator’s body. An 
L-type matching section was required. The 
range was about the same as that described 
above. 

Tests in which the antenna wire was sewed 
into the clothing were not so successful as the 
other schemes devised. 




BIBLIOGRAPHY 


Numbers such as Div. 13-101-MI indicate that the document listed has been microfilmed and that its 
title appears in the microfilm index printed in a separate volume. For access to the index volume and to the 
microfilm, consult the Army or Navy agency listed on the reverse of the half-title page. 


Chapter 1 

1. High Frequency Direction Finder Research, Karl 

G. Jansky, OSRD 209, NDSrc-155, Research Proj¬ 
ect C-16, Bell Telephone Laboratories, Inc., Novem¬ 
ber 1941. Div. 13-101-MI 

High Frequency Direction Finder Research, Karl 
G. Jansky, OSRD 699, NDCrc-155, Research Proj¬ 
ect C-16, Bell Telephone Laboratories, Inc., June 1, 
1942. Div. 13-101-M2 

2. “The Optical Behavior of the Ground for Short 
Radio Waves,” C. B. Feldman, Proceedings of the 
Institute of Radio Engineers, Vol. 21, June 1933, 
pp. 764-801. 

3. “Some Principles Underlying the Design of Spaced 
Aerial Direction Finders,” R. H. Barfield, Journal 
of the Institution of Electrical Engineers, Vol. 76, 
No. 460, April 1935, p. 425. 

Chapter 2 

1. High Frequency Direction Finder Apparatus Re¬ 
search, Harry M. Diamond, Harold Lifschutz, and 
LaVerne M. Poast, Research Project C-18, National 
Bureau of Standards, July 1, 1942. Div. 13-101-M3 
la. Ibid., pp. 61-62. 

2. “Some Principles Underlying the Design of Spaced 
Aerial Direction Finders,” R. H. Barfield, Journal 
of the Institution of Electrical Engineers, Vol. 76, 
No. 460, April 1935, pp. 424-443. 

3. High-Frequency Direction Finder Apparatus Re¬ 
search, National Bureau of Standards, Nov. 15,1941. 

4. Wave Collectors for Semi-Portable Radio Direction 

Finders for High Frequencies, D. G. C. Luck and 
L. E. Norton, OSDR 337, NDCrc-149, Research 
Project C-17, Radio Corporation of America, Jan. 21, 
1942. Div. 13-101.2-MI 

5. The Polarization of Downcoming Ionospheric Radio 
Waves, K. A. Norton, Report 60047, Federal Com¬ 
munications Commission—National Bureau of 
Standards, May 1942. 

6. Nature of Sky-Wave Propagation, K. A. Norton, 
Fourth Annual Broadcast Engineering Conference, 
Ohio State University, February 1941. 

7. Program of Continued Research and Development 
on Polarization Errors in Short-Wave Direction 
Finding, NDRC, International Telephone and Radio 
Laboratories, June 3, 1942. 


8. “Radio Propagation Over Plane Earth-Field 
Strength Curves,” Charles R. Burrows, Bell System 
Technical Journal, Vol. XVI, No. 1, January 1937, 
pp. 45-75. 

9. “Addendum To: Radio Propagation Over Plane 
Earth-Field Strength Curves,” Charles R. Bur¬ 
rows, Bell System Technical Journal, Vol. 16, No. 4, 
October 1937, pp. 574-577. 

10. “The Effect of the Earth’s Curvature on Ground- 
Wave Propagation,” Charles R. Burrows and Marion 
C. Gray, Proceedings of the Institute of Radio Engi¬ 
neers, Vol. 29, January 1941, pp. 16-24. 

11. A Course in Modern Analysis, Edmund Taylor 
Whittaker and G. N. Watson, (Cambridge Univer¬ 
sity Press, British Edition, 1927,) Cambridge Uni¬ 
versity Press and Macmillan Company, American 
Edition, 1942, p. 341. 

12. “Uber die Methode der Kleinster Quadrate,” J. F. 
Encke, Berliner Astronomisches Jahrbuch, 1834, 
pp. 249-304. 

13. Transactions of the Royal Society of Edinburgh, 
Vol. XXXIX, 1900, p. 257. 

14. Functionentafeln, Eugene Jahnke and Fritz Emde, 
B. G. Teubner, Leipzig, 1933. 

15. Summary of British DF System, E. D. Blodgett, 
NDRC, July 1942. 

16. Radio Research Board Report, W. Ross, Project 
RRB/C-43, February 1939. 

17. Radio Research Board Report, R. H. Barfield and 
B. G. Pressey, Project RRB/C-4, February 1939. 

18. Radio Research Board Report, W. Ross, Project 
RRB/C-3, March 1940. 

19. Correlations for Polarization Errors of Bearings 
with the DY Direction Finders, National Bureau of 
Standards, Dec. 11, 1941. 

20. Preliminary Measurements of a Navy Model DY 
Direction Finder, National Bureau of Standards, 
Dec. 9, 1941. 

21. Polarization Errors of the SCR-551 Direction 
Finder, National Bureau of Standards, Mar. 12, 
1942. 

22. Polarization Errors of the W.E.-C.A.A. Direction 
Finder, National Bureau of Standards, Apr. 20, 
1942. 


278 


BIBLIOGRAPHY 


279 


1 23. “Simultaneous Radio Range and Telephone Trans¬ 
mission,” W. E. Jackson and D. M. Stuart, Proceed¬ 
ings of the Institute of Radio Engineers, Vol. 25, 
March 1937, pp. 314-326. 

24. (Unpublished report) H. W. Kohler, Civil Aero¬ 
nautics Authority, Mar. 30, 1942. 

25. Polarization Errors of a Direction Finder with 
Spaced Loop Antennas, National Bureau of Stand¬ 
ards, June 25, 1942. 

26. Polarization Error Test of the CXAL Direction 
Finder, Collins Radio Company and National Bu¬ 
reau of Standards, Mar. 10, 1942. 

Chapter 3 

1. Study of Radio Pulse Propagation, Karl G. Jansky, 
OSRD 599, OEMsr-310, Western Electric Company, 
Inc. and, Bell Telephone Laboratories, Inc., May 1, 
1942. Div. 13-101.1-MI 

2. “A Note on the Theory of Night Errors in Adcock 
Direction Finding Systems,” J. F. Coales, Journal 
of the Institution of Electrical Engineers, Vol. 71, 
No. 429, September 1932, pp. 497-506. 

3. “Echoes from Nearby Short-Wave Transmitters,” 
C. F. Edwards and Karl G. Jansky, Proceedings of 
the Institute of Radio Engineers, Vol. 29, June 1941, 
pp. 322-329. 

Chapter 4 

1. Ultra High Frequency Direction Finding Study, 
E. D. Blodgett, L. L. Lakatos, and others, OSRD 
4285, OEMsr-1009, Froject 13.1-82, RCA-Victor, 
July 29, 1944. Div. 13-102.2-M2 

2. “Some Principles Underlying the Design of Spaced 
Aerial Direction Finders,” R. H. Barfield, Journal 
of the Institution of Electrical Engineers. Vol. 76, 
No. 460, pp. 423-443, April 1935. 

3. High Frequency Direction Firtder Apparatus Re¬ 
search, Harry Diamond, Harold Lifschutz, and 
LaVerne M. Poast, Research Project C-18, National 
Bureau of Standards, July 1,1942. Div. 13-101-M3 

4. “A Determination of the Electrical Constants of 
the Earth’s Surface at Wavelengths of 1.5 and 0.46 
M,” J. S. McPetrie, Proceedings of the Physical So¬ 
ciety, Vol. 46, 1934, p. 637. 

5. “Steady State Solutions of Electromagnetic Field 
Problems, Part III, Forced Oscillations of a Prolate 
Spheroid,” J. A. Stratton and L. J. Chu, Journal of 
Applied Physics, Vol. 12, 1941, p. 241. 

6. “The Self-Impedance of a Symmetrical Antenna,” 
Ronold King and F. G. Blake, Jr., Proceedings of 
the Institute of Radio Engineers, Vol. 30, July 1942, 
p. 335. 


7. “The Distribution of Current Along a Symmetrical 
Center Driven Antenna,” Ronold King and C. W. 
Harrison, Jr., Proceedings of the Institute of Radio 
Engineers, Vol. 31, October 1943, p. 548. 

“The Radiation Field of a Symmetrical Center 
Driven Antenna of Finite Cross Section,” C. W. 
Harrison, Jr. and Ronold King, Proceedings of the 
Institute of Radio Engineers, Vol. 31, December 
1943, p. 693. 

8. “Theory of Antennas of Arbitrary Size and Shape,” 
S. A. Schelkunoff, Proceedings of the Institute of 
Radio Engineers, Vol. 29, September 1941, p. 493. 

9. Electromagnetic Waves, S. A. Schelkunoff, D. Van 
Nostrand Company, New York, 1943. 

10. “Directional Antennas,” G. H. Brown, Proceedings 
of the Institute of Radio Engineers, Vol. 25, January 
1937, p. 78. 

11. “Antenna Theory and Experiment,” S. A. Schelku¬ 
noff, Journal of Applied Physics, Vol. 15, No. 1, 
January 1944, pp. 54-60. 

12. “A Modification of Hallen’s Solution of the Antenna 
Problem,” Marion C. Gray, Journal of Applied 
Physics, Vol. 15, No. 1,1944, pp. 61-65. 

13. Transient Electrical Phenomena and Oscillations, 
Charles P. Steinmetz, McGraw-Hill Book Company, 
New York, 1920. 

14. Functionentalfeln, Eugene Jahnke and Fritz Emde, 
B. G. Teubner, Leipzig, 1933. 

15. “High-Frequency Models in Antenna Investiga¬ 
tions,” G. H. Brown and Ronold King, Proceedings 
of the Institute of Radio Engineers, Vol. 22, April 
1934, p. 457. 

Chapter 5 

1. Wave Collectors for Semi-Portable Radio Direction 
Finders for High Frequencies, D. G. C. Luck, OSRD 
337, NDCrc-149, Research Project C-17, Radio Cor¬ 
poration of America, Jan. 21, 1942. 

Div. 13-101.2-MI 

2. Further Studies of Errors in High-Frequency Direc¬ 

tion Finders, D. G. C. Luck, OSRD 908, OEMsr-338, 
Research Project C-38, Radio Corporation of Amer¬ 
ica, Aug. 25, 1942. Div. 13-101.2-M3 

3. Polarization Errors of Shielded-U Adcock Direction 
Finders, D. G. C. Luck and L. E. Norton, OSRD 
1653, OEMsr-838, Research Project C-57, Radio 
Corporation of America, July 20, 1943. 

Div. 13-101.21-M2 

The Measurement of Errors of Radio Direction 
Finders, D. G. C. Luck and L. E. Norton, OSRD 1884, 
OEMsr-838, Research Project C-78, Radio Corpora¬ 
tion of America, June 10, 1943. Div. 13-101.2-M6 













280 


BIBLIOGRAPHY 


4. “Radio Propagation over Plane Earth,” Charles R. 
Burrows, Bell System Technical Journal, Vol. 16, 
January 1937, p. 45. 

5. High Frequency Direction Finder Apparatus Re¬ 
search, Harry Diamond, Harold Lifschutz, and La- 
Verne M. Poast, Research Project C-18, National 
Bureau of Standards, July 1, 1942. 

Div. 13-101-M3 

6. “Theory of Reflection of the Light from a Point 
Source by a Finitely Conducting Flat Mirror with 
an Application to Radiotelegraphy,” B. van der Pol, 
Physics, Vol. 2, August 1935, pp. 843-853. 

7. The Polarization of Downcoming Ionospheric Radio 
Waves, K. A. Norton, Report 60047, Federal Com¬ 
munications Commission—National Bureau of 
Standards, May, 1942 

8. Polarization Errors of Shielded-U Adcock Direction 
Finders, D. G. C. Luck and L. E. Norton, OSRD 
1653, OEMsr-838, Research Project C-57, Radio 
Corporation of America, July 20, 1943. 

Div. 13-101.21-M2 

The Measurement of Errors of Radio Direction 
Finders, D. G. C. Luck and L. E. Norton, OSRD 1884, 
OEMsr-838, Research Project C-78, Radio Corpora¬ 
tion of America, June 10, 1943. Div. 13-101.2-M6 

9. Study of Direction Finder Fundamentals, H. 
Busignies and D. Baker, OSRD 6608, OEMsr-745, 
Research Project C-58, Federal Telecommunication 
Laboratories, Inc., Dec. 17, 1945. Div. 13-101.2-M7 

10. Demountable Short Wave Direction Finder, Type 

SCR-502, H. Busignies and A. G. Richardson, 
OSRD 1634, OEMsr-262, Research Project C-34, 
Federal Telephone and Radio Corporation, July 1, 
1943. Div. 13-102-MI 

11. Improvement of Band U of NLS-505 Direction 
Finder, Trevor H. Clark and Henry B. Scarborough, 
OSRD 3318, OEMsr-1026, Project 13.1-84, Federal 
Telephone and Radio Corporation, December 1943. 

Div. 13-101.21-M3 

Investigation of Site Characteristics Which Lead to 
Errors in Direction Finders, Trevor H. Clark and 
Henry B. Scarborough, OSRD 5022, OEMsr-1026, 
Project 13.1-84, Federal Telephone and Radio Cor¬ 
poration, Mar. 15, 1945. Div. 13-101.21-M4 

12. Investigation of Compensation in Direction Finders, 
Joseph M. Pettit, OSRD 508, NDCrc-159, Research 
Project C-19, Stanford University, Apr. 7, 1942. 

Div. 13-101.2-M2 

13. Miscellaneous Current Direction Finding Problems, 
Trevor H. Clark and N. Marchand, OSRD 6657, 
OEMsr-1490, Project 13-122, Federal Telephone and 
Radio Corporation, Sept. 30, 1945. 

Div. 13-101-M6 


14. Lorenz: German Patent No. 624706; Hell: German 
Patent No. 601-904-1933; Telefunken: British Pat¬ 
ent No. 464075-1937. 

15. “Compensation Loop Direction Finders,” Joseph M. 
Pettit and A. W. Terman, Proceedings of the Insti¬ 
tute of Radio Engineers, May 1945. 

16. Study of Direction Finder Fundamentals, H. Busig¬ 
nies and D. Baker, OSRD 6608, OEMsr-745, Re¬ 
search Project C-58, Federal Telecommunication 
Laboratories, Inc., July 31, 1943. 

Div. 13-101.2-M7 

17. Study of the Direction Finder Fundamentals, H. 
Busignies, Research Project C-58. Federal Tele¬ 
phone and Radio Corporation, Sept. 28, 1943. (Re¬ 
port summarized in Div. 13-101.2-M7.) 

18. Study of Direction Finder Fundamentals, Polariza¬ 
tion Study, H. Busignies, Research Project C-58, 
Federal Telephone and Radio Corporation, May 28, 

1943. (Report summarized in Div. 13-101.2-M7.) 

Chapter 6 

1. Coordinated Study of Ionospheric Transmission and 

Direction Errors at High Radio Frequencies, T. R. 
Gilliland, Research Project C-13, National Bureau 
of Standards, Apr. 26, 1943. Div. 13-101.3-Ml 

2. Coordinated Study of Correlations of High-Fre¬ 
quency Direction-Finder Errors with Ionospheric 
Conditions, La Verne M. Poast, Research Project 

13.2- 92, National Bureau of Standards, Aug. 31, 

1944. Div. 13-101.3-M5 

3. Correlation of Direction Finder Errors with Iono¬ 

spheric Measurements, R. A. Helliwell, OSRD 3982, 
OEMsr-1122, Project 13.2-88, Stanford University, 
July 18, 1944. Div. 13-101.3-M4 

4. Direction Finder Measurements Research, G. W. 
Kendrick, OSRD 4528, OEMsr-1101, Project 13.2-90, 
University of Puerto Rico, June 30, 1944. 

Div. 13-101.3-M2 

5. Correlation of Direction-Finder Errors with Iono¬ 
spheric Conditions, College, Alaska, August 16,19US 
to June 30, 19UU, H. W. Wells, S. L. Seaton, and 
E. H. Bramhall, OSRD 4225, OEMsr-1151, Project 

13.2- 91, Carnegie Institution of Washington, Sept. 

8.1944. Div. 13-101.3-M6 

6. Correlation of Direction-Finder Errors with Iono¬ 

spheric Measurements, Harry Rowe Mimmo, OSRD 
3981, OEMsr-1252, Project 13.2-99, Harvard Uni¬ 
versity, July 15, 1944. Div. 13-101.3-M3 

Chapter 7 

1. Tests on Direction-Finder Systems, Harry Rowe 
Mimno, OSRD 6280, Final Report, Part III, on Con¬ 
tract OEMsr-1441, Problem No. 1, Service Project 
AN-30, Cruft Laboratory, Harvard University, Dec. 

1.1945. Div. 13-100-M2 



BIBLIOGRAPHY 


281 


2. Survey of Airborne Direction Finders, John L. 
Allison, OSRD 5038, Service Project AN-22, Feb. 
15, 1945. Div. 13-100-MI 

Chapter 8 

1. Ultra High Frequency Radio-Sonde Direction 
Finder, Luke Chia-Liu Yuan, OSRD 1256, OEMsr- 
217, Research Project C-33, California Institute of 
Technology, Feb. 8, 1943. Div. 13-102.2-MI 

Chapter 9 

1. Demountable Short Wave Direction Finder, Type 
SCR-502, H. Busignies and A. G. Richardson, OSRD 
1634, OEMsr-262, Research Project C-34, Federal 
Telephone and Radio Corporation, July 1, 1943. 

Div. 13-102-MI 

War Department Manual TM 11-256. 

Chapter 10 

1. Direction Finding by Improvised Means, A. J. 
Aikens and A. G. Chapman, OSRD 4608, OEMsr- 
1410, Project 13-101, Western Electric Company, 
Inc., and Bell Telephone Laboratories, Inc., Nov. 30, 
1944. Div. 13-101-M5 

Chapter 11 

1. Portable Radio Assault Beacon, Samuel J. Snyder, 
OSRD 4058, OEMsr-1261, Project 13.1-100, Wilmotte 
Laboratory, Inc., Aug. 15, 1944. Div. 13-102-M2 

Chapter 12 

1. Ultra High Frequency Direction Finding Antenna 
Study, Trevor H. Clark and E. Daubaras, OSRD 
5103, OEMsr-961, Research Project C-80, Federal 
Telephone and Radio Corporation, Apr. 15, 1945. 

Div. 13-104-M3 

Chapter 13 

1. Locating Tanks by Radio, C. W. Harrison, OSRD 
963, OEMsr-787, Research Projects C-60 and SC-31, 
Bell Telephone Laboratories, Inc., Oct. 15, 1942. 

Div. 13-102.1-MI 

2. Locating Tanks by Radio, C. G. Fick, OSRD 1542, 

OEMsr-737, Research Project C-61, General Electric 
Company, June 4, 1943. Div. 13-102.1-M2 

Chapter 14 

1. U-H-F Friendly Aircraft Locator, G. C. Larson and 
A. V. Loughron, OSRD 102, NDCrc-193, Research 
Project C-12, Report 1430-W, Hazeltine Service Cor¬ 
poration, Nov. 11, 1942. Div. 13-102.21-M8 

Instruction Book, U-H-F Friendly Aircraft Locator, 
NDCrc-193, Service Project SCr-552, Report 1429- 
W, Hazeltine Service Corporation, Dec. 2, 1942. 

Div. 13-102.21-M9 


Chapter 15 

1. Electrical Direction Finder Evaluator, John L. Al¬ 
lison, John H. Lewis, and H. C. Fryer, OSRD 6551, 
OEMsr-1472, Service Project SC-130, J. A. Maurer, 
Inc., Oct. 31, 1945. Div. 13-102-M3 

Chapter 16 

1. Meteorological Information from Sferic Pulses, 
Progress Report, April 27 to May 31, 19U5, R. E. 
Holzer, OEMsr-1485, Report UNM/SC-2, University 
of New Mexico, June 1, 1945. Div. 13-103.1-MI 

2. A Study of Sferics and Weather Information, R. E. 
Holzer, OSRD 6442, OEMsr-1485, Report UNM/ 
SC-5, University of New Mexico, Nov. 30, 1945. 

Div. 13-103.1-M2 

3. “Electrical Structure of Thunderstorms,” E. J. 
Workman, R. E. Holzer, and G. T. Pelzor, Technical 
Note 864, National Advisory Committee for Aero¬ 
nautics, November 1942. 

4. Technical Manual, Static Direction Finder AN/ 
GRD-1, War Research Laboratory, Engineering Ex¬ 
perimental Section, University of Florida. 

5. “Wave Form, Energy and Reflection by the Iono¬ 
sphere of Atmospherics,” T. H. Laby, J. J. McNeill, 
F. G. Nicholls, and A. F. B. Nickson, Proceedings of 
the Royal Society, Ser. A, Vol. 174, February 1940, 
pp. 145-163. 

6. “The Wave Form of Atmospherics at Night,” B. F. 
J. Schonland, J. S. Elder, D. B. Hodges, W. E. Phil¬ 
lips, and J. W. van Wyk, Proceedings of the Royal 
Society, Ser. A. Vol. 176, August to November 1940, 

pp. 180-202. 

7. Roy Lutkin, Journal of the Meteorological Society of 
London, Vol. 67,1941, p. 345. 

8. “Veber Gewitterregistrierung,” Jean Lugeon, Bul¬ 
letin d’Association Suisse des Electriciens, Vol. 34, 
No. 2, Jan. 27,1943, pp. 29-43. 

9. Boswell and Wark, Journal of the Royal Meteorolog¬ 
ical Society, Vol. 62, 1936, p. 499. 

10. Report on Statistical Direction Finder Research, 
Lockhart, Navy Department, Bureau of Engineer¬ 
ing, May 28, 1937. 

11. Harry Lowe Mimno, Reviews of Modern Physics, 
Vol. 9, 1937, p. 1. 

Chapter 17 

1. Antenna Patterns for Aircraft, George Sinclair, 
OSRD 886, NDCrc-100, Service Project SC-17, Ohio 
State Uniyersity, Aug. 31, 1942. Div. 13-104.1-MI 

C-ll Antenna Patterns for Aircraft, George Sin¬ 
clair, NDCrc-100, Service Project SC-17, Ohio State 
University,: Aug. 24, 1943. Div. 13-104.1-M2 




282 


BIBLIOGRAPHY 


2. A New Method for Measuring Dielectric Constant 
and Loss in the Range of Centimeter Waves, S. Rob¬ 
erts and Arthur R. von Hippel, OEMsr-262, The 
Massachusetts Institute of Technology, March 1941. 

CP-521-M1 

3. “Electrical Measurements at UHF,” Ronold King, 
Proceedings of the Institute of Radio Engineers, 
Vol. 23, August 1935, pp. 885-934. 

4. “Coupled Networks in Radio Frequency Circuits,” A. 
Allford, Proceedings of the Institute of Radio En¬ 
gineers, Vol. 29,1941, p. 59. 

5. “Tie Analogie zwischen Sende und Empfangsan- 
tennen,” K. Franz, Hochfrequenztechnik und Elek- 
troakustik, Vol. 56, 1940, pp. 118-119. 

6. Electromagnetic Theory, J. A. Stratton, McGraw- 
Hill Book Company, Inc., 1941. 

7. “Reflexion und Absorption von Dezimeterwellen an 
evenen, dielektrischen Schichten,” Hochfrequenz¬ 
technik und Elektroakustik, Vol. 51, 1938, pp. 156- 
162. 

8. Reflexion am Geschlichtenen Medium,” W. Pfister 
and O. H. Roth, Hochfrequenztechnik und Elektro¬ 
akustik, Vol. 51, 1938, pp. 156-162. 

9. “A Survey of Ultra-High Frequency Measure¬ 
ments,” L. S. Nergaard, RCA Review, October 1938. 

10. “Scheinwiderstandsmessungen im Dezimeterwellen- 
Gebiet,” H. Kaufman, Hochfrequenztechnik und 
Elektroakustik, Vol. 53, 1939, pp. 61-67. 

11. Communications Network, E. A. Guillemin, John 
Wiley and Sons, Inc., Vol. II, 1935, p. 52. 

12. “Das Paralleldrahtsystem als Messinstrument in der 
Kurzeellentechnik,” O. Schmidt, Hochfrequenztech¬ 
nik und Elektroakustik, Vol. 41, 1933, pp. 2-16. 

13. “Resonance Curve Method for the Absolute Measure¬ 
ment of Impedance at Frequencies of the Order of 
300 mc/sec,” R. A. Chipman, Journal of Applied 
Physics, Vol. 10, January 1939, pp. 26-38. 

14. “A Generalized Reciprocity Theorem for Transmis¬ 
sion Lines at UHF,” Ronold King, Proceedings of the 
Institute of Radio Engineers, Vol. 28, May 1940, pp. 
223-225. 

15. “A Generalized Coupling Theorem for UHF Cir¬ 
cuits,” Ronold King, Proceedings of the Institute of 
Radio Engineers, Vol. 28, February 1940, pp. 84-87. 

Chapter 18 

1. Airborne Antenna Design at VHF and UHF, R. S. 
Wehner, OSRD 4794, OEMsr-1396, Project 13-105, 
Radio Corporation of America, Dec. 2, 1944. 

Div. 13-104-MI 

2. Antenna Pattern Measurements, P. S. Carter, Re¬ 
port CM-45-9, Radio Corporation of America, 
August 1944. 

3. Radio Engineers Handbook, F. E. Terman, Mc¬ 
Graw-Hill Book Company, Inc., 1943, pp. 865-869. 


4. Circular Loop Antennas at High Frequencies, P. S. 

Carter, OEMsr-895, Research Project RP-260, Re¬ 
port 895-31, Radio Corporation of America, Jan. 10, 
1945. Div. 15-333.21-M8 

5. Theory and Applications of Loop Antennas, Donald 
Foster, OEMsr-411, Research Project RP-107, Tech¬ 
nical Memorandum 411-TM-122, Harvard Uni¬ 
versity, Radio Research Laboratory, July 22, 1944. 

Div. 15-333.22-M3 

6. “Circular Loop Antennas at UHF,” J. B. Sherman, 
Proceedings of the Institute of Radio Engineers, 
Vol. 32, September 1944, p. 534. 

7. The Average Characteristic Impedance of Multiwire 

Cylindrical Cage Dipoles, W. C. Babcock, OEMsr- 
966, NDRC Report 966-8, Bell Telephone Laborato¬ 
ries, Inc., July 1943. Div. 15-333.1-M5 

8. A Flush Surface Antenna of the Slot-Cavity Type 

Having Wide Band Characteristics, N. E. Linden- 
blad, OEMsr-895, Research Project RP-260, NDRC 
Report 895-32, Radio Corporation of America, Mar. 
26,1945. Div. 15-333.53-MI 

9. “Rectangular Hollow-Pipe Radiators,” W. L. Bar- 
row and F. M. Green, Proceedings of the Institute of 
Radio Engineers, Vol. 26, December 1938, p. 1,498. 

Chapter 19 

1. Development of Faired-in Antennas for Naval Air¬ 
craft, K. S. Kunz, H. Faulkner, and others, OSRD 
6422, OEMsr-1441, Project 13-110, Problem 5, Har¬ 
vard University, Dec. 1, 1945. Div. 13-104-M5 

Chapter 20 

1. Location of Slot-Type AN/APN-1 Altimeter Anten¬ 
nas on Naval Aircraft, B. C. Dunn, W. D. Woo, and 
Ronold King, OSRD 6422, OEMsr-1441, Project 
13-110, Problem 7, Harvard University, Dec. 1, 1945. 

Div. 13-104-M5 

2. Study of Problems Arising from Closely Grouped 
Antennas, R. W. Grigg and W. R. Young, OSRD 
5503, OEMsr-1412, Project 13-103, Bell Telephone 
Laboratories, Inc., Aug. 1, 1945. Div. 13-104-M4 

3. Systems Engineering for Army Air Forces Com¬ 
munications, Part I, A. B. Clark, OSRD 1442, 
OEMsr-1018, Service Project AC-54, Report 2519, 
Bell Telephone Laboratories, Inc., Apr. 27, 1943. 

Div. 13-200.1-MI 

Systems Engineering for Army Air Forces Com¬ 
munications, Part II, A. Tradup, OSRD 1925, 
OEMsr-1018, Service Project AC-54, Bell Telephone 
Laboratories, Inc., Oct. 1, 1943. Div. 13-200.1-M2 
Systems Engineering for Army Air Forces Com¬ 
munications, Part III, A. Tradup, OSRD 4293, 
OEMsr-1018, Service Project AC-54, Bell Telephone 
Laboratories, Inc., Aug. 30, 1944. Div. 13-200.1-M3 

4. War Department Publication TM 11-486, Apr. 25, 
1945, Chapter 6. 

5. Study of Improvised VHF Antennas, H. W. Nylund 

and R. W. Grigg, OSRD 4604, OEMsr-1411, Project 
13-102, Bell Telephone Laboratories, Inc., and West¬ 
ern Electric, Dec. 30,1944. Div. 13-104-M2 




OSRD APPOINTEES 


Division IB 

Chief 

C. B. J olliffe (December, 1942 to May, 1945) 
Haraden Pratt (May, 1945 to May, 1946) 

Deputy Chief 

K. C. Black 

Technical Aides 

J. L. Allison J. F. McClean 

C. F. Dalziel A. F. Murray 


K. C. Black 
0. E. Buckley 
J. H. Dellinger 
W. L. Everitt 
G. C. Fick 


Members 

R. H. George 
C. H. G. Gray 
A. Hazeltine 
J. A. Hutchinson 
C. M. Jansky 
L. F. Jones 


D. G. Little 
R. K. Potter 
H. Pratt 
C. A. Priest 
F. M. Ryan 


Section 13.1 
Section 13.2 
Section 13.3 
Section 13.4 
Section 13.5 
Section 13.6 


Section Heads 
Direction Finding 
Radio Propagation Problems 
Speech Secrecy 

Special Communications Problems 
Precipitation Static 
Miscellaneous Projects 


L. F. Jones 
J. H. Dellinger 
R. K. Potter 

C. A. Priest 
H. Pratt 

D. G. Little 


Consultants 

L. V. Berkner E. D. Blodgett 

H. H. Beverage D. G. Little 

R. K. Potter 


Interference Reduction Committee 


K. C. Black 
H. D. Doolittle 
R. G. Fluharty 


A. Hazeltine 
J. C. R. Licklider 
C. T. Morgan 


A. F. Murray 
S. S. Stevens 
0. W. Towner 


CONTRACT NUMBERS, CONTRACTORS, AND SUBJECT OF CONTRACTS 


Contract Number 

NDCrc-100 

NDCrc-149 

NDCrc-155 

NDCrc-159 

NDCrc-193 

OEMsr-217 

OEMsr-263 

OEMsr-310 

OEMsr-338 

OEMsr-737 

OEMsr-745 

OEMsr-787 

OEMsr-838 

(Projects C-57, 

OEMsr-961 

OEMsr-1009 

OEMsr-1026 

OEMsr-1101 

OEMsr-1122 

OEMsr-1151 

OEMsr-1252 

OEMsr-1261 

OEMsr-1396 

284 


Name and Address of Contractor 


Refer to 

Subject Chapter 


C-78) 


The Ohio State University Research 
Foundation 
Columbus, Ohio 

Radio Corporation of America 
Princeton, New Jersey 

Western Electric Company, Inc. 

New York, New York 

Leland Stanford Junior University 
Stanford University, California 

Hazeltine Electronics Corporation 
Little Neck, New York 

California Institute of Technology 
Pasadena, California 

Federal Telephone and Radio Corporation 
New York, New York 

Western Electric Company, Inc. 

New York, New York 

Radio Corporation of America 
Camden, New Jersey 

General Electric Company 
Schenectady, New York 

Federal Telephone and Radio Corporation 
New York, New York 

Western Electric Company, Inc. 

New York, New York 

Radio Corporation of America 
Princeton, New Jersey 


Federal Telephone and Radio Corporation 
New York, New York 

Radio Corporation of America 
Camden, New Jersey 

Federal Telephone and Radio Corporation 
New York, New York 

University of Puerto Rico 
Rio Piedras, Puerto Rico 

Stanford University 

Stanford University, California 

Carnegie Institution of Washington 
College, Alaska 

Harvard University 
Cambridge, Massachusetts 

Raymond M. Wilmotte, 

Washington, D. C. 

Radio Corporation of America 
Rocky Point, New York 



Antenna patterns for aircraft 7 


H-F direction finder research 3 

H-F direction finder research 1 

Investigation of compensa- 3 
tion in direction finders 

U-H-F friendly aircraft lo- 5 
cator 

U-H-F Radio-Sonde direc- 5 
tion finder 

Transportable direction finder 3, 5 

Study of radio pulse propa- 1 
gation 

H-F direction finder 3 

Locating tanks by radio 5 

Study of direction finding 3 

fundamentals 

Locating tanks by radio 5 

Polarization errors of 3 

shielded-U Adcock direc¬ 
tion finders; the measure¬ 
ment of errors of radio 
direction finders 

U-H-F direction finding an- 5 

tenna study 

U-H-F direction finding 2 

study 

Study of effect of the earth 
on direction finding 

Correlation of D-F errors 4 

with ionospheric conditions 

Correlation of D-F errors 4 

with ionospheric conditions 

Correlation of D-F errors 4 

with ionospheric conditions 

Correlation of D-F errors 4 

with ionospheric conditions 

Portable radio assault beacon 5 

Optimum aircraft antenna 7 

patterns 










CONTRACT NUMBERS, CONTRACTORS, AND SUBJECT OF CONTRACTS ( Continued) 


Contract Number Name and Address of Contractor 


Refer to 
Subject Chapter 


OEMsr-1410 

OEMsr-1411 

OEMsr-1412 

OEMsr-1441 
Problem 1 

OEMsr-1441 
Problem 5 

OEMsr-1441 
Problem 7 

OEMsr-1441 
Problem 10 

OEMsr-1472 

OEMsr-1485 

OEMsr-1490 
Project C-18 
Project C-2-13 
Project 13-109 
Project 13.1-81 
Project 13.2-92 


Western Electric Company, Inc. 
New York, New York 

Western Electric Company, Inc. 
New York, New York 

Western Electric Company, Inc. 
New York, New York 


Central Communications Research 
Cruft Laboratory 
Harvard University 
Cambridge, Massachusetts 

Central Communications Research 
Cruft Laboratory 
Harvard University 
Cambridge, Massachusetts 

Central Communications Research 
Cruft Laboratory 
Harvard University 
Cambridge, Massachusetts 

Central Communications Research 
Cruft Laboratory 
Harvard University 
Cambridge, Massachusetts 

J. A. Maurer, Inc. 

New York, New York 

University of New Mexico 
Albuquerque, New Mexico 


Federal Telephone and Radio Corporation 
New York, New York 

National Bureau of Standards 
Washington, D. C. 

National Bureau of Standards 
Washington, D. C. 

Division 13, NDRC 
New York, New York 

National Bureau of Standards 
Washington, D. C. 

National Bureau of Standards 
Washington, D. C. 


Direction finding by impro- 4 
vised means 

Study of improvised anten- 7 
nas 

Study of problems arising 7 
from closely grouped an¬ 
tennas 

D-F standards 6 


Development of faired-in 7 
antennas 


Location of slot-type anten- 7 
nas 


Antennas for V-H-F, U-H-F, 7 
and S-H-F communications 


Electrical direction finder 5 

evaluator 

Investigation of practica- 6 

bility of direction finding 
on storms 

Miscellaneous current direc- 3 
tion finding problems 

H-F direction finder re- 3 

search 

Coordination of D-F with 4 

ionospheric measurements 

Survey of airborne direction 6 
finders 

Facilities for D-F research 3 

Correlation of D-F errors 4 

with ionospheric conditions 


285 










SERVICE PROJECT NUMBERS 


The projects listed below were transmitted to the Executive 
Secretary, NDRC, from the War or Navy Department through 
either the War Department Liaison Officer for NDRC or the 
Office of Research and Inventions (formerly the Coordinator of 
Research and Development), Navy Department. 


Service 

Project 

Number 

Subject 

Refer to 
Chapter 

AC-54 

Correlation of DF errors with ionospheric conditions 

4 

AN-22 

Survey of airborne direction finders 

6 

AN-30 

Direction finder standards 

6 

NA-212 

Development of faired-in antennas 

7 

NA-236 

Location of slot-type antennas 

7 

SC-7 

H-F direction finder research 

3 

SC-7 

Transportable direction finder 

3,5 

SC-7 

H-F direction finder research 

1 

SC-7 

H-F direction finder 

3 

SC-7 

Study of direction finding fundamentals 

3 

SC-7 

U-H-F direction finding antenna study 

5 

SC-7 

U-H-F direction finding study 

2 

SC-7 

Study of the effect of the earth on direction finding 

3 

SC-7 

Miscellaneous current direction finding problems 


SC-17 

Antenna patterns for aircraft 

7 

SC-31 

Locating tanks by radio 

5 

SC-87 

Portable radio assault beacon 

5 

SC-123 

Investigation of the practicability of direction finding 



on storms 

6 

SC-130 

Electrical direction finder evaluator 

5 

SC-142 

Antennas for V-H-F, U-H-F and S-H-F communications 



equipment 

7 


286 







INDEX 


The subject indexes of all STR volumes are combined in a master index printed in a separate volume. For 
access to the index volume consult the Army or Navy Agency listed on the reverse of the half-title page. 


Adcock antenna systems 
accurate bearings, 9-10 
amplitude-comparison method, 9- 
12 

comparison between differentially 
connected screen arrays and 
H Adcocks, 88-90 
compensating networks, 9 
definition, 9-10 
direct-reading system, 10-12 
disadvantage, 115 
elevated-counterpoise, 101-104 
generation of pure fields, 38 
history, 3, 24 

Holmdel Adcock system, 101-104 
measurement of azimuthal an¬ 
gles, 133 

polarization error measurement, 
69-71 

rotatable balanced H antenna, 
typical calculation, 34-35 
shielded-U, 100-101 
“swinging” bearings, 107-108 
Air drag, aircraft antennas, 263- 
266 

drag reduction means, 264-265 
electrical considerations, 265 
form drag, 263-264 
frictional drag, 263 
low-drag antennas, 265 
measurement of drag, 264 
Airborne direction-finder survey, 
122-123 

Aircraft antenna design 
air drag, 263-266 
antenna patterns at various fre¬ 
quencies, 204 

balanced antennas, 210-211 
electrically short transmitting 
antennas, 221-222 
elliptically polarized fields, 211 
field patterns, 228-238 
horizontal polarization, 248-254 
impedance measurements, 222- 
223 

measurement methods for pat¬ 
terns, 207-210 
models, 203, 211-213 
patterns for u-h-f and v-h-f, 219- 
266 

power capacity, 261-262 
precipitation static, 262 
propeller modulation, 211 


radiation characteristics, 224- 
225 

receiving antenna, 222-224 
resonant lines, 220 
spacing, 4, 275-276 
SWR effect on line voltage, 220 
transmission line losses, 220-221 
transmitting antenna character¬ 
istics, 221-222 
typical patterns, 204-207 
“uniform” horizontal plane pat¬ 
terns, 249-252 

vertical polarization, 238-248 
Aircraft antenna types 
see also Direction finders 
broad band, 239-244 
faired-in for naval aircraft, 267- 
273 

fish-hook, 254 

for horizontal polarization, 248- 
254 

for “uniform” horizontal plane 
patterns, 249-252 
for vertical polarization, 238-248 
less-than-quarter-wavelength,247 
stingeree, 255-256 
surface antennas, 256-261 
trailing-wire, 255 
Aircraft locator, friendly 

see U-h-f friendly aircraft lo¬ 
cator 

Altimeter (AN/APN-1), antenna 
location, 274-275 
Amplidyne servo system, 97 
Amplitude-comparison method of 
direction finding, 3, 9-15 
Adcock antenna, 9-12 
crossed buried U antenna system, 
12-15 

loop antenna, 9 

Anomalous effects in radio propa¬ 
gation at high frequencies, 
227 

Antenna location for AN/APN-1 
altimeter, 274-275 
Antenna patterns, reciprocity theo¬ 
rem, 217-218 

Antenna patterns for aircraft 
see Aircraft antenna design; 
Field patterns of aircraft 
antennas 

Antenna spacing, 4, 275-276 


Antenna transformers and buried 
conductors, 12-14 
Antennas, Adcock 

see Adcock antenna systems 
Antennas, aircraft 

see Aircraft antenna design; Air¬ 
craft antenna types 
Antennas, broadside cage Musa, 6 
Antennas, crossed buried U sys¬ 
tem, 12-15 

Antennas, disguised, 276-277 
Antennas, improvised 

see Improvised direction finders 
Antennas, rotatable balanced H an¬ 
tenna, 34-35 
Antennas, tank, 213-215 
Antennas, u-h-f 

see U-h-f direction finding 
Arrays 

balanced-to-unbalanced trans¬ 
formers, 90-92 

comparison between V and flat 
arrays, 87-88 
flat array, 79-85 
indicators, 86-87 
switches, 85-86 
V-l array, 60-77 
V-2 array,. 77-80 

Assault beacon, British, 160-161, 
163, 167-169 

Assault beacon, portable radio 
see Portable radio assault beacon 
Assault bearer, British, 163 
Azimuth angle determination of 
incoming wave, 133-134 

Balanced-to-unbalanced transform¬ 
ers, V and flat arrays, 84, 90- 
92 

Beacon, portable radio assault 
see Portable radio assault beacon 
Bell Telephone Laboratories 

see BTL high-frequency direc¬ 
tion finders 

British assault beacon, 160-161, 
163, 167-169 

Broad-band antennas for aircraft, 
239-244 

fan antennas, 242-243 
inverted-L antenna, 243-244 
sleeve antenna, 240 
u-h-f cone type, 239-240 
whip antennas, 240-242 


287 




288 


INDEX 


Broadside cage Musa system, 6 
Brown’s antenna, 252 
BTL high-frequency direction find¬ 
ers, 3-21 

amplitude-comparison method, 9- 
12 

background material, 3-4 
conclusions, 16 

crossed buried U antenna system, 
12-15 

d-f system, 17-20 
phase-comparison method, 3-9 
range extension, 20-21 
receiver specifications, 16-17 
Signal Corps adaptation, 20 
tests on complete system, 15-16 
wave error, 21 

Buried cables, required depth, 51 
Buried U direction finders 

see Crossed buried U antenna 
system 

Capacitive goniometer for u-h-f d-f 
antenna, 173-175 
Carter antenna, 259-260 
Cathode-ray indication and auto¬ 
matic control, 97-99 
amplidyne servo system, 97 
CR tube, 97, 99 
indication presentation, 97-99 
L-R indicator meter, 97 
resistor, 97 

Circular array, phase comparison 
method, 5-6 

Circular polarization, fish-hook air¬ 
craft antenna, 254 
Collins Radio Company, 42, 43 
Collector parallax, 43 
Compensated-loop direction finder, 
116-118 

Corner-type reflectors, shielding 
properties, 129-135 
Counterpoise tests, 101-104 
CR tube, 97, 99 

Crossed buried U antenna system, 
12-15 

antenna transformers and buried 
conductors, 12-14 
bearing error, 14 
broad-band balanced-to-unbal- 
anced transformers, 14 
construction details, 13-14 
frequency characteristics, 14 
injection-signal system receiving 
arrangements, 14-15 
polarization error measurement, 
36-38 

Cylindrical dipole impedance be¬ 
fore reflector, 94-97 
CXK direction finder, 3, 5 


DAB spaced-loop direction finder 
calibration, 120 

d-f error measurements, 119-121 
Demountable short-wave direction 
finder, 136-147 
antenna system, 138-139 
calibration, 144-145 
characteristics and advantages, 
136-137 

choice of site, 143 
description of equipment, 137 
goniometer drive units, 141 
indicators, 138, 142-143 
interpretation of patterns, 145- 
147 

operation, 137-143 
remote indicator assembly, 142- 
143 

SCR-502 d-f system, 136 
sense circuit, 138-141 
synchronization system, 141-142 
wave collectors, 138-139 
Differentially connected screen ar¬ 
rays and H Adcocks, com¬ 
parison, 88-90 

Dipole dimensions and impedance 
characteristics, 62-64 
design of corner array, 63 
dipole limitations, 63-64 
impedance considerations, 64 
statement of problem, 62-63 
Dipole spacing for flat array, 82-83 
Dipoles, horizontal electric and 
magnetic, 40-42 

Dipoles, vertical electric and mag¬ 
netic, 40-42 
Direction finders 
see also Adcock antenna systems; 
Antennas 

BTL high-frequency systems, 3- 
21 

buried U, polarization errors, 36- 
38 

CXK, 3, 5 

DAB spaced-loop, 119-121 
demountable short-wave, 136-147 
improvised, 148-159, 276-277 
loop, 114-118 

NBS high-frequency system, 22- 
54, 68-72 

radio-sonde (u-h-f), 127-135 
rotatable balanced H antenna, 34- 
35 

spaced-antenna, 24 
V-l array, 76-77 
Direction finding, u-h-f 

see U-h-f direction finding 
Direction-finder errors 

see Errors in direction finders 


Direction-finder evaluator, electri¬ 
cal 

see Electrical direction-finder 
evaluator 

Direction-finder receivers, 17-18 
Direction-finder survey for aircraft, 
122-123 

Direction-finding methods 
amplitude comparison, 3 
phase comparison, 3 
Direct-reading system with crossed 
Adcock antennas, 10-12 
Disguised antennas, 276-277 
Double wire antenna method, 153- 
155 

Downcoming sky waves, polariza¬ 
tion errors 

see Polarization errors for down¬ 
coming sky waves 

Electrical circuit for u-h-f direction 
finders, 178-182 
band I wave collector, 178 
band II wave collector, 180-181 
goniometer, 179-181 
indicator unit, 181-182 
receiving unit, 181 
Electrical direction-finder evalua¬ 
tor, 190-194 

basic principles and mechanisms, 
192-193 

data obtaining operation, 194 
description, 190 
d-f bearing input, 194 
geometric evaluation, 191 
gnomonic chart distortion cor¬ 
rection, 193 

group d-f system of evaluation, 
191 

pantographs, 193 
visual d-f evaluation, 190 
Electrical elements for flat array 
see Flat array, electrical elements 
Electrically short antennas for air¬ 
craft, 221-222 

Elevation angle determination of 
incoming wave, 133-134 
Elevation angles measured by di¬ 
pole antenna, 136 
Errors in direction finders, 100-121 
correlated with ionosphere meas¬ 
urements, 119-121 
counterpoise tests, 101-103 
ground characterstics study, 
112-114 

loop direction-finder errors, 144- 
118 

polariscope, 107-112 
polarization errors, 103-104 




INDEX 


289 


sRielded-U Adcock direction 
finder errors, 100-101 
wave interference errors, 110- 
112 

Errors in direction finders, meas¬ 
urement techniques, 104-107 
see also Polarization error, 
measurement 

distant-signal observations, 105 
frequencies, 105 
local signal sources, 105 
output-ratio method, 106 
purity of transmitter polariza¬ 
tion, 106 
signal field, 105 
signal source, 106 
site selection, 107 
test source supported from air, 
107 

types of errors, 104 
unwanted emission, 106 
Evaluator, direction-finder 
see Electrical direction-finder 
evaluator 

Fading effects, 26-31 
bearing error, 26-27 
causes, 26 

distribution law, 26-27 
improvements proposed, 27 
phase interference, 26 
Faired-in antennas for naval air¬ 
craft, 267-273 
bar antennas, 268-269 
conclusions, 272-273 
current exciting devices, 267-271 
measurement methods, 270-272 
slot antennas, 267, 270 
Fan antennas for aircraft, broad 
band, 242-243 

Field intensity meter, 44-45 
Field patterns of aircraft antennas, 
228-238 

absolute vs. relative field strength 
patterns, 231 

comparison with ideal antenna 
pattern, 233-234 
conclusion, 238 
cross-polarization, 238 
data presentation, 231 
effect of near-by structure, 238 
flight measurements, 228-229 
model measurements, 229 
pattern calculation, 229 
polarization defined, 230-231 
vertical antennas, 234-238 
Fish-hook antenna for circular 
polarization (aircraft), 254 
Fixed multiple antenna systems 
for improvised d-f, 152-155 
double wire system, 153-155 


fading signal, 153 
failure symptoms, 152 
receiving site, 152 
Flat array, 80-85 

operation theory, 79-81 
physical arrangement, 81-82 
Flat array, electrical elements 
amplitude ratio, 82 
balanced-to-unbalanced trans¬ 
former, 84 

dipole spacing, 82-83 
gain of array, 85 
impedance characteristics, 85 
polarization errors, 85 
reflector dimensions, 84 
relative response in azimuth, 84 
spurious response lobes, 82 
transmission lines, 83-84 
Free-space pattern for systems us¬ 
ing reflectors, 71-72 
Fresnel equations, 27-30 
Fresnel plane wave reflection co¬ 
efficient, 74 

Friendly aircraft locator, u-h-f 
see U-h-f friendly aircraft loca¬ 
tor 

Gain control by injection signal, 
14-15 

Goniometer for u-h-f direction find¬ 
ers, 173-175, 179-181 
Ground characteristics for d-f in¬ 
stallations, 112-114 
antennas and transmission lines, 
113-114 

audio-frequency methods, 114 
bridge circuit measurement, 113 
resistance as a function of depth, 
114 

r-f measurements, 113 
soils as dielectrics, 113 
treatment recommended, 114 
Wenner-Gish-Rooney method, 114 
Ground constants, measurement, 
51-54, 92-94 

Ground reflection, effect on total 
field, 27-31 
equations, 27-31 
magnetic field components, 31 
optimum height for antenna sys¬ 
tem, 31 

total field components, relative 
values, 30 

total field intensity, 30-31 

H Adcock antenna 
see Adcock antenna systems 
Height of V-l array, 74-75 

electromagnetic wave propaga¬ 
tion, 74 



Fresnel plane wave reflection co¬ 
efficient, 74-75 
interference phenomena, 74 

High-frequency direction finding 
see also BTL high-frequency di¬ 
rection finders; NBS high- 
frequency direction finders 
radio pulse propagation, 55-58 
ultra-high frequency, 59-99 

Holmdel 

antenna location, 15-16 
counterpoise tests, 101-104 
Musa receiving equipment, 6, 56, 
112-113 

polarization error, 101-104 
tests on horizontally polarized 
waves, 16 

Horizontal electric and magnetic 
dipoles, 40-42 

Horizontal polarization, aircraft 
antennas, 248-254 
broad-band balance transform¬ 
ers, 249 

Brown’s antenna, 252 
flush-mounted antennas, 254 
A/2 dipole type antennas, 252-253 
multiple antennas, 254 
polyphase antennas, 253-254 
“uniform” horizontal plane pat¬ 
tern, 249-252 
wire dipoles, 253 

Impedance characteristics for V-l 
array, 62-64 

Impedance of cylindrical dipole be¬ 
fore reflector, 94-97 

Impedance measurements of an¬ 
tennas by means of models, 
216-217 

Improvised direction finders, 148- 
159, 276-277 

antenna selection, 148, 276-277 
experimental work, 148-155 
fixed multiple antenna systems, 
152-155 

general operating notes, 155 
loop antenna, 148-151 
low horizontal wires as direc¬ 
tional antennas, 151 
possible refinements, 155 
supplementary tests, 155 
test procedure for antenna 
schemes, 148 

theoretical development, 155-159 
walked wires, 151-152 

Incoming wave, elevation angle de¬ 
termination, 133-134 

Indicator for u-h-f direction find¬ 
ers, 181-182 

Indicators for V and flat arrays, 
86-87 





290 


INDEX 


Injection-signal system, 14-15 
Interaction effects in phase-com¬ 
parison method of direction 
finding, 6-8 

antenna arrangement, ground 
plan, 6 

fixed antenna, amplitude of cur¬ 
rent, 7 

interaction among antennas, 6-8 
vertical cage antennas, 6 
Interservice Radio Propagation 
Laboratory (IRPL), 119 
Inverted-L antenna for low u-h-f, 
243-244 

Ionosphere measurements, correla¬ 
tion with d-f errors, 119-121 
auroral absorption zone, influence 
on bearing accuracy, 120 
bearing deviation measurements, 
119-121 

geomagnetic disturbances, 120 
information sources, 119-120 

Less-than-quarter-wavelength air¬ 
craft antennas, 247 
bent half-folded-dipole antenna, 
247 

dielectric antenna, 247 
helical antenna, 247 
inverted-L antenna, 247 
Lindenblad’s antennas, 258-259 
broad-band slot antenna, 258-259 
double-slot antenna, 258 
Lobe intersection determination, 
V array, 65-66 

Local transmitter measurements 
generation of pure fields, 38-39 
methods, 38 
Loop antennas 

as amplitude-comparison type, 9 
for improvised direction finding, 
148-151 

for “uniform” horizontal plane 
pattern, 252 

Loop direction finder, errors, 114- 
118 

compensated-loop direction find- 
* er, 116-118 
conclusions, 118 
coupling network, 117-118 
modes of attack, 115-116 
normal loop operation, 115 
polarization errors, 115 
results obtained, 116 
wave errors, 115 
L-R indicator meter, 97-99 

Maxwell’s equations, 25, 103 
McPetrie normal incidence method 
for determining ground con¬ 
stants, 92 


Model planes for antenna pattern 
measurements, 229 
Models used for impedance meas¬ 
urements, 216-217 
Models used for tank antenna pat¬ 
terns, 213-215 

Multiple antenna systems in im¬ 
provised direction finders, 
152-155 

Multiple-resonant antennas, 248 
Musa system, 6, 56, 112-113 

National Bureau of Standards 
see NBS high-frequency direc¬ 
tion finders 

Naval aircraft, faired-in antennas 
see Faired-in antennas for naval 
aircraft 

Navy’s CXK direction finder, 3, 5 
NBS high-frequency direction find¬ 
ers, 22-54, 68-74 
experimental technique, 44-49 
fading effects, 26-31 
historical development, 24 
objectives, 22 

polarization error, measurement, 
35-44, 68-74 

polarization errors, analysis, 22- 
35 

site problems, 49-54 

Pattern interpretation, demount¬ 
able short-wave d-f, 145-147 
Patterns, aircraft antenna 

see Aircraft antenna design; 
Field patterns of aircraft 
antennas 

Phase-comparison method of direc¬ 
tion finding, 3-9 
circular array of antennas, 5-6 
components, 4-5 
future possibilities, 9 
interaction effects, 6-8 
phase difference, equation, 4 
Plane wave measurements, 35-36 
azimuthal response patterns, 36 
pickup factors, definition, 36 
sample treatment, 36 
Poast method of measuring polari¬ 
zation errors, 72-74 
construction, 72 
objections, 72-73 
operation, 72 
Polariscope, 108-112 

analysis of observations, 110 
construction, 108-110 
operating procedure, 110 
Polarization definition, 230-231 
Polarization error, measurement, 
32-49, 69-74, 103-104 
Adcock antenna, 69-71 



azimuthal angle, 32 
buried U direction finders, 36-38 
collector parallax, 43 
counterpoise tests, 101-104 
d-f systems examined, 44 
equations, 32-33 
experimental technique, 44-49 
field generated by local radiator, 
39-42 

field intensity, 44-45 
fixed-type direction finders, 33 
horizontal dipole adjustment, 44 
local transmitter measurements, 
38-39 

maximum polarization error, 45 
NBS method, 35-44, 68-74 
plane wave measurements, 35-36 
Poast method, 72-74 
radiator parallax, 42-44 
systems using reflectors, 71-72 
test conclusions, 46-49 
typical calculation, 34-35 
Polarization errors for downcoming 
sky waves, 22-35 
collector parallax, 23 
fading effects, 26-31 
ground reflection, 27-31 
historical development, 24 
nature of downcoming waves, 
24-26 

pickup factor, 23 

pickup ratio, 23, 24 

radiator parallax, 23 

site selection, 23 

state of polarization, 22, 26 

theoretical aspects, 22-24 

total field, 23 

Polarization errors for flat array, 
85 

Polarization errors for loop direc¬ 
tion finder, 115-116 
Polyphase antennas for horizontal 
polarization, 253-254 
Portable radio assault beacon, 160- 
171 

A-N modulation, 164 
antenna, 165, 170 
automatic volume control (AVC), 
161 

British system, 160-161, 163, 168- 
169 

comparison of beacon systems, 171 
crossed-loop beacon, 162-163 
experiments, 162-169 
factors affecting operation, 164- 
165 

frequency selection, 161-162 
key click elimination, 160, 169-170 
modulation, 161, 163 
obstruction, 166 
polarization, 162, 166 




INDEX 


291 


selection of type, 160 
sky-wave effect, 167 
switching relay design, 169-170 
unequal currents, 166 
weather effect, 167 
Power capacity of aircraft anten¬ 
nas, 261-262 

Q-meter in soil resistivity measure¬ 
ment, 113 

Quarter-wave aircraft antennas for 
vertical polarization, 245-247 
Brown-Epstein antennas, 245-246 
half-folded-dipole antenna, 246 
skin-back antenna, 246 
thick stub antenna, 245 

Radiation characteristics of air¬ 
borne antennas, 224-225 
Radiator parallax, 42-43 
Radio assault beacon, portable 
see Portable radio assault beacon 
Radio location of tanks, 183-184 
Radio pulse propagation, 55-58 
experimental conclusions, 56-58 
experimental procedure, 56 
measurement results, 56-57 
objectives of project, 55 
sources of error, 55-56 
Radio wave propagation, u-h-f and 
v-h-f, 225-228 

anomalous effects at high frequen¬ 
cies, 227 

distance effect, 225-226 
elevation effect, 226 
frequency effect, 226 
ground-reflection coefficients, 225 
man-made interference, 227 
multi-path interference, 228 
polarization, 226-227 
propeller modulation, 228 
space wave, 225 

Radio-sonde direction finder (u-h-f) 
see U-h-f radio-sonde direction 
finder 

Rayleigh distribution law, 26-27 
Receiver specification, 16-18 
Receiving antennas for aircraft, 
222-224 

antenna efficiency as function of 
frequency, 223 
impedance matching, 223 
line input impedance, effect of 
line losses, 223-224 
statement of problem, 222-223 
weak signal reception, 224 
Reciprocity between transmitting 
and receiving antennas, 217 
Remote indicator assembly for de¬ 
mountable shortwave direc¬ 
tion finder, 142-143 


Research facilities for u-h-f studies, 
59-60 

Research recommendations 

airborne direction-finder survey, 
122-123 

surface antennas, 261 
Resistivity of soil, measurement 
methods, 113-114 
audio-frequency methods, 114 
bridge methods, 113 
methods using antennas and 
transmission lines, 113 
r-f measurements, 113 
W enner-Gish-Rooney methods, 114 
Rotatable balanced H antenna, 
typical calculation, 34-35 
d-f azimuthal directional pattern, 
34-35 
dipoles, 34 
equations, 34-35 
horizontal wave error, 35 
maximum polarization error, 35 
pickup ratio, 34-35 
standard wave error, 35 
undesired response, 34-35 
Rotating antenna system for u-h-f 
direction finding, 175 

Schelkunoff’s formula for imped¬ 
ance of cylindrical dipole, 
94-97 

SCR-502 d-f system 

see Demountable short-wave di¬ 
rection finder 

Sferics and weather information, 
197-200 

conclusions, 200 
equipment in mobile unit, 198 
equipment utilized, 197-198 
lighting flashes, 199 
potential gradient change re¬ 
corder, 197 

purpose of project, 197 
waveform analysis, 198-199 
Shield oxidation, 134 
Shielded-U Adcock direction finder 
errors, 100-101 

Shielding properties for corner-type 
reflectors, 129-135 
Short-wave direction finder, de¬ 
mountable 

see Demountable short-wave di¬ 
rection finder 

Signal Corps adaptation of BTL di¬ 
rection finder, 20 
Signal source 
distant, 105 
local, 105 
self-contained, 106 
Site problems, 49-54 
bearing errors, 50-51 



buried cable depth, 51 
classification of problems, 49 
horizontal loop-antenna, 50 
measuring method for ground 
constants, 51-54 

Site selection for direction finder 
operations, 143 

Sites, good ground characteristics, 
112-114 

Sleeve antenna, 240 
Soil resistivity measurement, 113 
Spaced-antenna direction finder, 24 
“Standard wave error”, 24, 47, 48 
Stingeree aircraft antenna, 255-256 
Surface antennas for aircraft, 256- 
261 

conclusions, 261 
double-slot antenna, 258 
Lindenblad’s broad-band slot an¬ 
tenna, 258-259 

Lindenblad’s double-slot antenna, 
258 

louvre (wedge) antenna, 259-260 
semicylindrical cavities, 260 
single-slot antenna, 256-258 
waveguide antenna, 259 
Surface wave component, 39-40 
equations, 39 
practical importance, 39 
space wave, 39 

“Swinging” bearings, causes, 107- 
108, 110-112 
Switched cardioids, 81 
Switches for V and flat arrays, 85- 
86 

Switching relay design for assault 
beacon, 169-170 

Tank antenna patterns, 213-215 
Tank location by radio, 183-184 
accuracy improvement, 184 
SCR-508 tests, 183-184 
simplified radar method, 184 
Trailing-wire aircraft antennas, h-f 
and low v-h-f, 255 
Transformers, broad-band bal- 
anced-to-unbalanced, 14 
Transformers, V and flat arrays, 
84, 90-92 

Transmitter measurements, local, 
38-39 

Transmitting antennas for aircraft, 
221-222 

Type B indicator for u-h-f direction 
finders, 181-182 

U-h-f aircraft antenna design 
see Aircraft antenna design 
U-h-f cone type antenna, 239-240 
U-h-f direction finding, 59-99, 172- 
182 

antenna performance, 176 




292 


INDEX 


cathode-ray indication and auto¬ 
matic control, 97-99 
comparison between differentially 
connected screen arrays and 
H Adcocks, 88-90 
comparison between V and flat 
arrays, 87-88 

design of balanced-to-unbalanced 
transformers, 90-92 
determination of ground con¬ 
stants, 92-94 

electrical circuit theory, 178-182 
final antenna designs, 175 
flat array, 80-85 
goniometer, 173-175, 179-181 
impedance of cylindrical dipole 
before reflector, 94-97 
introduction, 59 
problems encountered, 172 
receiver, 174, 181 
research facilities, 59-60 
rotating collector, 175 
shielding, 174 

switching and indicating devices, 
85-87 

system experiments, 172-175 
V-l array, 60-77 
V-2 array, 77-80 

U-h-f friendly aircraft locator, 185- 
189 

antennas, 185-186 
apparatus limitations, 188 
cathode ray indicator, 187-188 
components, 185 
line receiver, 187 
line transmitter goniometer unit, 
186-187 

u-h-f receiver, 186 
U-h-f propagation 
see Radio wave propagation, 
u-h-f and v-h-f 


U-h-f radio-sonde direction finder, 
127-135 

azimuthal and elevation angle 
determination of incoming 
wave, 133-134 

copper screening as shield, 133 
description of apparatus, 127-129 
Fort Monmouth experiments, 135 
object of development, 127 
shield oxidation, 134 
shielding properties of corner- 
type reflectors, 129-131 
test results, 129-135 
Ultra-high-frequency direction find¬ 
ing 

see U-h-f directing finding 
“Uniform” horizontal plane pattern 
aircraft antennas, 249-252 
bent-sleeve dipole antenna, 249- 
250 

coax-fed V-dipole antenna, 250 
loop antennas, 252 
split-can antenna, 251 

V-l array, 60-77 
construction, 60 
design, 76 

determination of lobe intersec¬ 
tion, 65-66 

dipole dimensions and impedance 
characteristics, 62-63 
directivity in azimuth, 64-65 
electrical balance, 76 
gain, 75-76 

optimum height selection, 74-75 
polarization errors, 67-74 
reflectors, 61 

relative response in elevation, 67 
support pole effect, 76-77 
V-2 array, 77-80 
comparison with V-l; 77-79 




experimental work, 77 
gain, 79 

impedance characteristics, 79 
polarization errors, 79 
relative response in azimuth, 77- 
79 

V and flat arrays 
comparison, 87-88 
switches, 85-86 
Vertical cage antennas, 6, 56 
Vertical electric and magnetic di¬ 
poles, 40 

Vertical polarization, aircraft an¬ 
tennas, 238-248 

broad-band antennas, 239-244 
half-wave grounded-loop antenna, 
247 

less-than-quarter-wavelength an¬ 
tennas, 247 

quarter-wave antennas, 245-247 
V-h-f aircraft antenna design 
see Aircraft antenna design 
V-h-f antennas, improvised, 276 
V-h-f propagation 
see Radio wave propagation, 
u-h-f and v-h-f 

Walked wires for improvised direc¬ 
tion finding, 151-152 
Wave collectors, 138-139 
Wave components, 25-26 
Wave interference errors, 110-112 
Weather information 

see Sferics and weather informa¬ 
tion 

Wenner-Gish-Rooney method for 
site selection, 114 

Whip antennas for aircraft, 240-242 


















DECLASSIFIED 
By authority Secretary of 

SEP 2 31960 

Defense memo 2 August 1960 
LIBRARY OF CONGRESS 





